Prototype of the Front-end Circuit for the GOSSIP (Gas On Slimmed Silicon Pixel) Chip in
the 0.13um CMOS Technology.
V.Gromov, R.Kluit, H. van der Graaf.
NIKHEF, Kruislaan 409, Amsterdam, the Netherlands.
vgromov@nikhef.nl
Abstract.
Owing to a novel concept of the detection of the single
electrons in gas, the GOSSIP chip will hold certain
advantages over an ordinary silicon pixel readout chip. Of
these, no need for silicon sensor at all, low detector parasitic
capacitance and none of the bias current at the pixel are the
attractive features to design a compact low-noise and lowpower integrated front-end circuit.
A prototype of the integrated circuit has been developed in
the 0.13um CMOS technology. The prototype includes a few
channels equipped with the preamplifier, the discriminator
and the digital circuit to study the feasibility of the TDC-perpixel concept.
The design demonstrates very low input referred noise
(60e RMS) in combination with a fast peaking time ( 40ns)
and low analog power dissipation (2uW) per channel for 1.2V
supply. High frequency switching activity on the clock bus
(up to 100MHz) in the close vicinity of the sensitive analog
inputs does not cause noticeable extra noise.
The value of the parasitic capacitance at the input of the
front-end circuit is determined by the area of the pixel pad and
consequently could be very low (5fF…30fF). This feature
enables to design a low-noise (60-80e rms) and the same time
very low power circuit (2uW per pixel).
The low power aspect is of primary importance since any
additional cooling system involves an increase of the material
budget. We expect the GOSSIP chip to dissipate 100mW/cm2.
In this work we have developed a prototype of the front-end
circuit in the 0.13um CMOS technology.
Cathode (drift) plane
Cluster1
Cluster2
When a minimum ionizing (MIP) particle passes the drift
gap, some 10-50 electron-ion pairs will be created along the
track. Driven by an electric field the electrons will drift
towards the pixels. In the InGrid-pixel gap an avalanche
multiplication occurs making the charge sufficient to activate
an on-pixel integrated circuit [2]. The activated pixels will
show the projection of the track on the array surface.
Moreover the drift time measurements at the activated pixels
will indicate the polar angle of the track.
A number of features make the GOSSIP chip
advantageous for future particle detectors. It has no thick
silicon sensor bulk (slimmed pixel chip). Therefore it has a
low material budget and is free from the radiation damage
effects taking place in the depletion layer of the silicon sensor.
The on-pixel circuit will be radiation hard due to the internal
properties of the up-to-date deep-submicron CMOS
technology.
Cluster3
InGrid
50um,
400Volts
Silicon read-out chip
Input pixel
I. Introduction.
The GOSSIP (Gas On Slimmed Silicon Pixel) detector
[1] (see Fig.1) is seen as a CMOS pixel array with a
Micromegas grid placed at the distance of 50um on top of it
by means of wafer post-processing technology (Integrated
Grid or InGrid). One mm above this grid a cathode foil is
built. The cathode foil and the grid are put at -800V and 400V, respectively, and the pixel array surface is at ground
potential. The volume between the drift foil and the pixel
array is filled with a suitable gas mixture.
1mm,
400Volts
50um
Figure 1:Layout of the GOSSIP detector.
II. Inputs and requirements for the design of the
readout circuit.
A. Single electron efficiency.
The simulations show, that we will reach the single
electron efficiency around 90% even at a low gas of 2000 in
the case if the readout circuit operates at a threshold of 400
electrons (see Fig.2). It means that ENC referred to the input
must not exceed this value.
1
Gain=8000
0.9
0.8
Gain=4000
0.7
0.6
0.5 0
Gain=2000
500
0
1000
1500
2000
Threshold, electrons
Figure 2: Single electron efficiency as a function of operating
threshold.
B. Shape of the detector current.
The shape in the detector current is formed by the
electron and the ion components. The ion component mostly
determines the shape and the magnitude of the signal due to
slow motion of the ions in the InGrid-to-pixel gap (see Fig.3).
i(t)
ion component
electron
component
5 - 30 ns
III. The prototype of the front-end circuit.
A. Parasitic capacitance at the input of the
front-end circuit.
Detector parasitic capacitance is the basic input
parameter for any front-end circuit design. It plays the leading
role in trade-off between speed, noise and power
consumption. By decreasing the detector capacitance value
we can design a faster circuit with the same noise and power
consumption or reduce power consumption keeping constant
noise and speed. In the GOSSIP chip, detector capacitance or
parasitic capacitance on the input pad consists of three
components. These are pad-to-InGrid capacitance (Cp-grid),
pad-to-pad capacitance (Cp-p) and pad-to-substrate capacitance
(Cp-sub) (see Fig.5).
Figure 3: Shape of the detector current.
C. Single electron drift time measurements.
According to simulations it is feasible to get time
resolution of the order of 2ns rms with a realistic gas gain. It
corresponds to the spatial resolution 100um rms (see Fig.4).
Gain=8000
Entries
Gain=4000
Gain=2000
0
2ns
4ns
6ns
8ns
10ns 12ns
Measured drift time
Figure 4: Simulations of the single electron drift time
measurements at various gas gain factors.
In order to measure drift time of every cluster on the
track each pixel must be equipped with an individual time-todigital converter.
D. Analog-digital crosstalk.
Data taking and readout will occur simultaneously and
constantly. The high sensitive analog front-end circuit will
operate in the close vicinity of the high speed switching gates.
This urges us to make every effort to keep the front-end away
from the switching noise.
E. Power consumption.
Low power dissipation let us minimize (or even exclude)
a cooling system and therefore essentially reduce material
budget. We intent not to go beyond 2uW per pixel or 100mW
per cm2.
InGrid
Cp-grid
Cp-grid
Cp-p
Cp-grid
Cp-p
Input pad
Substrate of the wafer
Cp-sub
Cp-sub
Figure 5: Parasitic capacitances associated with the input pad.
The pad-to-substrate capacitance is formed on the readout
chip. It depends on the pad size and has an essentially larger
value than the others for the very thin dielectric layer in the
body of the chip. With no need for silicon sensor and bonding
bumps a very small size of the input pad is possible. As a
result, a very low value of the input parasitic capacitance
(around 10fF) can be reached (see Fig.6).
50
40
Cp-sub, fF
20
10
0
0
10
20
30
40
50
Size of the pad, um
Figure 6: The pad-to-substrate capacitance as a function of the
size of the pixel.
B. Gain in the charge-sensitive preamplifier.
Charge-to-voltage gain of the charge sensitive
preamplifier (see Fig.7) is inversely related to the feedback
capacitance.
Charge-to-voltage gain ≈ Cfb-1 (1)
To get the biggest gain we should decrease the feedback
capacitance. However, its value must be larger than the
parasitic capacitance value divided by the open loop gain
factor as follows:
phase margin even when the input parasitic capacitance
approaches 10fF.
Cfb >> Cpar / A (2)
Since Cpar is 10fF and A is around 100, we can consider
the feedback capacitance value as low as 1fF.
Vdd=1.2V
M3
M4
M5
Cfb
Qin
Rfb
Iin(t)
Ib=1nA
Cfb
=1fF
A
Cpar
M2
M1
Vb1
Output
Open loop voltage gain
of the OPAMP
Input
OPAMP
Figure 7: The charge-sensitive preamplifier.
Vb2
We use a coaxial-like structure in order to make a
physical layout of such a tiny component requiring a high
quality (see Fig.8). In this structure fringe or vertical plate
coupling forms the feedback capacitance. The high
performance is due to accuracy of lithography and thickness
of the metal layer. In addition, this layout let us electrically
isolate the high sensitive input.
M8
M6
M9
M7
Figure 9: Schematic of the charge sensitive preamplifier.
Input pad
Cpar = 10fF
LM
Ground planeM6
M3
Ground
Output
M2
M1
Substrate
Cfb=1fF
Parasitic metalto-metal fringe
capacitances.
Figure 8: Layout of the input interconnection.
C. Schematic of the preamplifier.
The scheme proposed by Krummenacher [3] is a common
way to implement a preamplifier for hybrid pixel detectors.
The circuit amplifiers the input signal and compensates for the
leakage current of the silicon sensor. However, it will become
unstable if parasitic capacitance at the input is very low.
The GOSSIP detector has no leakage current. This let us
modify Krummenacher’s scheme as to improve stability (see
Fig.9) We have fixed the current in the load of the differential
pair M1-M2. As a result, the circuit will demonstrate a safe
DC feedback in the circuit allows for discharging of the
feedback capacitor. The virtual resistor is a sum of inverse
transconductances of transistors M1 and M2 takes the value
80 MΩ. It also biases the input of the circuit so as to keep the
voltage at the output end equal to the reference voltage.
Because of use of differential stages and floating current
sources the biasing is highly insensitive to temperature drift
and power supply voltage instability. Statistical spread of the
offset at the output is 20mV rms corresponding to the input
referred signal of 170 electrons.
Operational amplifier (see Fig.9) is formed by a cascode
differential pair, loaded with a current mirror (see Fig.10). A
voltage follower provides a low output impedance of the
amplifier. The input-to-output small signal transfer function
has a pole with time constant 14ns. The DC gain of the
amplifier is 130. The circuit draws about 1.2uA current from
the 1.2V power supply source. The overall power
consumption is 1.5uW.
characteristics of the analog devices. There is a very nice
feature in the 0.13um CMOS technology. We are allowed to
make a floating p-well separated from the substrate. In the pwell we can place analog N-type transistors and keep them
away from the substrate noise (see Fig.12). By this means the
analog-digital crosstalk has been essentially put down.
Vdd=1.2V
M5
M6
M7
Output
M3
Ib2=0.2uA
M4
Digital N-type Analog P-type Analog N-type
FET area
FET area
FET area
Guard rings
P-well
N-well
substrate current
Vb3
Input
M1
P-type substrate
M2
Figure 12: Triple well layout in the 0.13um CMOS technology.
Vb2
E. Measurements.
Ib1=1uA
The measurements demonstrate that the preamplifier has
sufficient stability margin an expected shape of the pulse
response (see Fig.13). Equivalent input noise charge is 60
electrons rms.
Figure 10: Schematic of the operational amplifier.
D. Objectives for the first prototype.
In the first submit we intended to measure the most
important characteristics of the front-end circuit such as
stability, pulse response, noise and channel-to-channel spread.
For this purpose a bare preamplifier with a voltage follower
have been used. We also designed a complete front-end
including a CMOS comparator and a counter. This let us
study cross-talk between the high sensitive analog inputs and
the high speed switching CMOS blocks. A general view of the
physical layout of the prototype is given in Figure 11.
Figure 13: Delta-pulse response of the preamplifier. Input signal
is 410 electrons.
Delta pulse response at the output of the CMOS
comparator is given in Figure 14. No indication of afterpulsing or oscillation supports our contention that the parasitic
output-input crosstalk is very low.
Fig.11. The first prototype of the GOSSIP chip, submitted
on December 12, 2005.
Substrate noise is the most significant cause of the
analog-digital crosstalk. The local substrate potential will
fluctuate due to current flow originated by the switching of
the digital gates. The substrate fluctuations will affect the
The channel-to-channel spread of the threshold is 160
electrons rms. This result is consistent with simulation.
A new prototype featuring TDC-per-pixel concept will be
submitted in the end of 2006.
V. Acknowledgements.
The authors would like to thank M.Campbell, X.Llopart
and R. Ballabriga Sune of CERN, Geneva, Switzerland for
useful discussions, remarks and advices, Joop Rovekamp and
JJ.P. Fransen of NIKHEF, Amsterdam, the Netherlands for
technical support.
VI. References.
Figure 14: Delta-pulse response of the output of the CMOS
comparator. Input signal is 410 electrons. Threshold is 300electrons.
We also studied the analog-digital crosstalk by putting
the 100MHz clock on the CMOS counter clock in the close
vicinity of the high sensitive front-end. Even at very low
threshold no significant crosstalk at the output of the
comparator has been noticed (see Fig.15).
Figure 15 Output of the CMOS comparator along with the clock
signal running at 100MHz.
IV. Conclusions and plans.
Front-end readout circuit of the GOSSIP chip will benefit
from the low detector parasitic capacitance and no need to
compensate for the leakage current.
The first prototype of the fast (40ns peaking time), lownoise (ENC=60e rms) and low-power (2uw per channel)
front-end circuit has been successfully implemented in the
0.13um CMOS technology.
Owing to the triple well layout used in the prototype, the
high sensitive analog inputs have been effectively isolated
from the high speed switching gates.
[1] M.Campbell et al, “GOSSIP: A vertex detector
combining a thin gas layer as signal generator with a CMOS
readout pixel array”, Nucl. Instr. & Methods in Physics
Research, A560, pp.131-134, 2006.
[2] H. van der Graaf et al, “The detection of single
electrons by means of a Micromegas-covered MediPix2 pixel
CMOS readout circuit”, Nucl. Instr. & Methods NIM, A340,
pp.295-304, August 2004.
. [3] F. Krummenacher, “Pixel Detectors with Local
Intelligence: an IC Designer Point of View”, Nuclear
Instruments & Methods in Physics Research, A305, 1991,
pp.527-532, .