# Lecture 21 REMINDERS

```Lecture 21
REMINDERS
• Review session: Fri. 11/9, 3-5PM in 306 Soda (HP Auditorium)
• Midterm #2 (Thursday 11/15, 3:30-5PM in Sibley Auditorium)
OUTLINE
• Frequency Response
–
–
–
–
–
–
Review of basic concepts
high-frequency MOSFET model
CS stage
CG stage
Source follower
Cascode stage
EE105 Fall 2007
Lecture 21, Slide 1
Prof. Liu, UC Berkeley
Av Roll-Off due to CL
• The impedance of CL decreases at high frequencies, so
that it shunts some of the output current to ground.
1
 0
 
p
RD C L

1 

Av   g m  RD ||
jCL 

• In general, if node j in the signal path has a smallsignal resistance of Rj to ground and a capacitance Cj to
ground, then it contributes a pole at frequency (RjCj)-1
EE105 Fall 2007
Lecture 21, Slide 2
Prof. Liu, UC Berkeley
Pole Identification Example 1
 0
 p1
EE105 Fall 2007
1

RG Cin
 p2
Lecture 21, Slide 3
1

RD C L
Prof. Liu, UC Berkeley
Pole Identification Example 2
 0
 p1
1


1 
 RG ||
Cin
gm 

EE105 Fall 2007
 p2
Lecture 21, Slide 4
1

RD C L
Prof. Liu, UC Berkeley
Dealing with a Floating Capacitance
• Recall that a pole is computed by finding the resistance
and capacitance between a node and (AC) GROUND.
• It is not straightforward to compute the pole due to CF
in the circuit below, because neither of its terminals is
grounded.
EE105 Fall 2007
Lecture 21, Slide 5
Prof. Liu, UC Berkeley
Miller’s Theorem
• If Av is the voltage gain from node 1 to 2, then a
floating impedance ZF can be converted to two
grounded impedances Z1 and Z2:
V1  V2 V1
V1
1

 Z1  Z F
 ZF
ZF
Z1
V1  V2
1  Av
V1  V2
V2
V2
1

 Z 2  Z F
 ZF
ZF
Z2
V1  V2
1 1
EE105 Fall 2007
Lecture 21, Slide 6
Av
Prof. Liu, UC Berkeley
Miller Multiplication
• Applying Miller’s theorem, we can convert a floating
capacitance between the input and output nodes of
an amplifier into two grounded capacitances.
• The capacitance at the input node is larger than the
original floating capacitance.
ZF
Z2 
1 1
1
1
jCF


1
1
j 1  1 CF
Av
Av
Av 

1
ZF
1
jC F
Z1 


1  Av
1  Av
j 1  Av C F
EE105 Fall 2007
Lecture 21, Slide 7
Prof. Liu, UC Berkeley
Application of Miller’s Theorem
 0
1
in 
RG 1  g m RD C F
EE105 Fall 2007
out 
Lecture 21, Slide 8
1

1 
C F
RD 1 
 g m RD 
Prof. Liu, UC Berkeley
MOSFET Intrinsic Capacitances
The MOSFET has intrinsic capacitances which affect its
performance at high frequencies:
1. gate oxide capacitance between the gate and channel,
2. overlap and fringing capacitances between the gate and the
source/drain regions, and
3. source-bulk &amp; drain-bulk junction capacitances (CSB &amp; CDB).
EE105 Fall 2007
Lecture 21, Slide 9
Prof. Liu, UC Berkeley
High-Frequency MOSFET Model
• The gate oxide capacitance can be decomposed into a
capacitance between the gate and the source (C1) and
a capacitance between the gate and the drain (C2).
– In saturation, C1  (2/3)&times;Cgate, and C2  0.
– C1 in parallel with the source overlap/fringing capacitance  CGS
– C2 in parallel with the drain overlap/fringing capacitance  CGD
EE105 Fall 2007
Lecture 21, Slide 10
Prof. Liu, UC Berkeley
Example
CS stage
EE105 Fall 2007
…with MOSFET capacitances
explicitly shown
Lecture 21, Slide 11
Simplified circuit for
high-frequency analysis
Prof. Liu, UC Berkeley
Transit Frequency
• The “transit” or “cut-off” frequency, fT, is a measure
of the intrinsic speed of a transistor, and is defined as
the frequency where the current gain falls to 1.
Conceptual set-up to measure fT
I out  g mVin
I in 
Vin
Z in
 1 
I out
  1
 g m Z in  g m 
I in
 jT Cin 
g
 T  m
Cin
gm
2f T 
CGS
EE105 Fall 2007
Lecture 21, Slide 12
Prof. Liu, UC Berkeley
Small-Signal Model for CS Stage
 0
EE105 Fall 2007
Lecture 21, Slide 13
Prof. Liu, UC Berkeley
… Applying Miller’s Theorem
 p ,in
1

RThev Cin  1  g m RD CGD 
 p ,out 
1




1
CGD 
RD  Cout  1 

g
R
m
D




Note that p,out &gt; p,in
EE105 Fall 2007
Lecture 21, Slide 14
Prof. Liu, UC Berkeley
Direct Analysis of CS Stage
• Direct analysis yields slightly different pole locations
and an extra zero:
gm
z 
C XY
1
 p1 
1  g mRD C XY RThev  RThevCin  RD C XY  Cout 

1  g m RD C XY RThev  RThev Cin  RD C XY  Cout 
 p2 
RThev RD CinC XY  CoutC XY  CinCout 
EE105 Fall 2007
Lecture 21, Slide 15
Prof. Liu, UC Berkeley
I/O Impedances of CS Stage
 0
Z in 
j CGS
EE105 Fall 2007
1
 1  g m RD CGD 
Z out 
Lecture 21, Slide 16
1
j CGD  C DB 
|| RD
Prof. Liu, UC Berkeley
CG Stage: Pole Frequencies
CG stage with MOSFET capacitances shown
 p, X
 0
1


1 
 RS ||
C X
gm 

C X  CGS  CSB
 p ,Y
1

RD CY
CY  CGD  CDB
EE105 Fall 2007
Lecture 21, Slide 17
Prof. Liu, UC Berkeley
AC Analysis of Source Follower
 0
vout
vin
CGS
1   j 
gm

2
a  j   b  j   1
EE105 Fall 2007
• The transfer function of a
source follower can be
obtained by direct AC
analysis, similarly as for
the emitter follower (ref.
Lecture 14, Slide 6)
RS
CGDCGS  CGDC SB  CGS C SB 
a
gm
b  RS CGD 
Lecture 21, Slide 18
CGD  C SB
gm
Prof. Liu, UC Berkeley
Example
vout
vin
CGS
1   j 
gm

2
a  j   b  j   1
RS
CGD1CGS1  (CGD1  CGS1 )(CSB1  CGD 2  CDB 2 )
a
g m1
CGD1  C SB1 C GD 2 C DB 2
b  RS CGD1 
g m1
EE105 Fall 2007
Lecture 21, Slide 19
Prof. Liu, UC Berkeley
Source Follower: Input Capacitance
• Recall that the voltage gain of a source follower is Av 
Follower stage with MOSFET capacitances shown
 0
RS
1
 RS
gm
• CXY can be decomposed into
CX and CY at the input and
output nodes, respectively:
C X  1  Av CGS 
CGS
1  g m RS
CGS
Cin  CGD 
1  g m RS
EE105 Fall 2007
Lecture 21, Slide 20
Prof. Liu, UC Berkeley
Example
0
1
Cin  CGD1 
CGS1
1  g m1 rO1 || rO 2 
EE105 Fall 2007
Lecture 21, Slide 21
Prof. Liu, UC Berkeley
Source Follower: Output Impedance
 0
• The output impedance of
a source follower can be
obtained by direct AC
analysis, similarly as for
the emitter follower (ref.
Lecture 14, Slide 9)
jRG CGS  1
vX

iX
jCGS  g m
EE105 Fall 2007
Lecture 21, Slide 22
Prof. Liu, UC Berkeley
Source Follower as Active Inductor
Z out
jRG CGS  1

jCGS  g m
CASE 1: RG &lt; 1/gm
CASE 2: RG &gt; 1/gm
• A follower is typically used to lower the driving impedance,
i.e. RG is large compared to 1/gm, so that the “active inductor”
characteristic on the right is usually observed.
EE105 Fall 2007
Lecture 21, Slide 23
Prof. Liu, UC Berkeley
Example
3  0
Z out
EE105 Fall 2007
j rO1 || rO 2 CGS 3  1

jCGS 3  g m 3
Lecture 21, Slide 24
Prof. Liu, UC Berkeley
MOS Cascode Stage
• For a cascode stage, Miller multiplication is smaller
than in the CS stage.
Av , XY
 1 
vX
  1

  g m1 
vY
 gm2 
 CX  2CXY
EE105 Fall 2007
Lecture 21, Slide 25
Prof. Liu, UC Berkeley
Cascode Stage: Pole Frequencies
 0
Cascode stage with MOSFET capacitances shown
(Miller approximation applied)
 p ,out 
 p ,Y 
 p, X 
1


g
RG CGS1  1  m1
gm2


EE105 Fall 2007


CGD1 


1
RD C DB 2  CGD 2 
1
1
gm2


gm2

C

C

1

 DB1
GS 2

g m1


Lecture 21, Slide 26


CGD1  C SB 2 


Prof. Liu, UC Berkeley
Cascode Stage: I/O Impedances
 0
Z in 
1



g m1 
CGD1 
j CGS1  1 
 gm2 


EE105 Fall 2007
Z out  RD ||
Lecture 21, Slide 27
1
j CGD 2  C DB 2 
Prof. Liu, UC Berkeley
```