PCIM Europe 2016, 10 – 12 May 2016, Nuremberg, Germany
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Melanie Adelmund, University of Bremen, Germany, melanie.adelmund@uni-bremen.de
Christian Bödeker, University of Bremen, Germany, christian.boedeker@uni-bremen.de
Nando Kaminski, University of Bremen, Germany, nando.kaminski@uni-bremen.de
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The development of fast power semiconductors increases the requirements for current
sensors. Due to the higher switching frequencies and the higher gradients of the current and
voltage transients, the capacitive and inductive coupling can affect the measurement signal
considerably. The requirements increase especially in the characterisation process, but also
for mass applications of fast semiconductors. In addition, some current sensor types introduce
an inductance into the load circuit, which affects the switching process and, thus, has to be
avoided. Therefore, the aim is to build a shunt with a minimum inductance and a high signal
fidelity simultaneously.
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Precise current measurements are required in most areas of power electronics. There is a
variety of sensor principles: Closed- and open-loop hall-effect sensors, Pearson current
transformers, Rogowski coils, and shunts. Due to the inherent isolation and the still adequate
bandwidth the Rogowski coil is widely used in power electronics. However, the maximum di/dt
is limited by the integrator circuit and the measurement signal is affected by capacitive coupling
into the windings. Recent publications show good results with a differential Rogowski coil, in
which the capacitive component of the signal can be eliminated [1]. The other standard
measurement devices are shunts, especially coaxial shunts, due to their very high achievable
bandwidth. However, also this device has its drawbacks: There is no galvanic isolation
between the shunt and the measurement instrument. Furthermore, the shunt has to be inserted
into the circuit to be measured. Thus, the inductance of the shunt adds to the circuit and can
affect the switching transient under investigation. In this work the focus is on optimising the
inductances of shunts to get a minimum influence on the switching process and to measure
without inductive coupling effects at the same time.
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The coaxial shunt (Fig. 1a and Fig. 2a) offers the advantage to place the measurement taps in
the field free region of the inner conductor [2] [3]. Due to the cylindrically symmetrical shape
the space in the inner conductor is not affected by magnetic fields caused by the current
through inner and outer conductor. The only influence on the measurement signal is caused
by the effective inductance of this circuit, i.e. the inner inductance of the inner conductor, which
is in the low picohenry range. Therefore, also high di/dt values will not affect the measurement
significantly. However, the construction of the shunt is demanding and the cooling of the actual
shunt material, i.e. the inner conductor is not easy to handle. To improve the heat dissipation
from the inner conductor, the space between inner and outer conductor needs to be filled with
ISBN 978-3-8007-4186-1
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© VDE VERLAG GMBH · Berlin · Offenbach
PCIM Europe 2016, 10 – 12 May 2016, Nuremberg, Germany
sense
sense
force
b)
sense
force
sense
force
a)
c)
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Fig. 1: Shunt structures: a) coaxial shunt b) hair-pin shunt c) Möbius shunt
Fig. 2: Self-made shunts: a) coaxial shunt b) hair-pin shunt c) Möbius
shunt. The additional BNC-connector is for the evaluation of the DCresistance
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a compound material. Resistance alloys like Manganin® offer the advantage of low temperature
coefficients and, therefore, low temperature drifts. However, also these alloyed conductors can
be damaged due to overheating. Thus, coaxial shunts can only be used with low continuous
power or only in pulse operation. Furthermore, the inductance introduced to the load circuit by
commercial coaxial shunts is at least in the range of 3 - 5 nH. Pipe or squirrel cage shunts are
“a kind of” coaxial shunt with a better continuous power rating but without the advantage of an
absolutely field free measurement in the inner conductor due to the openings in the walls to
avoid eddy currents. Also the inductance brought to the load circuit will be much higher due to
the bigger shape.
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© VDE VERLAG GMBH · Berlin · Offenbach
PCIM Europe 2016, 10 – 12 May 2016, Nuremberg, Germany
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With a planar shunt structure called hair-pin shunt [4] (Fig. 1b and Fig. 2b) very low
inductances can be achieved. Compared to the coaxial shunt the cooling is much easier due
to the accessibility of the resistance material. Test structures with the resistance part made of
Manganin® (Fig. 3) show inductances in the range of 1 nH or even lower in the load circuit.
Thus, the inductances of the constructed hair-pin shunts are anyway lower than those of
commercial coaxial shunts. However, the inductance also affects the measurement signal and,
thus, the measured voltage Vshunt contains a large inductive component VL (Fig. 5, blue line).
The inductive component can be compensated by an RC-network across the shunt (Fig. 4)
and its parasitic inductance [2] [5]. The resulting voltage VC and, therefore, the current signal
is measured across the capacitor CC (Fig. 5, red line). It is easy to see that the signal is well
compensated and shows similar curves like the best available reference, i.e. the coaxial shunt
by T&M Research Products, Inc.
I
IC
RC
Ishunt
Rshunt
Vshunt =
VR+VL
CC
VC
Lshunt
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Fig. 3: Hair-pin shunt and compensation
element
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Fig. 4: RC network for signal compensation
[2] [5]
60
hairpin-shunt uncomp.
hairpin-shunt comp.
coaxial shunt (T&M)
55
Current Ishunt, measured [A]
50
45
40
35
VL
30
25
20
15
10
5
VR
0
-5
5.08
5.1
5.12
5.14
Time t [μs]
5.16
5.18
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Fig. 5: Different shunt signals: reference signal of a coaxial shunt (green), uncompensated signal
of a hair-pin shunt (blue) and compensated signal of the hair-pin shunt (red)
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© VDE VERLAG GMBH · Berlin · Offenbach
PCIM Europe 2016, 10 – 12 May 2016, Nuremberg, Germany
To determine the values for the compensating RC-network [5] [6] the voltage drop Vshunt across
the uncompensated shunt can be expressed in the Laplace domain:
Vshunt (s)
Rshunt ˜ Ishunt (s) L shunt ˜ s ˜ Ishunt (s)
(Rshunt s ˜ L shunt ) ˜ Ishunt (s)
(1)
The current IC into the compensation branch consisting of the resistor RC and the capacitor CC
can be calculated by:
IC (s)
Vshunt (s)
R C 1/ s ˜ C C
Vshunt (s) ˜ s ˜ CC
s ˜ R C ˜ CC 1
(2)
Thus, the compensated measurement signal VC across the capacitor CC is given by:
VC (s)
IC (s) ˜
1
s ˜ CC
R shunt ˜ Ishunt (s) ˜
s ˜ L shunt / R shunt 1
s ˜ R C ˜ CC 1
(3)
With the matching condition of Lshunt / Rshunt = RC ∙ CC the voltage VC equals the voltage VR
across the resistor Rshunt.
VC (s) R shunt ˜ Ishunt (s)
(4)
Fig. 5 shows the compensated signal VC (red line), which is similar to the signal of the
commercial coaxial shunt (green line).
For the compensation network, the boundary condition RC >> |Rshunt + s∙Lshunt| [5] has to be
fulfilled to avoid a significant current, which would affect the measurements. Therefore, the
resistance RC is chosen to be about 1000 times higher than the shunt resistance Rshunt.
Recommended values for the resistor RC in the compensation network are in the range
30 Ω … 1000 Ω and for the capacitor CC within the range 47 pF … 1 nF [2]. Thus, the
compensation network (100 Ω / 220 pF) is in the recommended ranges. However, it has to be
taken into account, that elements of the compensation network (Fig. 4) also have additional
parasitic elements and, hence, are frequency depended. Also the resistance and capacitance
of the connected probe have to be considered (here: 10 MΩ / 8 pF).
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Another type of shunt is the Möbius shunt (Fig. 1c and Fig. 2c), which was patented in 1966
as a “non-inductive electrical resistor” [7]. Investigations showed that this shunt does not offer
a lower inductance or capacitance than planar shunt structures. The term “non-inductive”
seems to be related to bifilar wired shunts and not to planar constructions. In addition, the
measurement of the resulting voltage cannot be performed in a field free space like in the
coaxial shunt and inductive coupling will affect the measurement. Also the Möbius shunt needs
to be four times longer to get the same resistance like a hair-pin shunt due its parallel kind of
structure. Therefore, the inductance in the load circuit is for sure larger than that of a hair-pin
shunt of the same resistance. Thus, it does not offer any advantage over the hair-pin shunt.
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© VDE VERLAG GMBH · Berlin · Offenbach
PCIM Europe 2016, 10 – 12 May 2016, Nuremberg, Germany
sense
sense
2∙Rshunt
2∙Lshunt
2∙Lmeas
force
force
Fig. 6: Structure of the M-shunt
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Fig. 7: Prototypes of the M-shunt and the coaxial
shunt by T&M Research Products, Inc. used for
inner
comparison. The additional BNC-connectors were
inductance
G used for determination of the DC-resistances and
the inductances the shunts add to the load circuit.
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A novel shunt structure, which should combine the advantages of coaxial and hair-pin shunts
is a doubled hair-pin shunt. This planar M-shaped structure corresponds to the cross section
(Fig. 6) of the coaxial shunt. The taps for the measurement signal are in a field free space, like
in the case of the coaxial shunt, if both halves of the M-shaped structure are identical and the
connection to the measuring taps are between them. In this case, only the inner inductances
of the inner conductors will affect the measured signal. Furthermore, like the hair-pin shunt this
kind of shunt introduces only a low inductance between its force connectors and, therefore to
the load circuit.
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All transient measurements were performed
on a double pulse test bench, which is
configured as a buck converter. A printed
circuit board (PCB) was designed specifically
for the tests (Fig. 8). As switching device, a
SiC-MOSFET (Cree C2M0160120D) was
used. The load inductance was a handmade
spider web coil (LL = 500 μH, Cpar = 4.1 pF),
which freewheels over a SiC-Schottky-Diode
(Cree C4D05120E). An oscilloscope was
used to capture the measurement data
(Tektronix MSO4104). The test current was
Itest = 14.5 A and the voltage of the DC-link
was Vtest = 150 V.
The investigated shunts are the coaxial shunt
by T&M, a hair-pin shunt, and three different
M-shunts (all Fig. 7 except the hair-pin
shunt). The signal of the coaxial shunt was
used as reference current iref for the
calculation of the inductances of each shunt.
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Fig. 8: PCB used during the double pulse tests
for the investigation the parasitic inductances of
the shunts (LPCB = 31.8 nH)
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© VDE VERLAG GMBH · Berlin · Offenbach
PCIM Europe 2016, 10 – 12 May 2016, Nuremberg, Germany
All shunts were measured with the PCB (LPCB = 31.8 nH) shown in Fig. 8. The inductance, which
each shunt adds to the load circuit was determined by eq. 5. An assumption for the calculation
of the inductances from the shunt voltages Vshunt is that the inductances of the shunts are all
similar and quite low compared to the stray inductance of the PCB (Lshunt << LPCB) and,
therefore, will not affect the switching transients by themselves. This method was used for the
shunts #1 - #3. For even better precision, the coaxial shunt #1 and the M-shunts #4 and #5
were connected in series to guarantee an identical di/dt during the measurements. The
investigated values are shown in Tab. 1.
L shunt
Vshunt R shunt ˜ iref
diref / dt
(5)
The hair-pin shunt (shunt #2) adds only one third of the inductance of the coaxial shunt to the
load circuit. For the M-shunt, the evolution can be pursued by the results of the shunts #3 - #5.
The resistance value of shunt #3 is 30.7 mΩ and the stray inductance effective in the load
circuit is about 5.2 nH. The part of the inductance affecting the measurement (inner inductance
of the inner conductor) is about 177 pH higher than that of the coaxial shunt, which is the
reference. From these initial results further improvements have been derived: By using a
resistance alloy instead of brass and optimising the shunt structure the inductance in the load
circuit has decreased due to the reduced length of the resistance material and less space
between the layers of the shunt, respectively. Also the influence on the measurement signal
has decreased due to these improvements. By using Manganin© as resistance alloy (shunt #4)
the inductance, which adds to the load circuit is only 2.5 nH. Also the additional inductance
affecting the measurement signal is much lower with only 29 pH, whereby the measurement
signal is obviously improved. Shunt #3 and #4 came in the same housing, which was designed
for brass as resistance material. A further M-shunt (shunt #5) was again redesigned and, thus,
has much smaller structures specifically for Manganin©. Additionally, a thinner isolation of only
75 μm Mylar© was used. Therefore, this shunt only adds an inductance of 0.4 nH to the load
circuit. Even if the resistance would be four times higher and, thus, the shunt four times longer,
this shunt will have a lower inductance than the 3.3 nH of the coaxial shunt (shunt #1). Also
the inductance affecting the measurement signal is only 39 pH higher than that of the
reference.
Fig. 9 shows the actual measurements of the M-shunts during turn-off of the transistor. The
coaxial shunt is measured for comparison. All shunts were connected in series. Thus the di/dt
Shunt
#1
#2
Type
Coaxialshunt
Hair-pin
shunt
#3
M-shunt
#4
M-shunt
#5
M-shunt
Resistance
alloy
Isolation
material
Resistance
Rshunt
Inductance
Lshunt
Additional
inductance
Lmeas related
to reference
-
-
102.4 mΩ
3.3 nH
Reference
Manganin©
(17.5 μm)
Brass
(10 μm)
Manganin©
(17.5 μm)
Manganin©
(17.5 μm)
Mylar©
(125μm)
Mylar©
(125μm)
Mylar©
(125μm)
Mylar©
(75μm)
83.4 mΩ
1 nH
---1)
30.7 mΩ
5.2 nH
177 pH
21.4 mΩ
2.5 nH
29 pH
22.4 mΩ
0.4 nH
39 pH
Tab. 1: Results of the measured shunts
1)
same inductance as Lshunt in the uncompensated case
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© VDE VERLAG GMBH · Berlin · Offenbach
PCIM Europe 2016, 10 – 12 May 2016, Nuremberg, Germany
30
Coaxial shunt (#1)
M-shunt (#4)
M-shunt (#5)
Current Ishunt, measured [A]
25
20
15
10
5
0
-5
0
10
20
30
40
50
60
70
80
90 100
Time t [ns]
Fig. 9: Measurements of the M-shunts compared to the coaxial shunt
at all shunts is identical and the only influence on the measurement signal can appear due to
the inner inductance of the inner conductors, which affects the measurement signal. The higher
inductance Lmeas of the M-shunt #5 compared to the coaxial shunt and M-shunt #4 may be due
to deviations from the intended arrangement of the resistance alloy in the inner structure of the
shunt. Further improvement is on its way.
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The coaxial shunt is a high bandwidth shunt with measurement taps in the field free region of
the inner conductor but commercially available types add an inductance of 3 - 5 nH to the load
circuit. A hair-pin shunt offers a lower inductance compared to the coaxial shunt and is easy
to cool. However, the measurement taps are not placed in the field free space introducing a
large inductive component into the measured voltage. The proposed M-shunt combines the
advantages of the coaxial shunt and the planar hair-pin shunt. The inner inductance of the
inner conductors, which affects the measurement signal is quite low but not yet lower than that
of the coaxial shunt. All signals measured with Manganin© M-shunts differ only slightly from
the signal of the coaxial shunt and, therefore, are already sufficient for many applications.
Probably, the inductance, which affects the measurement signal can be further minimised by
a more precise arrangement of the inner structure. The inductance brought into the load circuit
by the best M-shunt is much lower than the inductance a commercial coaxial shunt adds to the
circuit. Another benefit of the M-shaped structure is the potentially better transfer of heat from
the resistive alloy to the ambient compared to the considered coaxial shunts.
A further advantage of the M-shunt is the simple construction with only plane parallel surfaces.
Therefore, the shunt is much easier and also cheaper to assemble than a comparable coaxial
shunt.
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© VDE VERLAG GMBH · Berlin · Offenbach
PCIM Europe 2016, 10 – 12 May 2016, Nuremberg, Germany
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The authors would like to thank Isabellenhütte Heusler GmbH & Co. KG for providing samples
of their Manganin® foil.
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[1] S. Hain and M.-M. Bakran, “New Rogowski coil design with a high DV/DT immunity and
high bandwidth,” presented at the 15th European Conference on Power Electronics and
Applications (EPE), 2013.
[2] S. A. Dyer, Wiley Survey of Instrumentation and Measurement. John Wiley & Sons, 2004.
[3] A. J. Schwab, “Low-Resistance Shunts for Impulse Currents,” IEEE Trans. Power Appar.
Syst., vol. PAS-90, no. 5, pp. 2251–2257, Sep. 1971.
[4] R. Davis, “Design Formulas for Nonreactive High-Voltage Pulse Resistors,” IEEE Trans.
Parts Mater. Packag., vol. 1, no. 2, pp. 3–23, Sep. 1965.
[5] D. Schröder, Leistungselektronische Bauelemente, 2nd ed. Springer Berlin Heidelberg
New York: Springer-Verlag, 2006.
[6] B. Hudoffsky, “Berührungslose Messung schnell veränderlicher Ströme,” Dissertation,
University of Stuttgart, Stuttgart, 2014.
[7] R. L. Davis, “Non-Inductive Electrical Resistor,” Patent, US 3,267,405, Aug. 1966.
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© VDE VERLAG GMBH · Berlin · Offenbach