PRODUCT APPLICATIONS
Using Current Sense PROFETs and
Speed TEMPFETs in a
High Current H-Bridge Motor Driver
A high power bridge driver can be constructed from Smart SIPMOS high side and
low side switches. Design issues are
addressed, including the use of simulation
models and auxiliary protection circuits.
by Jon Hancock, Infineon Technologies
Description/Overview of Bridge Drivers
TRILITHIC Smart Power full bridge drivers
have been introduced in a single package using
a combination of PROFET high side switches
with a low side MOSFET switching transistors.
This synergistic combination relies on the PROFET high side switch to implement motor direction control and a variety of protection
functions, while including two fast low cost
MOSFET transistors in the same package to
provide PWM control of the motor speed. With
the TRILITH IC bridge drivers it is possible to
implement H-Bridge drivers for specialized
functions, such as automotive door locks, as
well as more general purpose low power motor
control, using sign-magnitude modulation (Fig.
1). Available TRILITHIC H bridges range in continuous operating current range from 3A for PDSO-28 packaged parts like the BTS770, up to
15A in the case of the BTS780. However, there
are applications which require lower R DS[on]
than the BTS780, for reasons of maximum output current, efficiency or thermal performance.
For these a different solution is required.
Concept for High Power Bridge
Bridge driver for 12-24 volt applications using
discrete Smart SIPMOS transistors instead of
a package integrated bridge driver. This
approach supports optimizing the choice of the
high side and low side switch for very low overall conduction losses. Capabilities of these
transistors as regards switching speed, control
methods, and protection functions will be
explored and discussed, with some recommendations for options using external circuitry
to optimize the protection functions for specific
application requirements. This will include a
brief overview of new PSPICE simulation models for these transistors, with examples of circuit analysis of auxiliary protective circuits.
Current Sense PROFET
The high side switch used will be one of
the family of Current Sense PROFET's. These
are very low RDS[on] high side switches with a
current sense output from the power transistor.
This output pin provides a current output at a
ratio to the main transistor current, approximately 1/5000 for the BTS650P, which can be
monitored by a system controller, or used to
trigger protection circuits locally. In the case of
the bridge drivers, the PROFET is chosen to
have a higher conduction resistance and
higher thermal impedance, so that the maximum thermal stress is always on the PROFET,
which has very comprehensive protection circuits built in. Current Sense PROFET’s have
very low RDS[on] and low thermal impedance,
so instead of using a more rugged low side
switch, it is preferable at this power and current level to use a low side switch which also
has at least some basic protection functions.
This Product Application Note will describe
a concept for a higher power, high efficiency H1 of 13
PRODUCT APPLICATIONS
Introducing Speed TEMPFETs
TEMPFET's are thermally protected
Smart SIPMOS transistors, introduced by Siemens/Infineon over 10 years ago. A small sensor chip is mounted on top of the base power
chip, and connections made from the sensor
chip to the gate and source pins of the MOSFET base chip. The sensor chip functions as if
it was a thermally triggered SCR; when the
sensor chip temperature rises above 150°C,
the SCR fires, clamping off the gate drive to
the MOSFET and turning off the transistor.
Because the sensor chip must be made very
small to have fast thermal reaction time, the
clamp off capability of the SCR like structure is
low, so the gate input current must be limited,
typically by an input resistor, so that the gate
threshold voltage isn't exceeded in the triggered mode. This limitation on the gate input
current also limits the switching speed, typically to the range of 10 kHz or less.
Chip-on-chip construction of Speed
TEMPFET
FIGURE 1:
FIGURE 2:
Typical H-Bridge Configuration for Motor
drive with DMOS transistors
The Speed TEMPFET is a new family of
TEMPFET transistors which have additional
pin connections so that the temperature sensor connection is available separately, and can
be connected elsewhere based on the
designer's preference. In this way, t h e
designer has the choice of connecting the sensor pins as previously down with the original
series of TEMPFETs directly to the gate and
source pin, or of connecting them to some
PWM Modulation for Sign Magnitude
Operation of Bridge
2 of 13 Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver
PRODUCT APPLICATIONS
other circuitry used to disable gate drive. In
this way, the limitation of gate drive current can
be overcome, and the SPEED TEMPFET may
be used for higher frequency switching applications.
H-Bridge Modulation Method
Because of the new connection method
of the protection sensor in SPEED
TEMPFETs, these transistors can be used for
switching frequencies from a few hundred
hertz up to 100's of kilo-hertz. This gives a fast
modulation capability for the low side of the
bridge. However, PROFETs, which feature an
internal charge pump and driver circuit, are
designed for controlled, low EMI switching,
and cannot be switched practically above a
few hundred Hz. This imposes some limitations on the motor control PWM modulation
scheme which may be used. The combination
of High side PROFET and low side SPEED
TEMPFET is suited to Sign-Magnitude control
(Fig. 2), where the high side switch is used to
control direction, and also may be used for
braking, but is not run at the PWM clock frequency. In this mode, Synchronous Sign-Magnitude control or Locked-Antiphase control are
not practical, because they require fast switching of the high side transistor as well as the
low side transistor.
"F ORWARD"
"BRAKING"
When using Sign-Magnitude control with
PROFETs and TEMPFETs, the high side
switch is used principally for direction steering,
and may also be used for braking. Referring to
Fig. 1 and 3, if the switch MH1 is turned on,
then a PWM signal is applied to the diagonal
transistor ML2 to control the driving torque of
the motor. Braking may be accomplished by
turning on both high side switches, while
reversing direction entails turning on the other
high side switch MH2, and applying the PWM
torque control to ML1. Changing modes
between different directions and braking must
account for the switching time of the PROFET
high side switch, so setup delays of 350 - 500
µsec should be planned, as shown in Figure 3.
There are some additional issues regarding switching behavior which have to be taken
into account for a bridge driver using PROFET's as the upper transistor. Because of a
technology limitation in the driver circuit for
PROFETs, the impedance clamping off the
internal gate of the power MOSFET is determined by the value of the polysilicon resistor
used to limit turn-on speed. When the output
voltage of one leg of the bridge (see Fig. 1) is
high, the gate to drain voltage on that PROFET is low, and the internal gate to drain
capacitance is high, as is the case for all
DMOS transistors. When the transistor in the
"REVERSE"
"BRAKE TO STOP "
OFF
On
MH1
Off
On
MH2
Off
Braking delay Low-side
On
ML1
Off
On
ML2
Setup delay Low-side
Brake to reverse actuate delay
Braking delay Low-side
Off
FIGURE 3:
Sign-Magnitude modulation scheme for four phases of H-Bridge used as motor drive
Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver3 of 13
PRODUCT APPLICATIONS
lower leg of the bridge turns on, pulling the
source low, the drain to gate capacitance
forms a divider network with the source input
capacitance and the gate resistor. At higher
dv/dt's, it is possible for enough charge to be
injected from the drain to gate to enhance the
gate beyond the threshold voltage for conduction, turning on power base chip of the PROFET. This will cause a brief current spike
through the drain to source pins. As the rate of
dv/dt increases, the enhancement voltage
generated at the gate internally will increase,
raising both the level and duration of current
pulse through the PROFET that is nominally in
an “off” mode. This affect shows up even in the
simulation model of the BTS650P, and can be
seen in the simulation output in Fig. 14.
SIPMOS Simulation Models
“What-If” evaluation of potential circuit
configurations and components through simulation can be a useful part of design process
only if device models with good accuracy and
the required characteristics are available to
the designer. To this end, Infineon has spent
considerable effort developing component
models for both discrete SIPMOS transistors
and Smart SIPMOS transistors which are
strongly based on device physics, and incorporate a number of characteristics necessary
for power circuit simulation and evaluation.
The “core” SIPMOS power transistor
model which is now available is an electrothermal macro model based closely on mathematical models of the device physics. Simulators such as SPICE, PSPICE, etc. do have
semiconductor models built in, but these are
single temperature models based on the structures and processes used in small signal integrated circuit components. There are many
and substantial differences in structure and
electrical characteristics between the lateral
MOSFET transistors used in typical IC applications, and the vertical DMOS power transistor
structures widely used in MOSFET power transistors. Because of this, modeling power
MOSFET transistor requires a different
approach, combining a number of semiconductor model elements with analog behavioral
elements with auxiliary equations, in order to
create a model macro which behaves as the
physical device.
A built-in SPICE Level 3 model is used
for the core of the device model, but this is
augmented with additional semiconductor and
Miller capacitance C dg
I dg =
[
d
C ox ⋅ (V dig − V depl )
dt
with

V
V depl = V dig + 2 V n  1 − 1 + dig

Vn

]
2

 k dg 



V
=
 and n  C 
 ox 

Drain Resistance
V Rd
 Tj
= R do ⋅ I d ⋅ 
 300



3
2
Quasi-Saturation
V Qsat = α ⋅ V dX2 1
FIGURE 4:
Physics based SIPMOS Macro Model for SPICE and SABER
4 of 13 Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver
PRODUCT APPLICATIONS
35
30
Id [A]
25
Vds [V]
20
15
2x Vgs [V]
10
5
0
10
FIGURE 4:
20
time [usec]
40
50
Physics based Capacitance Model with dynamic ID as a function of VGS results in
accurate switching behavior
passive components, plus a several mathematical function blocks which implement specific functional areas of power MOSFET. The
basic block diagram for the fixed temperature
version of this model, with some of the key
analog behavioral functions, is shown in Figure 4. Special function blocks are used to calculate the non-linear gate to drain
capacitance, the temperature dependent drain
resistance, and the quasi-saturation behavior
of the drain. The operation and derivation of
these functions has been described previously
[1], and won’t be covered in detail here.
The net result of this development effort
was a model structure which provides a good
description of the static and dynamic behavior
of the power MOSFET in it’s primary operating
modes, within the normal device SOA. This is
shown in Figure 4, where comparison is made
between measured switch response and the
simulated switching behavior using the Level 1
model for a 55V BUZ103S SFET transistor.
However, the requirements for modeling
many power conversion applications go
beyond this capability, and require the development of an electro-thermal power MOSFET
model. The concept for this type of model is
built upon the foundation of the device model
just described, but also includes elements to
calculate the instantaneous power dissipation
Rd (TJ )
Pv (t ) =
Vds ⋅ I ds
G
Vth (TJ)
Tj
Rth1
Cth1
Rth6
Cth2
Tc
External Heatsink
Cth6
Tamb
Gfs(TJ)
S
FIGURE 5:
Electrothermal SIPMOS model using physically based thermal network, with key
parameters instantaneously dependent on Tj
Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver5 of 13
PRODUCT APPLICATIONS
700
measurements
600
Short circuit
simulation
Id [A]
500
400
Level-3 model
300
200
10x Vds
100
0
-2
FIGURE 6:
0
2
4
6
Time [usec] 14
16
18
Measured and simulated short circuit behavior for a BUZ100SL transistor in a low
impedance short circuit test fixture
in the transistor, elements representing the
transient thermal impedance of the transistor,
and a feedback system to modify key device
parameters affecting the transfer function
which are dependent on the junction temperature (Fig. 5). These parameters include the
drain resistance RD=f(TJ), the gate threshold
voltage Vth =f(TJ), and the transconductance
Gfs=f(TJ). While there are other device parameters affected by operating temperature, these
are the key parameters affecting the conducting transfer function within the normal device
SOA. Accounting for this aspect of the MOSFET’s characteristics makes it possible to predict the circuit behavior even in many overload
conditions, as long as the junction operating
temperature is within the range where parasitic
bipolar currents don’t have an appreciable
impact on the transistor behavior (typically less
than 250°C).
Figure 6 illustrates this capability comparing the measured and simulated behavior for a
BUZ100SL 55V transistor in a low impedance
short circuit test fixture. Note the close tracking
of drain current as a function of time between
the measurement and the simulation. This is
only possible if both the static transfer function
based on temperature and the transient thermal impedance are accurately modeled.
Bridge Design Example for 12V Application
Next, some design hints for a specific
application will be reviewed, including issues
which must be observed for all applications,
and optional circuits for additional levels of
functionality or protection. This design example will address a motor drive for a small DC
motor having a 5-6A current unloaded, and up
to 12A current loaded, and a nominal stall current of 30 - 36A. Selection of the most appropriate devices for this application depends on
a variety of criteria which vary from application
to application. These include the maximum
ambient temperature for the electronics module, the available thermal impedance to ambient, and the assembly and packaging goals for
the electronics.
A general trend in power electronics in
recent years as the available RDS[on] in low
voltage transistors plummets is the movement
away from generous heatsinking, and a reliance on low conduction losses to improve the
thermal behavior of power modules. For this
reason it is more common to select the transistors based on RDS[on] and thermal issues,
rather than conventional MOS current ratings,
though the ISO rating, based on a case temperature of 85°C instead of the MOS rating of
25°C, may still be useful. While a nominal 100-
6 of 13 Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver
PRODUCT APPLICATIONS
TJ max – TA max
150° – 85 °- = 4.0625W
----------------------------------- = ------------------------( 16 ° ) ⁄ W
Rθ J – A
Measured & Fitted Motor Transfer Function, IRMS = f(VRMS)
y=a(1-exp(-bx))+c(1-exp(-dx))
7
6
5
IRMS
125 watt transistor may have the current rating
to handle the described load with margin, if the
on-state resistance at maximum junction
results in 25 watts of power dissipation, the
resulting module will neither be small nor run
particularly cool. Working backwards instead
from thermal impedance and maximum ambient and junction temperatures, let’s take as a
given that the best Rth(J-A) that the proposed
packaging concept can offer is 16°C/W for
each TO-220 packaged transistor in the motor
bridge driver. With a maximum ambient of
85°C and a maximum operating junction temperature of 150°C, the allowable power dissipation per package is found from:
4
3
2
1
0
0
5
10
15
VRMS
Figure 7:
Nominal transfer function of example 12V
motor, NL conditions
(EQ 1)
This shows that for these conditions, we want
to limit maximum power dissipation per package to 4 watts. In the case of the high side
switch, switching only occurs when turning on
and off, and changing motor direction or braking. Switching losses therefore are negligible,
but in addition to MOSFET conduction losses
in the forward mode, conduction losses in the
body diode in the freewheeling mode will need
to be considered. These losses, unlike the
MOS conduction losses, are not easy to calculate because the current level is proportional to
the on time of the PWM switch, but freewheeling time is inversely proportional to the on time
of the PWM switch, and the conduction loss is
a function of these two plus the non-linear current dependent diode voltage drop. Because
of the switching speed limitations of PROFETs,
it is not feasible to operate in sign-magnitude
synchronous mode, a mode for which the free
wheeling current would be handled by the
MOS channel, not the body diode.
What are the worst case conditions for
free-wheeling current losses in the body
diode? It turns out that the answer is not obvious from looking at the rms voltage and current characteristic of the motor (Figure 7),
because the motor is a very non-linear load.
Figures 8 and 9 show switching waveforms
Vbb
VOut
IOut
Figure 8:
Voltage for switching leg and motor
current, Vbb, at 10% duty cycle
Vbb
VOut
IOut
Figure 9:
Voltage for switching leg and motor
current, Vbb, at 20% duty cycle
Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver7 of 13
PRODUCT APPLICATIONS
Tj MAX – Ta MAX
150 – 85
R θ ( J – A ) = ------------------------------------ = --------------------- = 32.5 ( ° ⁄ w )
2
PD
Vbb
(EQ 3)
This is the starting point for the maximum thermal impedance junction to ambient for the high
side switch. In this example, staying with a TO220 packaged device, we’ll use a BTS650P.
IOut
Vout
Figure 10:
Voltage for switching leg and motor
current, Vbb, at 50% duty cycle
and the motor current for a light drive (10%
duty cycle, Figure 8) and at 20% (Figure 9)
and 50% (Figure 10). When near the stall condition at 10% duty cycle, back emf from the
generator effect is at the minimum, and the
freewheeling current time is relatively high,
with the characteristic inductive reset. At 20%
there is some reduction of the reset time compared to the charging time, as some generator
effects come into play. As the driver duty cycle
increases beyond 25%, approaching 50%, the
motor is running at much higher speed, and
generator effects cause the inductive reset to
be terminated very quickly, in this case in less
than 100 µsec. Clearly, the condition near stall
is the worst case in this example. From the
observed behavior, the approximate loss can
be calculated from:
I
30
P D = ----P × V F × D = ------ × 0.9 × 0.1 = 1.35W
2
2
A starting point for looking at the low side
transistor switching and thermal issues must
consider the off state dv/dt limitation of the
PROFET. As previously mentioned, to avoid
turn-on of the highside switch, the maximum
dv/dt in switching must be limited to a rate
which avoids injecting charge from the drain to
gate capacitance so rapidly into the gate to
source capacitance that the internal gate
threshold voltage of the PROFET is exceeded.
This maximum rate of dv/dt is roughly 0.7 volts
per microsecond. This places an upper limit on
turn-on speed which needs to be taken into
account when estimating switching losses.
However, this only affects the turn-on drive,
and is likely to have a beneficial side effect, in
that the body diode of the PROFET will not be
subjected to high peak Irrm. Turn-off switching
can be set without a specific dv/dt limit. Fortunately, the motor inductance limits the rate of
Vbb
(EQ 2)
This suggests that the minimum power dissipation to expect in this example in the high
side switch should be 1.5 to 2 watts, depending entirely on the reverse diode current. Using
Equation 1, and solving for R θ ( J – A ) ,
VOut
IOut
Figure 11:
Turn-on and turn-off for normal low duty
cycle stalled rotor condition.
8 of 13 Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver
PRODUCT APPLICATIONS
current rise under normal stall conditions at
low duty cycles, so the turn-on condition is relatively benign, in spite of peak current almost
five times the normal run current, as is seen in
Figure 11. Even at true locked rotor conditions
and 80% duty cycle, voltage reset occurs just
before the onset of conduction, and driver
power dissipation is not excessive (Figure 12).
Locked rotor impedance of the example motor
is 60 µH, and about 0.44 ohms.
Vbb
VDrvr
IOut
VOut
In the example circuit of Figure 13, drain
to gate feedback capacitors are used with a
controlled gate input current to limit the maximum dv/dt at the output nodes of the H-Bridge.
The external capacitors are used to help “linearize” the switching behavior of the low side
switch. Vertical DMOS power MOSFET’s
inherently have a large variation of drain to
gate capacitance with varying V DS ; it can
Figure 12:
Locked Rotor Driver, Half-bridge output,
and current at D = 80%
range over a 20 to 1 ratio or more. Though the
ideal driver would be a current source, a resistive mode driver is practical because drain
4
OUT
OUT
3
IS
IN
1
2
2
6
6
7
7
OUT
OUT
OUT
IN
OUT
3
U2
BTS650P
5
U1
BTS650P
5
OUT
1
VBB
OUT
IS
VBB
4
VBATT
R14
D14
D16
1N4148
1N4148
MH2
4.7K
R16
4.7K
Q12
2N3904
Q10
2N3904
Q11
R1
1N5228
10K
R2
D13
47n
R17
R19
3.9K
100K
1N4148
D17
1N5231B
D18
C6
47n
R20
3.9K
100K
C3
1N4148
R5
10K
1N4148
D1
1N4148
D3
R7
1N4148
10K
1N5227
R6
D7
R3
Q3
2N2222A
1N5226
Q1
BTS244Z
D10
1N5227
A
K
A
D9
D11
1N5242
K
2N2222A
Q4
2N2222A
Q2
Figure 13:
R10
4.7K
D4
D2
4.7K
10K
1N4750
1N4750
R9
R11 1K
+12V
2.7nF
D6
D5
ML1
D15
1N4148
C4
2.7nF
+12V
2N3904
2N3904
4.7K
R18
1N5231B
Q8
R15
J?
BP-DUAL
1
C5
4.7K
D20
1N5228
10K
R13
Q7
2N3904
2
Q9
2N3904
D19
Q6
C1
1n
1K
BTS244Z
D12
1N5242
R4
1.0k
1.0k
R12
D8
1N5226
C2
1n
Q5
R8
10K
2N2222A
Application Circuit, with optional overcurrent/short circuit protection areas
shaded
Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver9 of 13
ML2
PRODUCT APPLICATIONS
voltage switching occurs during the gate plateau region when the gate voltage remains relatively flat, and the gate charge drives the
integrator function of the gate to drain capacitance. During this time the gate voltage of the
driving transistor changes relatively little.
Driver charging current is defined by the effective applied driving voltage divided by the
series resistance. The effective driving voltage
is the applied drive voltage minus the gate plateau voltage for the actual load current. As an
example, if the driver voltage available is 10
volts, and an inspection of the transistor’s
transfer function in the data sheet shows the
effective plateau voltage for the normal load
current will be about 3.5 volts, and the initial
value for the gate resistor is 2K, then the driver
current is found from
V Drvr – V Plat
12 – 3.5- = 3.25mA
IG Drvr = -----------------------------= -----------------2000
RG
(EQ 4)
For a 14V nominal switching voltage, the total
switching interval at a maximum dv/dt is 20
µsec. The capacitor value desired gate to
drain is calculated based on the total gate
driver charge over the switching interval, so
IG Drvr ( T Switch )
3.25mA ( 20 µ s )
C Drain = ------------------------------------ = ------------------------------------ = 4.64nF
V Drain
14
(EQ 5)
The value of Crss at the rated VDS for the
low side transistor should be subtracted from
the value for CDrain, to select the value for the
external resistor. For this example, using the
BTS244Z, this value is approximately 500 pF,
leaving the nearest standard value for the
external capacitor at 3.9 nF. With the additional paralleled gate to drain capacitance and
the specified driver, the maximum dv/dt will be
limited to a safe value. If this limiting doesn’t
occur during the initial voltage fall of the HBridge output node, then the charge transfer
may turn-on the BTS650P, and shoot through
current will occur on that side of the bridge.
This phenomena is illustrated also by the simulation results in Figure 14, because the
device model is just as susceptible to this phe-
Date/Time run: 08/02/99 14:46:09
Temperature: 27.0
(A) TEMPFET Test LS Switch H-Bridge.dat
80
60
“Shoot-through” Current from excess dv/dt
on BTS650P turning on internal MOSFET
40
20
-0
50usV(VGate)60usV(Q1:C)
V(V3:+)
V(Isense650)
80us
V(VDrain)100us
V(X2:TJ)
-I(R5)
120us
140us
Time
Figure 14:
Simulation with “Fast” driver showing affect of excessive dv/dt during critical initial fall
interval and internal turn-on of BTS650P
10 of 13 Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver
150us
PRODUCT APPLICATIONS
nomena as the actual transistor. The critical
interval is in the initial fall time, when Crss is at
a minimum. In the simulation example shown,
though most of the switching interval is this
case is below the limit, the critical initial dv/dt is
too high.
Optimizing the low side switch
Other factors to consider for the low side
switch are the total expected losses, thermal
performance with the intended mounting, and
optionally, optimization for short circuit current
limiting vs. RDS[on]. With the typical range of
PWM switching frequencies for motor drives
between a few hundred Hz up to 20 kHz,
switching losses for Speed TEMPFETs will be
low compared with the conduction losses. The
latter are variable, depending on the current
profile, which in turn is dependent on PWM
setting, motor speed, and required torque.
Device selection for SMD mounting is
driven by RDS[on] and total power dissipation,
rather than by conventional device current ratings, because the standard current ratings are
based upon low external thermal impedance
with high junction temperature, and represent
the upper bound for the transistor’s capability.
The preferred trend is to minimize the total
power loss in an SMD environment when possible, or when using minimal external heatsinking. In either case, keeping power dissipation
in the range of 2-3W, and Rth(J-A) in the range
of 15 - 30 K/W will provide a compact power
module with minimal heatsinking requirements. Using the same conditions defined earlier for the high side switch, an Rth(J-A) of 16
K/W is assumed, which gives a maximum
power dissipation “budget” of 4 watts.
For this example we assume a worst
case rms current of 10A under load. Speed
TEMPFETs are logic level devices, but they
further reduce the RDS[on] with increasing gate
drive. At 4.5 V of gate drive, the BTS244Z has
a maximum RDS[on] at Tj = 25°C of 18 milliohms, but with 10V of gate drive this reduces
to 13 milli-ohms. Over the full operating range,
this RDS[on] will double at maximum junction
temperature. With 10V of gate drive, the contribution of conduction losses to power dissipation at 10A rms with worst case TJ will be
2.6 watts, requiring (from Equation 3 above)
an Rth(J-A) of 25 K/W if the maximum ambient
is 85°C. This fits within the goals above, making the BTS244Z a good choice.
Protection of the Low Side Switch
A remaining consideration and important
one is optimizing the protection for the Speed
TEMPFET. Because of the mass of the thermal sensor chip, there is a finite response time
for the sensor chip to match the base chip temperature. Under short circuit conditions with
high drain to source voltage and high gate
drive, the short circuit current may be so high
and the power dissipation so great that junction temperature can exceed a safe operating
value before the sensor can respond and trigger, resulting in degradation or failure of the
base power chip. This consideration was
addressed in the short circuit safe operating
area graph in the TEMPFET data sheet, which
shows the maximum gate drive voltage which
can be used, while limiting the short circuit current to a safe value, depending on the maximum voltage applied drain to source. The
drawback to this is that this usually results in a
relatively low gate drive voltage, which raises
RDS[on] and increases power dissipation.
The shaded area in the low side switch
drivers of Fig. 13 shows one method of dealing
with this problem. This is a driver circuit with
short circuit and current limit protection functions, which under normal load conditions
allows the full 10 volts of gate drive to be used,
optimizing the R DS[on] . However, when the
driver is turned on, and should the drain circuit
voltage rise due to high current flowing drain to
source, the NPN protection transistors Q3 or
Q5 will turn on and clamp the gate voltage to a
lower level. For mild overloads, this results in a
relatively slow, gentle response, as seen in
Figure 15. With hard short circuits, the
response is much faster, and prevents the
Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver11 of 13
PRODUCT APPLICATIONS
possibility of exceeding the short circuit SOA
of the Speed TEMPFET. Figure 16 shows a
simulation schematic for this circuit with the
BTS244Z. The additional R/C network connected to the lower right-hand part of the
BTS244Z is the equivalent circuit model for
SMD mounting with 6 cm2 of copper foil. Figure 17 shows the response to a “hard” short
circuit in what is called Short Circuit Mode 2.
This is a short circuit test condition where the
short is applied after the transistor is turned on
and in full conduction, with minimum drain to
source voltage, and maximum gate to source
voltage. This is the highest possible stress
mode for short circuit, and produces the highest peak short circuit current. This is the case
because the gate to drain capacitance is at it’s
highest, and when the transistor starts to desaturate from the sudden short, the rise in
drain voltage is transferred to the gate, and
increases the gate enhancement voltage
unless the gate driver is very low impedance.
VDrvr
Vbb
VOut
IOut
Figure 15:
V
tClose=400us
1
U2
L2
2
0.8mH
Low side protection circuit response to
locked rotor condition
R_Short
R7
0.01
1.4
R8
0.004
L1
120NH
V
C6
2.7n
VDrain
+
V1
12.0V
D6
R3
4.7K
-
D3
R6 10K 1N4148
0
V
GATE_DRIVE
2N2222A/ZTX
V
D12
D1N5227
D9
+
-
TJ
D4
D1N5242
R2
10k
2N2222A/ZTX
Q2
D10
D1N5226
R9
1MEG
0
sRth1
sRth2
sRth3
sRth4
A
Tamb
K
sCth1
1.218m
0.8268
sCth2
0.4863
4.266
sCth3
3.336
26.11283
7.371
sCth4
37.977
+
V
Sens_Clamp
R4 1k
V
TC
1N4148
ts
R1 1k
Q1
V3
D1N4750 X2
BTS244Z
C1
1n
R5
0.003
V2
+
Vamb
85
16.0
-
-
0
0
Figure 16:
Simulation Example for Short Circuit Mode 2, where the short is applied after the
power transistor is turned on and operating at normal curre30
12 of 13 Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver
PRODUCT APPLICATIONS
Summary
Low RDS[on] Smart SIPMOS transistors
make it practical to implement an H-Bridge
motor driver in SMD technology at relatively
high current with minimal heatsinking. By
using sign-magnitude modulation for variable
speed DC motor control, PROFET high side
switches may be used for the direction steering and braking control, and Speed
TEMPFETs for the low side PWM control.
PSPICE and SABER electro-thermal simulation models are available which enable
designers to test and evaluate the power circuits under a variety of conditions. Analysis
capabilities with these electro-thermal models
include junction temperature rise from thermal
effects due to the power dissipation from overloads and normal loads, as well as switching
and protective circuit behavior.
* E:\Msim_8\Projects\PROFET-TEMPFET Bridge\TEMPFET Test LS Switch W Short2.sch
Date/Time run: 08/03/99 16:10:03
Temperature: 27.0
(A) TEMPFET Test LS Switch W Short2.dat
200
Current Pulse during
Type 2 short circuit
150
100
50
0
150usV(R1:1)
V(X2:TJ)
200us V(C1:2)
V(R_Short:1)
250us
V(VDrain)300usV(GATE_DRIVE) 350us
-I(R5)
400us
450us
Time
Figure 17:
Simulation Example for Short Circuit Mode 2, where the short is applied after the power transistor is
turned on and operating at normal current; SC current pulse is limited to 10 µsec duration
Using Current Sense PROFETs and Speed TEMPFETs in a High Current H-Bridge Motor Driver13 of 13