1
ERA Report 99-0164
ELECTROMAGNETIC COMPATIBILITY DIVISION
Practical
implementation
of
radiated emission reduction and
frequency stabilisation of ISM
equipment operating in the 27
MHz band
P S Bansal, A Finney, M Philippakis, C Martel
FINAL REPORT
ERA Report 99-0164
ERA Project 33-01-1263
Client
: Radiocommunications Agency
Client Reference
: Mr P K Sunda
ERA Report Checked by:
Approved by:
Dr A Finney
Manager
Advanced Projects Department
A J Maddocks
Manager
EMC Division
February, 16
Ref. PSB/vs/33/1263/Rep5117
2
ERA Report 99-0164

Copyright 1999
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PB/vs/33/Rep5117
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ERA Report 99-0164
Executive Summary
This work performed by ERA Technology Ltd for the Radiocommunications Agency (RA) concerns
practical but systematic approaches in combating radio frequency interference (RFI) from industrial,
scientific and medical (ISM) equipment operating at 13 MHz and 27 MHz. Previous work reported in
ERA Report 98-0221 identified three factors which will cause RFI: field strength of radiated
emissions, frequency of operation and harmonic emissions. The work presented in this report deals
with the practical implementation of three techniques to reduce RFI,
(i)
polarisers to reduce radiated emissions from buildings
(ii)
vestibules to reduce harmonic emissions from conveyorised systems
(iii)
frequency stabilisation procedure applied to an old C-frame plastic welder operating at
50 MHz.
1.
Polarisers
A polariser is made up of conducting rods or wires which are parallel to one another at a constant
spacing. If an electromagnetic wave is incident on the wire grid the wave would be attenuated through
the polariser if the electric component of the wave is parallel to the wires but will be transmitted (with
very little loss) if the electric component is perpendicular to the wires. The polariser also acts as a
transmission grating dispersing the incident em wave energy over a wide angle (depending upon the
wire spacing). The theory of polarisers is provided which takes into account the conductivity of the
wire material, expressions for the reflection and transmission intensities are given. A computer
program was written which calculates the transmission attenuation in dB as a function of incident
angle and also frequency for an infinite polariser. The input parameters are
(i)
diameter of the wires
(ii)
wire spacings
(iii)
number of wires
(iv)
electrical resistivity of the wires
(v)
relative permeability of the wires
(vi)
the frequency of operation of the ISM machine.
Measurements on the transmission attenuation through a polariser were made at ERA on two sites.
The polariser was 3m high by 4m wide, with copper wires of diameter 1 mm spaced at 100 mm (and
then at 50 mm) parallel to the height. Results show good agreement between theory and
measurements between 80 to 300 MHz. At lower frequencies there was poor correlation due to the
edge effects of the polariser. On ERA site 1, which was just outside the covered open area test site
PB/vs/33/Rep5117
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ERA Report 99-0164
(OATS), the problem of interference due to reflected or scattered waves was not so great, and hence a
good agreement was obtained between theory and experimental results. Similar measurements were
made on ERA site 2, which was located within the confines of buildings, the effects of scattering from
walls marred the results, but overall, there was approximately 10 dB of attenuation through the
polariser. The main advantage of the polariser is that they can be designed and applied at any
frequency.
2.
Waveguide Vestibules
For conveyorised systems, waveguide techniques were applied to reduce harmonic emissions. Firstly,
theoretical models were simulated using finite-difference time-domain (FDTD) techniques on
waveguide structures which included the use of cut-off techniques, introduction of lossy materials,
metallic obstacles and diversion of em wave propagation to another port terminated in a lossy load.
Initial simulations were conducted on vestibule dimensions already in use at a site which produced
edible products. The vestibule was treated as a waveguide, that is, the vestibule is in good electrical
contact with the conveyor belt. If this cannot be achieved then the use of vestibules would be of no
use, since the vestibule plus the isolated conveyor belt would create a coaxial structure and hence the
generation of the transverse electromagnetic (TEM) mode would prevail (note: the TEM mode can
exist from DC to very high frequencies, and this would readily extract all harmonic energy and radiate
this from the output port to the environment. From a theoretical point, simulation results obtained with
the waveguide lined with absorbing material does not produce any appreciable amount of attenuation.
The emphasis was then on diverting the energy into another port, that is, with the aid of a T-piece. A
series of simulations were performed with the T-piece, with the inclusion of an angle diverting plate
which directed the em wave energy out of the guide into the T-piece which was terminated in a load.
A good agreement between the simulations and the final T-piece waveguide was obtained with
variations within + 5 dB over most part of the frequency range considered. The main advantage in
using a waveguide T-piece is that one can design vestibules with T-piece over any broad frequency
range which will offer ~ 20 dB of attenuation. Great interest has been shown by two leading dielectric
heater manufacturers in this technique. Recommendations provided for further work using waveguide
techniques are:
(a) to optimise the T-piece performance
(b) explore other waveguide techniques which may offer better attenuations over a broadband
(c) investigate the performance, use and placement of absorbing materials within the waveguide Tpiece
(d) implement the T-piece on existing ISM machines utilising conveyorised systems.
3.
Frequency Stabilisation
There are a large number of dielectric heaters operating around the 13 MHz and 27 MHz frequencies,
however the problem of frequency stability and operation remote from these frequencies are a major
PB/vs/33/Rep5117
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ERA Report 99-0164
concern when dealing with RFI cases. A frequency stabilisation procedure was written in the previous
work and so the main objectives of this part of the work was to:
(a) validate the procedure
(b) determine which parameters affect the operational frequency
(c) provide a training aid for RA staff in giving practical advice to ISM users in methods of
stabilising the frequency of operation to be within the ITU band.
The basic theory of the tank and applicator circuits is given along with the RF coupling mechanism
and the relevant equations. The frequency stabilisation procedure contains the following:
(a) equipment required
(b) system calibration
(c) measurement set-up
(d) measurement procedure.
The tank cavity is excited by a small single turn loop connected to the automatic network analyser
(ANA). The advantage of this technique is that all measurements are made at very low power (~0-10
dBm) and changes in various parameters could be made without damaging the machine or the
diagnostic equipment. The return loss measurements are continuously made whilst the following
machine parameters are adjusted:
(a) tank cavity volume
(b) applicator spacing
(c) coupling loop
(d) material parameters.
The frequency stabilisation procedure was applied to an old C-frame plastic welder. The parameters
varied were (a), (b) and (c) on the machine. From these variables, the best way to establish the
operation frequency was to vary the tank volume. In varying the applicator spacing over
approximately 10 cm, the frequency change was approximately 0.2 MHz. The problems encountered
when changing the power flap spacing were in obtaining good coupling and the possibility of
incurring electrical sparking and coronas. In reducing the tank volume by loading the tank with
conducting blocks, a 15.5% volume reduction produced a 3.5 MHz frequency rise in the tank
frequency. Recommendations provide for further work in frequency stabilisation are:
(a) complete enclosure of the cavity by sealing off the bottom open end
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ERA Report 99-0164
(b) uniform increase in the tank volume using a sliding plate, rather than conducting blocks
(c) increasing the Q of the cavity by lining it with good conducting material such as copper or
aluminium in order to promote frequency stabilisation.
4.
Overall Conclusion
Each of the methods outlined above offer a reduction in emissions from ISM equipment. The
reduction obtained depends on the type of machine that is operating. Where higher levels of
attenuation are required then a combination of individual techniques will achieve this objective e.g. a
polariser combined with a waveguide vestibule.
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ERA Report 99-0164
Contents
Page No.
1. Introduction
13
1.1 Nature of RFI Problem
13
1.2 Proposed Techniques and Technical Discussion on Initial Results
13
1.2.1
1.2.2
1.2.3
Polarisers for the reduction of radiated emissions
Vestibules – Waveguide techniques
Experimental procedure to control frequency of operation
1.3 Work Programme Objectives and Scope
2. Polarisers
2.1 Overview of Polariser Theory
2.1.1
2.1.2
Diffraction theory
Attenuation theory
2.2 Computer Program – POLGRID.Pas
2.2.1
2.2.2
2.2.3
Input parameters
Calculations
Program output
14
14
14
15
17
17
17
17
19
19
19
20
2.3 Experimental Work
20
2.4 Results and Discussion
20
3. Vestibules
22
3.1 Introduction
22
3.2 Attenuation Concepts
22
3.2.1
3.2.2
3.2.3
3.2.4
Waveguide attenuation
Filter concept
Modifying the RF path
Attenuator vane concept
3.3 Attenuation of RF Energy using T-Wave Guide Piece
3.3.1
3.3.2
Simulation method and geometry
Theoretical results
3.4 Experimental Results
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22
23
23
23
24
24
24
25
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ERA Report 99-0164
3.5 Conclusion
25
3.6 Further Work and Recommendations
26
3.7 Reference
26
4. Frequency Stabilisation
27
4.1 Introduction
27
4.2 ISM Dielectric Heaters – Typical Electrical Circuit
27
4.2.1
4.2.2
4.2.3
Tank or oscillator circuit
Applicator circuit
Coupling of RF power to the applicator
4.3 Frequency Stabilisation Procedure
4.3.1
4.3.2
4.3.3
4.3.4
Equipment required
System calibration
Measurement set-up
Measurement procedure
27
28
28
28
28
29
29
29
4.4 Experimental Results and Discussion
30
4.5 Conclusions
31
4.6 Recommendations for Further Work
31
Appendix A: Task 100 "Polarisers for reducing radiated emissions from
ISM machines operating at 13.56 MHz and 27.12 MHz
59-150
Appendix B: Task 200 "Assessment of vestibules and associated structures
for the spurious emission reduction from 27 MHz ISM heating
machines
151-246
Appendix C: Task 300 "Frequency stabilisation technique for ISM machines
operating at 13.56 MHz and 27.12 MHz
247-284
Appendix D: Flow Charts
Note: Page numbering in the appended reports is as per the original documents.
PB/vs/33/Rep5117
285-291
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ERA Report 99-0164
Tables List
Page No.
Table 1.1
Summary of the work programme presented in this report
16
Table 2.1
Summary of polariser attenuation as a function of frequency
33
Figures List
Page No.
Figure 2.1a
An array of wires of diameter a, equally spaced with periodicity, g
34
Figure 2.1b
An electromagnetic wave incident at an angle, phi, is diffracted at an angle,
theta, by the wire grating
34
Figure 2.2
Variation of the received power measured with and without a polariser grid
35
Figure 2.3
Variation of the received power measured with and without a polariser grid
36
Figure 2.4
Variation of the received power measured with and without a polariser grid
37
Figure 2.5
Variation of the received power with and without a polariser grid
38
Figure 2.6
Variation of the horizontally polarised (with respect to polariser) received
power measured with and without a polariser grid
39
Figure 2.7
Polariser grid attenuation – a comparison between theory and experiment
40
Figure 3.1
T-junction ports
41
Figure 3.2
Straight waveguide (Reference case)
41
Figure 3.3
T-junction + metallic plates + RAM inside the guide
41
Figure 3.4
RAM in the centre
42
Figure 3.5
Waveguide geometry
43
Figure 3.6
T-junction + metallic plates 1 and 2
44
Figure 3.7
T-junction + RAM + metallic plates 1 and 2
45
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ERA Report 99-0164
Figure 3.8
Test piece geometry (waveguide WG 11A)
46
Figure 3.9
RAM on sides and centre
47
Figure 3.10
T-junction + RAM on sides and centre + metallic plates
48
Figure 4.1
Mutual coupling between the tank and applicator circuits
49
Figure 4.2
Experimental layout for making return loss and impedance measurements of the
dielectric heater’s tank and applicator circuit
50
Schematic diagram of the C-type plastic welder machine showing the position of
the loop antenna within the tank cavity
51
Return loss spectra measured with the applicator ON and OFF on the 50 MHz
C-type dielectric plastic welder
52
Frequency change resulting in varying the position of the power flap above the
tank cavity of the 50 MHz C-type plastic welding dielectric heater
53
Variation of the resonant tank frequency due to loading with conducting blocks
inside the tank cavity of the 50 MHz C-type plastic welder
54
Proposed techniques to decrease and increase the tank cavity volume
55
Figure 4.3
Figure 4.4
Figure 4.5
Figure 4.6
Figure 4.7
PB/vs/33/Rep5117
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ERA Report 99-0164
Abbreviations List
RA
RFI
ISM
OATS
FDTD
TEM mode
ITU
EMC
RAM
DUT
PB/vs/33/Rep5117
Radiocommunications Agency
Radio Frequency Interference
Industrial, Scientific, Medical
Open Area Test Site
Finite-Difference Time-Domain
Transverse Electromagnetic mode
International Telecommunications Union
Electromagnetic Compatibility
Radio frequency Absorbing Materials
Device Under Test
12
ERA Report 99-0164
This page is intentionally left blank
PB/vs/33/Rep5117
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ERA Report 99-0164
1.
Introduction
The use of dielectric heating for use in many industrial applications is not new. Since the mid to late
1940s that has become an industry in its own right. Previous to that time, most equipment was built as
a sideline by some of the large electronic engineering companies such as STC, GEC and Redifon.
Until recently, interference from 13 MHz and 27 MHz ISM machines could be resolved simply by
retuning the fundamental so that any offending harmonic was moved to an area of the radio spectrum
that had no nearby radio communication usage. This is no longer a viable answer because the demand
for broadcasting, navigation, emergency services and communication has grown to the point where
there is no such spare capacity in the over-crowded spectrum. With the increasing use of
radiocommunications in society, the demand is for fault free communications, where frequency
stability and reduction in higher harmonic emissions is of prime concern for dielectric heaters.
1.1
Nature of RFI Problem
Cases of interference from industrial RF processing machines to various radio, television and
navigation users, including the emergency services, continue to occur. The radiated emission limits
that the industrial RF processing machines in use are expected to meet are presently set out in SI
1675/71. The harmonised European standard EN 55011, which reflects CISPR 11, sets the EMC
levels for the placing on the market or taking into service of new machines under the EMC
regulations. Legally, companies who have taken into service machines which do not comply, could be
forced to close their operations. However, such an action may have an adverse effect on industry in
terms of job losses or small business closures. Recent research work funded by the
Radiocommunications Agency highlighted the use of polarisers and vestibules as promising methods
of reducing disturbance from ISM machines at relatively low cost. Alternative methods of mitigation,
such as updating oscillators have relatively high levels of cost, especially when the modification is an
upgrade to an old ISM machine.
1.2
Proposed Techniques and Technical Discussion on Initial Results
ISM equipment are known sources of radiated emissions. Where the fundamental frequency remains
within the stipulated tolerances and where the harmonic content complies with the relevant standards,
then there are few EMC problems. However, the design and construction of these machines, many of
which are assembled on site to a customised specification, may undergo subsequent modification
during their lifetime which renders harmonic control extremely difficult. Often the fundamental nature
of the product, e.g. a conveyor belt type in food manufacture, requires the presence of large apertures
which are potential leakage points, satisfactory in many cases in the factory prototype, but difficult to
achieve control in the practical installation. The horizontal polar distribution of the radiation can also
be affected by reflections and enhancements by local metal structures which may act as parasitic reradiators. Measurements made on site must be conducted with care in order to ensure that variations
in polar radiation patterns are taken into consideration.
PB/vs/33/Rep5117
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ERA Report 99-0164
1.2.1 Polarisers for the reduction of radiated emissions
Some simple experimental trials showed that attenuation of the unwanted harmonics was possible,
using the polarising technique. This gave up to 20 dB attenuation in some cases, but unexpected
amplification at some spot frequencies resulted primarily due to interference of scattered waved from
buildings and diffracted waves from the edges of the finite size polariser.
1.2.2 Vestibules – Waveguide techniques
Alternatively, for the conveyor type of ISM machines, the use of vestibules appears to be an attractive
means of attenuating radiated harmonics to the statutory limits given in the standards.
The method of enclosing the conveyor belt used in a number of continuous processing units in a long
metallic duct was worth pursuing. The duct would act as a waveguide below cut-off and hence
provide significant attenuation for the fundamental and a number of lower order harmonics. What
must be noted, however, is that care needed to be taken so that on an enclosed conveyor system,
metallic parts do not transform the waveguide into a “square coaxial line” capable of supporting TEM
waves with zero cut-off frequency. If so, it is possible that little or no attenuation will be afforded by
this scheme, allowing the energy to propagate along the ducting and radiation to occur at the newly
formed external aperture.
As far as harmonic reduction is concerned, a refinement of the waveguide duct idea was a possibility.
In the first approach, partitioning plates may be used so as to make sure that the effective width of the
duct is smaller than the one required for the propagation of the propagating ‘waveguide mode’ for a
given harmonic frequency. This is a similar approach to the one employed over many years in the
radar industry to reduce out-of-band emissions and is termed ‘harmonic filtering’.
Lining the duct walls with lossy material is another approach but may not produce the best result in
attenuating modes of propagation. The expensive ferrite tile approach or a lower cost carbon loaded
epoxy board implementation both merited closer evaluation. Such materials are widely used to
suppress radar reflections, and a range of radio frequency absorbing materials (RAM) is available
commercially, or could be synthesised for optimum performance at the specified frequency and/or its
harmonics.
1.2.3 Experimental procedure to control frequency of operation
Most older machines use Class C operation, whereby RF energy generated in the tank circuit is
inductively coupled to the applicator circuit. However, changes occurring during the work cycle in the
applicator circuit tend to offset the tank and hence the operation frequency due to the reflected
impedance from the applicator to the tank circuit.
By the use of circuit measurement techniques i.e. S-parameters and impedance, an understanding of
the importance of the elements in the three sections of ISM machines can be achieved in order to
reduce the reflected complex impedance back into the tank circuit, and hence achieve greater stability
in the fundamental operating frequency.
PB/vs/33/Rep5117
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ERA Report 99-0164
Measurements of the response of the circuit performance of the system due to changes in the elements
can be made. These would enable the relative importance of changes to the elements to be identified,
and what control of changes may need to be implemented. A methodology which describes the
measurements to be made and the priority level for changing elements which control the fundamental
frequency, would enable remedial changes to be more effective, and at very little cost.
1.3
Work Programme Objectives and Scope
Frequency stabilisation, attenuation of radiated emissions and harmonic suppression were identified in
reducing RFI from dielectric heaters. The work programme given in this report concentrates on the
following objectives:
1. To interpret the theory of polarisers and vestibules in its simplest form such that it could readily
by applicable to practical situations.
2. To back theoretical and/or a computer modelling/program with experiments to validate the
practicality of the innovations proposed in combating RFI.
3. To provide procedures, computer programs and flow diagrams which will aid RA staff in
applying and conveying the information to RF users.
Table 1.1 summarises the key areas tackled and what has been delivered to RA. Although this report
concentrates on polarisers, vestibules and frequency stabilisation, the overall work, including the first
phase of the work, allows one to take a structured approach in combating RFI. This approach is
presented as flow diagrams which addresses frequency stabilisation, radiated emissions and harmonic
suppression as applied to four differently designed dielectric heaters.
The scope of the work presented in this report concentrates on bringing together the theoretical
aspects with experimental results of the three tasks in order to provide validity of the techniques used
in practical situations for reducing RFI.
These flow diagrams are presented in Appendix D.
PB/vs/33/Rep5117
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ERA Report 99-0164
Table 1.1
Summary of the work programme presented in this report
ISM Problem
Practical
Solution
Deliverable(s) to RA
Experimental
Investigation
Comments
Radiated
emissions
Polarisers
(optical
and
microwave
techniques)
Theoretical
model,
computer
program,
experimental results
Yes
Polarisation sensitive, providing 10 to 20 dB
attenuation to em waves with polarisation
parallel to the wires. Advantages are (1)
cheap, (2) easy to construct, (3) allows
frequency selection for attenuation with the
aid of a computer program, and (4) allows
cross polarised em waves to be transmitted
with very little loss.
Vestibules
(microwave and
waveguide
techniques)
Theoretical model and
results, and prototype of
waveguide T-piece and
experimental results
Yes
Application of microwave waveguide
techniques to low frequency em wave
propagation. Leaked em waves are guided
through a special vestibule and are diverted
into an absorbing chamber. A good
agreement between theoretical analysis and
measurements has been obtained. Difficult to
apply on C-frame and shuttle tray type
machines.
Low
power
evaluations using
an
automatic
network analyser
Experiments
procedure
for
conducting
measurements,
experimental results and
recommendations
Yes
The
test
procedure
provided
was
implemented on an old C-frame plastic
welder using low power techniques. The tank
frequency and Q of the tank cavity can be
accurately
determined.
Frequency
stabilisation can be achieved by altering
some of the tank parameters. This procedure
is most useful for RIS engineers and RF
users of dielectric machines.
Flow diagrams
which can be
systematically
applied
to
various types of
dielectric heaters
in
order
to
reduce
RFI
problems
Easy to use flow diagram
No
These flow diagrams are most useful when
reducing RFI problems. They encapsulate all
remedial steps which are systematically
applied, and includes the practical solutions
stated above.
(Task 100)
Radiated
harmonic
emissions from
coveyorised
systems
(Task 200)
Frequency
stabilisation
(Task 300)
Frequency
stabilisation,
harmonic
suppression,
radiated emissions
PB/vs/33/Rep5117
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ERA Report 99-0164
2.
Polarisers
A polariser can be considered to have two unique properties. Firstly, it allows transmission of
electromagnetic (em) waves which have the electric component perpendicular to the wires (assuming
the wavelength of the incident wave to be much longer than the wire spacing) but attenuates waves
which have the electric component parallel to the wires. Secondly, they also act as dispersers of em
energy, similar to a diffraction grating of the transmission type. These properties are utilised in
reducing leaked em energy from buildings containing dielectric heaters, and thereby reducing the
problem of RFI on nearby radio services. In optimising the attenuation of the transmitted em waves at
a frequency or a series of frequencies, the polariser’s parameters can be selected beforehand with the
aid of a computer program.
2.1
Overview of Polariser Theory
The polariser affects the incident wave by attenuating the transmitted wave if the polarisation is
parallel to the wires (attenuation theory) and disperses the transmitted wave (diffraction theory). The
full theoretical treatments are given in Appendix A.
2.1.1 Diffraction theory
Figure 2.1 shows a wire grid grating of N wires, where a is the diameter of the wires and g is the mean
wire spacing. The wires are assumed to be infinite in length. The normalised intensity of the
transmitted diffracted wave is
1  sin    sin N 

Normalised Intensity = 2 
 .
N     sin  
2
2
(2.1)
where the first term in brackets corresponds to the diffraction factor, and the second term in brackets
corresponds to the interference factor.
The terms  and  are given as
=
g
(sin  - sin )

(2.2)
=
 (g  a)
(sin  - sin )

(2.3)
2.1.2 Attenuation theory
The general formula for transmission through a wire grid is given as
1=T+R+A
PB/vs/33/Rep5117
(where T, R , and A are fractions i.e. less than 1)
(2.4)
18
ERA Report 99-0164
where T is the transmissivity, R is the reflectivity and A is the absorption. Lewis and Casey [7] have
considered a wire grid with finite conductivity. The grid parameters considered satisfied the condition

< (1 + sin ) where  is the angle of incident wave from the normal of the grid plane. The
g
quantities  2 and  2 , which are the intensity coefficients for reflection and transmission, are given
by

1
 2 = [([1  Ri' ]2  [ X p'  X i' ]2 ) 2 ]2
(2.5)
1
 2 = [(1 -  2 - 2  2 R i' ) 2 ] 2
(2.6)
where
R i' = (
Rs 2 g
)(
)
a  0
(2.7)
Rs 2 g
)(
)
a  0
(2.8)
X i' = i(
1
Rs = (f 0  ) 2
X p' =
2g cos 

(2.9)
[ln (
g
) + F]
a
(2.10)
where
F=
1
2


n 1
[([n + (
g

) sin] 2 - (
g

)2 )

1
2
+([n – (
g

)sin 2 - (
g

)2 )
where n is the order of the diffracted wave (n = 1 in our case)
 is the electrical resistivity of the wire (m)
 0 is the impedance of free space (~ 376.73 )
 is the magnetic permeability =  0 .  r , where  0 = 4 x 10 7 H/m
PB/vs/33/Rep5117

1
2
-
2
]
n
(2.11)
19
ERA Report 99-0164
2.2
Computer Program – POLGRID.Pas
This program was written and compiled in Turbo Pascal, version 7.1. Full details of the program are
given in Task 100, Appendix A.
2.2.1 Input parameters
After running the program, the following inputs are requested from the user:
1) Filename for the output data.
2) Wire diameter in metres, a.
3) Wire spacing in metres, g.
4) Total number of wires, N.
5) Electrical resistivity of the wire material,  (Ohm.m).
6) Relative permeability of the wire material,  r .
Once these have been entered, then the program requests “Please enter choice of calculation”. The
two choices displayed on the screen are:
A) “Reflection and transmission attenuation as a function of angle, please enter <1>”.
B) “Reflection and transmission attenuation as a function of frequency, please enter <2>”.
On choosing A), the program will then ask for the frequency (in MHz) for the calculations. On
choosing B), the program will ask for the operating frequency (in MHz) of the dielectric machine for
calculations. On entering these parameters the program performs the calculations.
2.2.2 Calculations
The following calculation are performed as a function of angle or frequency, depending upon the
choice of calculations required (see Appendix A for equations).
(i)
The normalised intensity, equation 2.1.
(ii)
The intensity coefficient for reflection, equation 2.10.
(iii)
The intensity coefficient for transmission, equation 2.11.
(iv)
Transmission intensity = normalised intensity x intensity coefficient for transmission.
(v)
Reflection intensity = normalised intensity x intensity coefficient for reflection.
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 Normalised Intensity 
Transmission intensity (dB) = 10 log 10 
 = 10 log 10 (intensity
 Transmission Intensity 
(vi)
coefficient transmission).
In addition, one can manipulate the results in Excel to get
(vii)
Reflection intensity (dB) = 10 log 10 (Intensity coefficient for reflection)
(viii)
Absorption loss (dB) = 10 log 10 [1 – (intensity coefficient for transmission + intensity
coefficient for reflection)]
2.2.3 Program output
The computer program first outputs all the input parameters the user has typed in, then the
calculations (either as a function of incident angle or as a function of frequency). The first column is
either angle in degrees or frequency in MHz, followed by the calculations from (i) to (vi) in that order
in the following columns, see Appendix A.
2.3
Experimental Work
A polariser was built at ERA which has the dimensions of height = 3m and width = 4m. The wires
used were copper, of diameter 1 mm and were spaced first at 100 mm and then later at 50 mm (the
total number of wires being 39 and 78, respectively). The wires were arranged parallel to the height.
The actual frame for supporting the wires was made of wood.
All measurements were made on two sites at ERA. Site 1 was close to the ERA covered open area test
site (OATS), and Site 2 was over an area close to the North Building at ERA.
In order to assess the polariser’s transmission attenuation, measurements of the received signals were
made with and without the polariser, so that the transmission attenuation could be determined from
Transmission attenuation (dB) = Received signal without polariser (dBm) – Received signal with polariser (dBm)
In all the results presented, the lighter solid trace represents measurements of the received signal
without the polariser and the darker solid trace represents measurements of the received signal with
the polariser.
2.4
Results and Discussion
Figures 2.2 to 2.5 show measurements of the received power in dBm versus frequency. The fine
curves are measurements made without the polariser grid and the dark curves are measurements made
with the polariser grid. The sharp dips observed in measurements made without the polariser grid
represent destructive interference of the direct wave with waves reflected off the ground and the walls
of the OATS chamber. These dips become less pronounced when the polariser grid is placed between
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the antennas and is akin to the dispersive properties of the polariser acting as a diffraction grating.
Figures 2.2 to 2.5 were made with the biconical antennas orientated in the same polarisation as the
wires in the grid. Figure 2.6 shows the same measurements, except the antennas were orientated
horizontally with respect to the polariser. The expected result is seen of the polariser having very little
influence on waves polarised perpendicular to the wires.
The summary table, Table 2.1, shows the measured attenuation at harmonics of the 27.12 MHz
frequency up to 300 MHz. Figure 2.7 shows plots of the theoretical attenuation curve (dark line) and
the mean experimental attenuation curve (thin line) and also the best fit experimental curve (dashed
line). The discrepancy below 70 MHz is due to the close proximity of antennas and the effects of
reflections.
Similar measurements were made on Site 2, which was located within the confines of buildings, the
effects of scattering marred the results, but overall there was ~10 dB attenuation through the polariser
(see Appendix A).
All measurements show encouraging results over certain frequency ranges, the attenuation through the
polariser grid can be as high as 30 dB. A computer program provided can be used to fine tune the
performance of the polariser such that one can achieve the adequate attenuation in order to prevent
RFI. The advantage of a two-dimensional grid (see Appendix A) is to provide equal attenuation to the
vertically and horizontally polarised waves radiated by an ISM equipment. Although measurements
and theoretical analysis were made up to 1 GHz, there is no limit in the design of polarisers to meet
the demands at much higher frequencies.
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3.
Vestibules
3.1
Introduction
For ISM machines with the fundamental operation at 27 MHz, interference problems can occur up to
the 30th harmonic. This part of the work reports on ways by which the spuriously emitted field
strength can be substantially reduced over a very broadband using waveguide techniques. Its primary
focus is on cases where a metal duct is used to transport goods in and out of a heating machine. The
waveguide techniques relies on the assumption that RF harmonic emissions escape along routes
confined inside these input and output transportation ducts. The methods presented are based on the
modification of the duct geometry in such way that electromagnetic emissions will find it hard to
escape along the duct routes while at the same time allows the processed materials to be conveyed
unimpeded in and out of the duct.
3.2
Attenuation Concepts
The main difficulties of the problem lies in the fact that:
1) Attenuation of radiated emissions is required over an extremely broad frequency range.
2) The primary excitation is not well defined since an arbitrary number of modes can be excited.
3) At high frequencies, the duct represents a highly overmoded wave guide system.
4) The solution found should be implemented with minimum cost and its introduction to the service
should take place with minimum or preferably zero modification to existing ISM heating
installation. The concepts available to the design can be summarise as follows:
a) Waveguide attenuation characteristic, waveguide cut-off techniques.
b) Filter concepts involving cavities.
c) Modification of RF preferred path and subsequent attenuation to dummy load.
d) RF vane attenuation concept using absorbing materials.
A combination of the above methods will all contribute to the proposed solution in attenuating
radiated emissions.
3.2.1 Waveguide attenuation
This method relies on the fact that a waveguide below cut-off can add significant attenuation to
signals below cut-off. For those signals above cut-off the basic waveguide offers almost perfect
propagation. Under these conditions, the only way to provide some attenuation is when the duct walls
are covered with lossy materials. A detailed study on the attenuation characteristics of ducts has
already been reported (see Appendix A). The main conclusion was the significant attenuation below
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cut-off, this can only be exploited if the duct has the minimum possible cross section dimensions. For
harmonics above cut-off the waveguide structure can offer little attenuation even in cases where the
walls are lined with absorbing materials.
3.2.2 Filter concept
The idea is to modify the duct geometry in a way that it serves as a filter to reject harmonic
frequencies up to 1 GHz, and possibly higher. The implementation of this concept is complicated by
the fact that no well-defined excitation exists which will allow the excitation and propagation of
several modes to occur simultaneously at higher frequencies. A consequence of this will be on what
type of working mode(s) will be generated in practice. Within the limited period of the current work,
it was not possible to study the filter concept into great detail. What has been examined was the case
where a lossy cavity was coupled to the main duct by means of a slot. The cavity would resonate at
some selected harmonics and the coupled power into the lossy cavity would be absorbed. The results
on coupled cavities have already been presented in Appendix B. The main conclusion is that although
significant attenuation can be offered to the coupled modes, the associated frequency response is quite
narrow band, and hence from a practical point of view tuning may be required for every installation
utilising cavity filters in order to couple and absorb the offending frequency(s) of emission.
3.2.3 Modifying the RF path
This concept is based on the use of a T-piece (Fig. 3.1) in such a way that path 2-3 offers minimum
attenuation and path 1-2 is the maximum attenuation path. If port 3 is terminated to a matched load,
then the radiated emissions appearing at port 3 will be absorbed.
Results from the application of this concept are given in Section 3.4.
3.2.4 Attenuator vane concept
It is well known that attenuation to a propagating em wave can be provided if an absorbing material is
placed inside the waveguide at locations of maximum field strength. This is the principle behind the
vane attenuator - a common microwave laboratory component. The limitation of this method is that
the depth of the absorbing material has to be limited, such that, it would not hinder the passage of
processed material conveyed to the output port. The selection of material is another consideration.
High performance radio frequency absorbing materials (RAM) exist, but tend to be expensive.
However, lower cost alternatives can be made by dispersing carbon or iron powder in epoxies which
can be shaped to the required form.
Results are presented to show the effectiveness of this approach. Although attenuation tends to
increase with frequency, this is limited to about 5-10 dB for the frequency range of interest. Although
this method on its own cannot provide realistic solution, it can be employed in conjunction with other
methods, for enhanced attenuation performance.
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3.3
Attenuation of RF Energy using T-Wave Guide Piece
3.3.1 Simulation method and geometry
The task of analysis/design of the RF emission attenuation structure was largely carried out using
theoretical modelling tools.
The electromagnetic simulations was based on the Finite-Difference Time-Domain (FDTD) technique
[1]. The principal reasons for this choice are the following:

A full wave analysis can be achieved since all waveguide modes which are excited and
propagated inside the vestibule will be automatically accounted for in the simulation.

The addition of absorbing materials and metallic plates inside the structure can be done in a very
flexible and powerful way.

Broadband results are given from a single computer run.
In FDTD computation, the structure to be analysed is divided into small cells of finite size. Material is
allocated to the different edges of each cell and a calculation of the electromagnetic field is
undertaken using the Maxwell's equations formulated in the time domain. The quantisation of the
geometry due the cell size gives rise to a stair case approximation of the real dimension of the
structure. The approximation may be made more and more accurate by increasing the number of cells.
Therefore, the larger the number of cells, the more accurate the computed field.
The theoretical modelling will deal with waveguide structure. The excitation has been accomplished
with the introduction of posts that can excite both the TE and TM modes. The quantity determined is
the scattering parameter S21 (that is, the signal out of port 2 from the signal in at port 1, see Fig. 3.1)
which is compared with the same S21 parameter (see Fig. 3.2) determined when a straight reference
waveguide is considered. The ducts were assumed to have a cross section of 1.32  0.2 metres. The
reference straight waveguide was 4 metres long. The longitudinal cross section of the reference case
and the proposed structure can be seen in Figs 3.2 and 3.3 respectively.
3.3.2 Theoretical results
The first structure analysed was a straight waveguide which had a thin RAM card placed along the
full length of the waveguide. This card was centred in the waveguides ‘a’ dimension and it protruded
0.1 m parallel to the b dimension inside the waveguide. The RAM materials electrical properties are
similar to ECCSORB MF-117 (r=9, tane=0.07, r=1.8, tanm=0.6). The attenuation characteristics of
this structure can be seen in Fig. 3.4 by the solid curve. The dashed curve represents S 21 spectrum of
the reference waveguide, Fig. 3.2.
It is verified that the straight attenuation card structure on its own cannot provide high attenuation
over the frequency range of interest.
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The attenuation characteristics of the modified T-piece was then analysed. The geometry and
dimensions can be seen in Figs 3.3 and 3.5. Metal plates 1 and 2 (Fig. 3.5) serve the purpose of
directing most of the energy to the loaded port 3 of the T-piece. The performance of the structure can
be seen in Fig. 3.6. Although there are isolated frequencies where attenuation offered is moderate, the
overall performance of this structure guarantees at least 15 dB of attenuation over a broader frequency
range. If we can combine the T-piece with a RAM card as shown in Fig. 3.3, an improved
performance resulted, as shown in Fig. 3.7. An overall attenuation in excess of 20 dB was achieved.
One can easily construct and install such a structure at low cost without drastically modifying the
converised system.
3.4
Experimental Results
The T-piece waveguide structure theoretically investigated showed a significant potential for reducing
harmonic emissions. An experimental verification was then made to validate predictions. Within the
context of this project a full size experimental set-up was proposed but due to financial/timescale
constraints an experimental model was designed and constructed to work at higher frequencies using
existing standard waveguide structures. This test piece made using waveguide WG11A (frequency
3.2-4.9 GHz). As time was short the objective of the experimental investigation was to check the
accuracy with which the adopted theoretical tools can predict real performance.
The geometry and dimensions of the test piece are shown in Fig. 3.8, where all the dimensions are in
millimetres. An equal length piece of straight waveguide was also made to provide reference S21 data.
In this figure, fc_TE10 refers to the cut-off frequency of TE10 mode of the waveguide. Very good
agreement between theory and practice are shown.
Figure 3.9 shows measurement spectrum of S21 (solid curve) made on a straight waveguide which had
a RAM card running central along the length of the waveguide. This curve is compared with the
simulation result of S21 (dashed curve) for the same waveguide configuration.
Figure 3.10 shows the frequency response of the T-piece (Fig. 3.8). The RAM cards were inserted in
the input and output guide as discussed previously (Fig. 3.3). The loaded port 3 was terminated with
flat absorbing material. The agreement between theory (dashed curve) and experimental measurement
(solid curve) are good as both the general trend and levels follow closely to one another.
3.5
Conclusion
Various methods have been studied in order to reduce the level of spurious RF emission escaping
from ISM heating machines. These techniques assume that harmonic emissions are leaked to the
outside through the ducts which are normally used to transfer items in and out the heating machine.
This work study backed by experimental results, show that one can successfully apply microwave
techniques to 27 MHz machines. From the findings, the following conclusions are given:
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
Straight waveguide can only be useful if the troublesome harmonic frequencies lie below the cutoff frequency of the fundamental TE 10 mode of the waveguide. Best results are expected when
the duct cross section is made as small as practically possible.

A parallel plate duct should be avoided as it can support a transverse electromagnetic mode
(TEM) with zero cut-off frequency.

For harmonics at frequencies above cut-off the basic waveguide structure offers limited
attenuation. This is also true in cases where the duct walls are lined with absorbers or where RAM
is protruded inside the waveguide in a way similar to a vane attenuator component.

A modified waveguide T-piece has been proposed as a solution in reducing broadband emissions
from vestibules on ISM machines. These offer an average attenuation in excess of 15 dB over the
frequency range considered.

Combination of the T-piece waveguide and the vane attenuator concept can provide increased
attenuations of over 20 dB for frequencies up to 1 GHz.

The theoretical analysis was validated by comparisons with experimental results. This provides
confidence in adopting the T-piece waveguide technique on conveyorised systems.

This technique can also be applied to conveyorised ISM machines operating at frequencies much
higher than 27 MHz.
3.6
Further Work and Recommendations
The T-piece waveguide technique in reducing radiated emissions from converised ISM machines can
be designed to meet the EN 55011 requirements at any frequency or frequency range. The theoretical
analysis and simulations are in no way complete and there is scope to do further work which will
(a) optimise the T-piece performance,
(b) look at other waveguide techniques for reducing radiated emissions,
(c) optimisation of RAM placement within the waveguide T-piece, and finally
(d) to implement the T-piece on existing ISM machines which utilise conveyorised systems.
3.7
[1]
Reference
K. Kuntz, R Luebbers, ‘Finite Difference Time Domain Method for Electromagnetics’
CRC Press, 1993
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4.
Frequency Stabilisation
4.1
Introduction
There are a large number of dielectric heaters operating around the 13.56 MHz and 27.12 MHz
frequencies, however the problem of frequency stability and operation remote to these frequencies are
a major concern when dealing with RFI cases. The aim of this work was to implement the frequency
stabilisation procedure in stabilising the frequency of operation for a given ISM machine.
The main objectives of this part of the work was to:
1) Validate the frequency stabilisation procedure.
2) Determine which parameters affect frequency drift.
3) Provide a training aid for Radiocommunications Agency staff.
Since a large portion of ISM machines operate under old designs, this work will aid RA staff in giving
practical advice to ISM users in methods of stabilising the frequency of operation to be within the ITU
band.
4.2
ISM Dielectric Heaters – Typical Electrical Circuit
The main components of a typical dielectric heater are:
i)
DC high voltage power source.
ii)
A triode.
iii)
Tank or oscillator circuit.
iv)
The applicator.
v)
Feedback loop.
In order to implement the frequency stabilisation procedure, the two circuits of interest are the tank
and applicator.
Figure 4.1 shows the tank and applicator circuit (parallel plate). The RF energy is inductively coupled
to the applicator circuit from the tank circuit. The components shown are made up of either distributed
parameters or lumped elements.
4.2.1 Tank or oscillator circuit
The tank circuit has two functions, firstly to govern the frequency of operation for the dielectric
heater, and secondly to couple RF energy to the applicator. The frequency of the RF energy generated
in the tank circuit during the operation is given by:
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f t = (2
L t C t ) 1
(4.1)
This frequency is fixed and can only be changed by altering the values of C t and L t . Since the tank
circuit is made up of lumped elements, one can view the tank as being made up as a metallic resonant
cavity.
4.2.2 Applicator circuit
The applicator circuit is inductively coupled to the tank circuit. This circuit is made up of an inductor,
L a (which is determined by particular shape of the applicator), a capacitor, C a (which is the function
of many parameters, such as electrode type, dimensions and spacing, dielectric properties of the
material, etc.) and a resistor, R a , which is dependent on the conductivity of the dielectric material.
The frequency of the applicator is given by
f a = (2
L a C a ) 1
(4.2)
4.2.3 Coupling of RF power to the applicator
The tank circuit, as the name suggests, is basically a metallic container which contains the triode.
Being a metallic container, the tank resonates when excited by RF energy delivered to it by the triode
(pulsed at
1
, time interval). This RF energy of frequency, f t is coupled out to the applicator by a
ft
metallic plate in the form of a single loop. The length of this plate which is inside the tank couples the
RF energy to one plate of the applicator, the other being grounded. The RF electric field is then
established between the applicator plates, into which the workpiece is inserted.
4.3
Frequency Stabilisation Procedure
This technique uses low power excitation of the tank cavity with the aid of an automatic network
analyser. Full details of this technique is discussed in Appendix C.
4.3.1 Equipment required
The following equipment and components are essential in order to adjust the dielectric heater’s
operation frequency.
1) An automatic network analyser (10-100 MHz bandwidth).
2) A compatible s-parameter test set.
3) A 50  N-type calibration kit comprising of an open, short and 50  load.
4) Two non-resonant loops, each connected to 50  connectors.
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5) Two 3 m long N-type 50  cables.
6) Two non-conducting supports for the loops which can elevate or rotate the loops.
7) A high quality through N-type 50  connector.
4.3.2 System calibration
A full 2-port calibration is required, this procedure is given in Appendix C.
4.3.3 Measurement set-up
The experimental set up for making impedance, return loss and insertion loss measurements on the
dielectric heater assembly is shown in Fig. 4.2. The launching loop connected to port 1 is orientated
inside the tank in order to excite the transverse electromagnetic mode (TEM mode). The RF energy is
coupled out of the tank by the coupling loop which is connected across the parallel plate applicator, as
shown in Fig. 4.2. The RF energy delivered to the parallel plates establishes an RF electric field
across the plate. The receive loop is positioned such that the RF magnetic flux threads through it
thereby inducing an RF current in the receive loop.
4.3.4 Measurement procedure
Making measurements of the return loss is straightforward. The ratio of the reflected signal to the
incident signal (that is, out of port 1 and into port 1, which is S11 (measurement of magnitude and
phase) is required to be measured. Using an ANA, select S11, press 'meas' and then option "Refl:FWD
 A
S11   " (which is usually default). Then press 'Format' and then "LOG MAG" which will display
R
S11 (dB) as a function of frequency. What will be viewed on the screen will be similar to Fig. 3.2(b),
in Appendix C.
In order to acquire the real/imaginary impedance (R+jX) data of S 11 (complex reflection coefficient),
press 'Format' and then "Smith". The Smith chart will show the complex impedance of the DUT over
the frequency range selected.
The amount of power reflected from the DUT is directly related to the impedance values of both the
device and the measuring system. In fact, each value of the reflection coefficient () uniquely defines
a device impedance. For example:-  = 0 occurs when the DUT and test set impedance are the same.
A short circuit has a reflection coefficient of  = 1 |_ 180 0 (= -1). An open circuit has a reflection
coefficient of  = 1 |_ 0 0 (=1). Every other value for  also corresponds uniquely to a complex device
impedance, according to
Zn =
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1 
1 
(4.3)
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ERA Report 99-0164
where Zn is the DUT impedance normalised (that is, divided by) the measuring system's characteristic
impedance (= 50 , in this case). By pressing 'Marker' and turning the knob, one can read the resistive
and reactive components of the complex impedance at any point on the trace (or frequency value).
Note the Smith Chart display is similar to Fig. 3.2(a) in Appendix C.
4.4
Experimental Results and Discussion
All measurements were done on a small c-type plastic welder which operated at about 50 MHz, see
Fig. 4.3 which shows a schematic diagram of the machine layout.
The tank cavity shown on the left contains the power triode which is connected to a rigid inductor
post onto which the capacitor plate is connected. The conductor post extracts the RF energy from the
tank cavity and couples it capacitively between the capacitor plate and the earth plate to the applicator
head and the earthed workpiece. The applicator is simply a parallel plate capacitor into which the
work is placed and RF energy applied for processing. The tank cavity is partially closed at the top by
the power flap and open at the bottom.
The actual resonant frequency of the tank circuit is governed by the lumped inductor and capacitor
elements within the tank circuit, and not the physical dimensions of the tank.
Figure 4.4 shows the return loss spectra measured by altering the applicator spacing. The dashed
curve was obtained by having the applicator spacing at maximum setting, and the solid curve
represents the return loss measured when the applicator spacing is set to it minimum value. There are
two things that are observed when the applicator spacing is reduced, firstly a frequency reduction
from 51.18 MHz to 50.96 MHz and secondly, the Q of the tank cavity is reduced from 290 to 250.
The frequency shift and the reduction in Q results from the change in the tank’s lumped parameters
due to the reflected impedance from the applicator circuit.
Figure 4.5 shows the effect of varying the power flap position. When the power flap is set at its
minimum position, that is into the tank cavity, the tank cavity’s volume is reduced and hence the
frequency measured is high as shown by the solid curve. On raising the power flap to its maximum
setting, and thereby increasing the tank cavity’s volume, the frequency of the tank is reduced, as
shown by the dashed curve. The overall frequency change is about 2.6 MHz. However, the actual
setting of the power flap will govern the degree of RF coupling to the applicator. If this flap is set too
close to the capacitor plate, then the electric field strength established between the plates will increase
to a level whereby sparking and coronas will be generated resulting in very little RF coupling to the
applicator. A low setting of the power flap would result in a reduction of RF coupling from the tank to
the applicator circuit. This power flap has to be finely tuned to a position which would cause good RF
coupling but without generating electrical sparks and coronas.
Figure 4.6 shows the effect of reducing the tank cavity’s volume. This was achieved by loading the
cavity with polystyrene blocks lined with aluminium foil to make them conducting. The tank volume
is 2.704 x 10 2 m 3 and the total volume of the conducting blocks is 4.1855 x 10 3 m 3 , thus the
reduction in the tank volume is 15.5%. In Fig. 4.6, the dashed curve represents the return loss
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measured from the unloaded tank cavity, and the solid curve represents the return loss measured from
the loaded tank cavity. The overall frequency increase on reducing the tank cavity volume is about 3.5
MHz. Thus by fine tuning of the tank volume it is possible to make the tank frequency, and hence the
operating frequency, to be within twice the fundamental frequency of 27.12 MHz, that is 54.24 MHz.
The return loss is also reduced, which signifies an increase in the Q of the cavity as a result of using
aluminium lined blocks which have higher conductivity than iron.
Figure 4.7 shows methods whereby one can decrease or increase the tank cavity volume in an ideal
manner rather that haphazardly loading the tank at various positions as shown in Fig. 4.3. The
advantage of uniformly increasing or decreasing the tank cavity volume is to maintain the purity of
the resonant mode excited in the cavity. Also, one can use a good conducting material to make the
sliding plates in order to increase the Q of the cavity and increase frequency stability.
4.5
Conclusions
From the three techniques employed, by far the best way to establish the tank, and hence the operation
frequency, is to vary the tank cavity volume, as demonstrated in Fig. 4.6. The problems encountered
when changing the power flap spacing are in obtaining good coupling and the possibility of incurring
electrical sparking and coronas. In varying the applicator spacing over approximately 10 cm, the
frequency change was approximately 0.2 MHz. However, the spacing change over the workpiece may
only be a few millimetres over the work cycle and the frequency shift may not be enough as compared
to the frequency shift due to the dielectric property change during RF processing.
In varying the tank cavity volume, using the methods shown in Fig. 4.6, one effectively changes the
resonant frequency of the fundamental mode. This frequency can then be finely tuned by varying the
power flap by small increments in order to achieve adequate RF coupling to the applicator.
Observations of the C-Frame dielectric plastic welder revealed the tank cavity was open at the bottom
and thus presents a major source of leaked electromagnetic radiation. This would also reduce the
Quality factor of the cavity since, by definition, the Q of a cavity is the energy stored per cycle
divided by the energy lost per cycle. The losses in the cavity are thus attributed to the aperture at the
bottom and I 2 R or ohmic losses within the cavity walls.
4.6
Recommendations for Further Work
Although the experimental work showed three methods by which the tank, and hence the operation
frequency, could be varied, there is still scope for further work which could be performed. These are
listed below:
1) Measurements of radiated emissions without and with a metal plate closure at the bottom of the
tank cavity.
2) Measurement of the frequency shift due to changing the tank cavity volume, as shown in Fig. 4.6.
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3) Increasing the Q of the cavity by lining it with copper or aluminium film in order to promote
frequency stability.
4) Since very little coupling to the applicator occurred, the resonant dip in the return loss
measurements is absent. It would be useful to excite the RF electric field at the parallel plates of
the applicator with the calibrated loop and then to measure the return loss. This should provide
information about the applicator and tank frequencies and also the RF coupling.
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Table 2.1
Summary of polariser attenuation as a function of frequency
Frequency
Theoretical
Figure 2.2
Figure 2.3
Figure 2.4
Figure 2.5
Mean
Best Fit
(MHz)
Attenuation
A (dB)
A (dB)
A (dB)
A (dB)
A (dB)
(dB)
(dB)
27.12
24.06
-7.6
-4.7
-6.4
-9.1
-6.95
-4.7
54.24
18.1
12.5
11.4
8.4
6.5
9.7
12.5
81.36
14.66
17.6
14.6
15.6
19
16.7
14.6
108.48
12.29
13.8
10.4
13.6
11.8
12.4
11.8
135.6
10.49
-1
-6
4.7
9.3
1.75
9.3
162.72
9.07
2.4
1.8
-9.7
1.3
-1.05
2.4
189.84
7.92
8.1
5.3
5.9
-3.7
3.9
8.1
216.96
6.97
12.2
8.1
5.8
6.3
8.1
6.3
244.08
6.18
1.4
2.11
8.3
11.1
5.7275
8.3
271.2
5.5
1.72
4.5
3.9
7.9
4.505
4.5
298.32
4.92
4.9
2.8
2.4
6.7
4.2
4.9
34
ERA Report 99-0164
E
H
S
a
N
g
(a)
N
(b)
Figure 2.1 : (a) An array of wires of diameter a, equally spaced with periodicity, g.
(b) An electromagnetic wave incident at an angle phi is diffracted at an angle,
theta by the wire grating.
PB/vs/33/Rep5117
35
ERA Report 99-0164
Figure 2.2 Variation of the Received Power measured with and without a Polariser
Grid (wire diameter = 1mm, wire spacing = 100mm, N = 39, width = 4m, height = 3m,
Tx {ht = 1.5m, d1 = 3m}, Rx {ht = 1.5m, d2 = 2m})
0
With Polariser Grid
-10
Received Power (dBm)
Without Polariser Grid
-20
-30
-40
-50
-60
-70
0.0E+00
5.0E+07
1.0E+08
1.5E+08
Frequency (Hz)
PB/vs/33/Rep5117
2.0E+08
2.5E+08
3.0E+08
36
ERA Report 99-0164
Figure 2.3 Variation of the Received Power measured with and without a
Polariser Grid (wire diameter = 1mm, wire spacing = 100mm, N = 39, width =4m,
height =3m, Tx {ht = 1.5m, d1 = 5m}, Rx {ht = 1.5m, d2 = 3m})
0
-10
Received Power (dBm)
-20
-30
-40
-50
With Polariser Grid
Without Polariser Grid
-60
-70
0.0E+00
5.0E+07
1.0E+08
1.5E+08
Frequency (Hz)
PB/vs/33/Rep5117
2.0E+08
2.5E+08
3.0E+08
37
ERA Report 99-0164
Figure 2.4 Variation of the Received Power measured with and without a Polariser Grid (wire
diameter = 1mm, wire spacing = 100mm, N = 39, width = 4m, height = 3m, Tx {ht = 1.5m, d1 =
7m}, Rx {ht = 1.5m, d2 = 2m})
0
-10
With Polariser Grid
Without Polariser Grid
Received Power (dBm)
-20
-30
-40
-50
-60
-70
0.0E+00
5.0E+07
1.0E+08
1.5E+08
Frequency (Hz)
PB/vs/33/Rep5117
2.0E+08
2.5E+08
3.0E+08
38
ERA Report 99-0164
Figure 2.5 Variation of the Received Power with and without a Polariser Grid (wire
diameter = 1mm, wire spacing = 100mm, N = 39, width = 4m, height = 3m, Tx {ht =
1.5m, d1 = 7m}, Rx {ht = 1.5m, d2 = 3m})
0
Received Power with Polariser Grid
-10
without Polariser Grid
Received Power (dBm)
-20
-30
-40
-50
-60
-70
-80
0.0E+00
5.0E+07
1.0E+08
1.5E+08
Frequency (Hz)
PB/vs/33/Rep5117
2.0E+08
2.5E+08
3.0E+08
39
ERA Report 99-0164
Figure 2.6 Variation of the Horizontally Polarised (wrt Polariser) Received Power
measured with and without a Polariser Grid (wire diameter = 1mm, wire spacing
= 100mm, N = 39,, width = 4m, height = 3m, Tx{ht = 1.5m, d1 = 2m}, Rx {ht =
1.5m, d2 = 2m})
0
Received Power (dBm)
-10
-20
-30
-40
Without Polariser Grid
-50
With Polariser Grid
-60
-70
0.0E+00
5.0E+07
1.0E+08
1.5E+08
Frequency (Hz)
PB/vs/33/Rep5117
2.0E+08
2.5E+08
3.0E+08
40
ERA Report 99-0164
Figure 2.7 Polariser Grid Attenuation - A Comparison between Theory and Experiment
30
25
Theoretical Attenuation
Attenuation (dB)
20
Mean Attenuation
15
Best Fit Attenuation
10
5
0
-5
-10
0
50
100
150
Frequency (MHz)
PB/vs/33/Rep5117
200
250
300
41
ERA Report 99-0164
3
1
2
Figure 3.1: T junction ports
0.2m
4m
Figure 3.2: Straight waveguide (Reference case)
loaded port
ISM machine side
0.2m
45°
0.2m
0.9 m
0.9 m
4m
Figure 3.3: T junction + metallic plates + RAM inside the guide
42
ERA Report 99-0164
43
ERA Report 99-0164
1.6
1.6
0.4
0.4
Metal plate 1
Loaded port
0.2
0.2
Metal plate 2
1.32
4
Figure 3.5: Waveguide geometry (all dimensions in meters)
PB/vs/33/Rep5117
44
ERA Report 99-0164
PB/vs/33/Rep5117
45
ERA Report 99-0164
PB/vs/33/Rep5117
46
ERA Report 99-0164
58
58
58
58
side view
58
29
29
14.5
232
f lange
f lange
plan view
58
Figure 3. 8
PB/vs/33/Rep5117
Test piece geometry (waveguide WG 11A )
47
ERA Report 99-0164
48
ERA Report 99-0164
PB/vs/33/Rep5117
49
ERA Report 99-0164
M
RT
RA
CA
LA
CT
LT
workpiece
Tank Circuit
Figure 4.1
Applicator
Mutual coupling between the tank and appli cator circui ts
50
ERA Report 99-0164
ANA
ref
po rt 1
R
A
B
Pa ral le l
pla te
a ppl ica to r
circu it
po rt 2
s-paramete r test set
3 m lo n g N-Typ e 5 0o hm ca bl e s
Ta nk circu it
T riode
valve
housing
d
dielectric load
launching loop
F igur e 4 .2
PB/vs/33/Rep5117
coupling loop
receiving loop
Exp er ime nta l layo ut for m akin g r etu rn los s an d im ped an ce
m easu rem en ts o f t he d ielec tr ic he ater 's t ank and ap plica tor
cir cu it
51
ERA Report 99-0164
copper strap
earth plate
appl icato r
capaci to r
plate
d
power
fl ap
D
B
conducting
b locks
Tank
cavity
To network
analyser
TRI ODE
C
E
A
Figure. 4. 3 Schem atic diagram of t he
C-Type P lasti c Welder machine showing
the position of t he loop ant enna wit hin
the tank cavit y.
g round
PB/vs/33/Rep5117
52
ERA Report 99-0164
Figure 4.4 Return Loss Spectra measured with the Applicator ON and OFF on the
50MHz C-Type Dielectric Plastic Welder
0
-0.5
-1
Return Loss (dB)
-1.5
-2
-2.5
-3
-3.5
Applicator pressed down
-4
Applicator off
-4.5
-5
50
50.2
50.4
50.6
50.8
51
Frequency (MHz)
PB/vs/33/Rep5117
51.2
51.4
51.6
51.8
52
53
ERA Report 99-0164
Figure 4.5 Frequency change resulting in varing the position of the Power Flap
above the Tank Cavity of the 50MHz C-Type Plastic Welding Dielectric Heater
0
-0.5
-1
Return Loss (dB)
-1.5
-2
-2.5
-3
-3.5
Power Flap raised to maximum setting
-4
Power Flap lowered to minimum setting
-4.5
-5
48
48.5
49
49.5
50
Frequency (MHz)
PB/vs/33/Rep5117
50.5
51
51.5
52
54
ERA Report 99-0164
Figure 4.6 Variation of the Resonant Tank Frequency due to Loading with conducting
Blocks inside the Tank Cavity of the 50MHz C-Type Plastic Welder
0
-0.5
-1
Return Loss (dB)
-1.5
-2
-2.5
-3
Tank cavity loaded with conducting blocks
-3.5
Unloaded Tank Cavity
-4
-4.5
50
51
52
53
Frequency (MHz)
PB/vs/33/Rep5117
54
55
56
55
ERA Report 99-0164
26cm
16cm
t riode
top view
t riode
sliding pl at e
for reduc ing
t he t ank
cavit y v olume
inductor post
ext ernal sliding
c ov er f or inc reas ing
t he t ank volume
top view
Fi gure 4.7 Proposed techni ques to decrease and increase the tank cavity volume
PB/vs/33/Rep5117
56
ERA Report 99-0164
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57
ERA Report 99-0164
Distribution
PB/vs/33/Rep5117
Radiocommunications Agency
(3)
Project Engineer
(1)
Project File
(1)
Information Centre
(1)
58
ERA Report 99-0164
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PB/vs/33/Rep5117