hH
Using the ATF-10236 in Low
Noise Amplifier Applications
in the UHF through 1.7 GHz
Frequency Range
Application Note 1076
Introduction
GaAs FET devices are typically
used in low-noise amplifiers in
the microwave frequency region
where silicon transistors can’t
provide the required gain and
noise performance. There are,
however, many applications in
the frequency range below 2
GHz where the low noise figures
and high gain of GaAs FETs can
improve receiver sensitivity.
Typical applications include low
noise amplifiers (LNAs) in the
800 to 900 MHz frequency range
for use in celluar telephone and
pager applications and spread
spectrum transceiver applications. Additional applications
include the 1228 and 1575 MHz
frequencies used for Global
Positioning System (GPS)
applications. Other applications
include VHF mobile radio,
IMMARSAT, and WEFAX, just
to name a few.
tion. The designs are centered
at 900 MHz and 1575 MHz, but
can be scaled for any frequency
within the region of 400 to 1700
MHz. Each amplifier has a
usable bandwidth of about 30 to
40 percent.
special consideration needs to
be given to the input circuit
design and to the tradeoffs
required to ensure low noise
figure while still achieving
moderate gain, low VSWR and
unconditional stability.
Using a high-gain, high-frequency GaAs FET at VHF poses
special problems. Of greatest
concern is the problem of designing the amplifier for unconditional stability. Typically, GaAs
FETs have greater gain as
frequency is decreased, e.g., 25
dB maximum stable gain at 500
MHz. A second problem is that
matching the typical microwave
GaAs FET at lower frequencies
for minimum noise figure does
not necessarily produce minimum input VSWR.
Device Family
Achieving the lowest possible
noise figure requires matching
the device to Γopt (the source
This application note describes
match required for minimum
two low-noise amplifiers that
noise figure). At higher microuse the Hewlett-Packard ATFwave frequencies this will
10236 low noise GaAs FET
generally produce a reasonable
device. Both designs use identi- input VSWR, since Γopt and the
cal circuit topology with the only complex conjugate of the device
input reflection coefficient S11
differences being in the proper
choice of three inductors depend- are usually close on the Smith
Chart. At lower frequencies,
ing on the frequency of opera-
This application note will
discuss the use of the ATF10236 series of low noise GaAs
FET devices. The device has a
500 micron gate periphery and
is most suitable for applications
in the VHF through 4 GHz
frequency range. The device is
tested for noise figure and gain
at 4 GHz where it is typically
used in satellite TVRO applications. The ATF-10136 is the
premium device being specified
at 0.6 dB maximum noise figure
while the ATF-10236 is specified at a 1.0 dB maximum and
the ATF-10736 is specified at a
1.4 dB maximum all at 4 GHz.
Typically a three stage device
lineup is used at C band with
the ATF-10136 device being
used as the first stage followed
by the ATF-10236 followed by
the ATF-10736. The higher
noise figure of the second and
third stages has minimal effect
on the over all cascade noise
2
figure. A three stage low-noise
amplifier with greater than 30
dB of gain is generally required
at C band to overcome filter/
mixer losses.
While there is considerable
difference in noise figure at 4
GHz between the three devices
(i.e. 0.8 dB), the difference in the
1 to 2 GHz frequency range is
less than several tenths of a dB.
In actual circuits built on low
cost FR-4 dielectric material, the
ATF-10236 device is capable of a
0.5 dB noise figure at 900 MHz
and a 0.75 dB noise figure at
1575 MHz. Most commercial
applications at frequencies below
2 GHz generally do not require
the high gain of a three stage
cascade so generally a single
stage LNA is used followed by a
bipolar device or a silicon MMIC.
Cascading the ATF-10236 with a
3.5 dB noise figure MMIC such
as the MSA-0686, will still result
in about a 1 dB noise figure LNA
with 25 to 30 dB gain.
The Hewlett-Packard ATF-10236
is supplied in the low cost
commercial 0.100 inch “micro-X”
metal/ceramic package. Examination of the data sheet reveals
that the device is capable of a 0.6
dB noise figure at frequencies
below 2 GHz with an associated
gain of greater than 16 dB. The
noise parameters and S-parameters of this transistor are
summarized in Table 1.
Design Technique
Obtaining the lowest possible
noise figure from the device
requires that the input matching
network convert the nominal
50Ω source impedance to Γopt.
This produces a deliberate
impedance mismatch that, while
minimizing amplifier noise
figure, produces a high input
VSWR. The ideal situation is
where Γopt is the complex conjugate of S11 (i.e., S11*). For this
condition, minimum noise figure
is achieved when the device is
matched for minimum VSWR.
This situation occurs predominantly above 2 GHz and tends
to diverge at lower frequencies,
where S11 approaches 1.
High input VSWR has varying
significance, depending on the
application. Most noteworthy is
the increased uncertainty of the
noise figure measurement due to
reflections between the noise
source and amplifier input.
Having a noise source with a
very low output VSWR and one
whose VSWR has minimal
change between the “on” and
“off” states will minimize this
uncertainty. One such noise
source is the Hewlett-Packard
HP346A with a nominal 5 dB
Excess Noise Ratio (ENR).
Similarly, when the amplifier is
connected to a receive antenna,
high input VSWR creates added
uncertainty in overall system
performance. The effect is
difficult to analyze unless an
isolator is placed at the input to
the amplifier. The use of an
isolator, however, adds excessive
loss and, at VHF frequencies, the
size of the isolator is often
prohibitively large.
several authors (References 1-3).
Source feedback, in the form of
source inductance, can improve
input VSWR with minimal noise
figure degradation. The drawback of utilizing source inductance is a gain reduction of up to
several decibels. However, GaAs
FET devices often have more
gain than desired at low frequencies, so the penalty is not severe.
The effect of source inductance
on amplifier input match is best
studied with the help of a computer simulation. The microwave design simulation program
from Hewlett-Packard EEsof
called Touchstone™ is used to
analyze S11 of the amplifier with
the proposed output matching
network. S11 was measured
looking directly into the gate of
the device with the source
inductance added between the
source and ground. With the
AFT-10236 at 500 MHz, adding
the equivalent source inductance
of two 0.10 inch leads causes the
value of S11 to decrease from
0.970 to 0.960. Angle remains
relatively constant at about -20
degrees. Γopt remains relatively
unchanged with the addition of
source inductance. Comparing
S11 to Γopt at 500 MHz now
shows them to be nearly identical. The result is that minimum
noise figure and minimum
VSWR will coincide more closely
To examine the alternatives,
with one another when matching
constant noise figure and conthe device for minimum noise
stant gain circles can be configure. Plotting Γopt for the ATF10236 device from 450 MHz
structed to assess the impact of
trading increased noise figure for through 2 GHz in Figure 1 shows
a decrease in input VSWR and a that Γopt lies very near the
R/Z0=1 curve. This implies that
corresponding increase in
a series inductance will provide
amplifier gain. In most inthe necessary match to attain
stances, the result is a much
minimum noise figure.
higher noise figure than really
desired. One option is to use
The simplest way to incorporate
source feedback. This subject
source inductance is to use the
has already been covered by
3
Table 1. Scattering and Noise Parameters for the Hewlett-Packard ATF-10236 GaAs FET,
Vds = 2 volts and Ids = 25 mA
Scattering Parameters: Common Source, Zo = 50 Ω
Freq
GHz
Mag
S11
Ang
dB
S21
Mag
Ang
dB
S21
Mag
Ang
0.5
1.0
2.0
.97
.93
.77
-20
-41
-81
15.1
14.9
13.6
5.68
5.58
4.76
162
143
107
-32.8
-26.0
-21.3
.023
.050
.086
76
71
51
S22
Mag Ang
.47
.45
.36
-11
-23
-38
Noise Parameters
Frequency
GHz
Noise Figure
dB
Gamma
Mag
Optimum
Ang
Rn/50
normalized
0.5
1.0
2.0
0.45
0.5
0.6
0.93
0.87
0.73
18
36
74
0.75
0.63
0.33
parallel. With the help of
Touchstone™, the effect of the
lead inductance can be analyzed
by simulating the inductance as
a high-impedance transmission
2.0
1.0
device source leads. Using
device leads as inductors produces approximately 1.3 nH per
0.100 inch of source lead, or
0.65 nH for two source leads in
5
0.
2000 MHz
1000
MHz
0.2
2.0
0.5
1.0
500
MHz
0.2
line. The TUNE mode is invaluable for determining the optimum lead length for a given
performance. Table 2 shows the
effect of lead length on gain,
noise figure, stability, and input
and output VSWR at 900 MHz.
It is clear that lead lengths of
0.06 inch or less have a minor
effect on noise figure while
improving input match substantially. Gain does suffer, but this
is not a major concern.
1.0
2.0
0.
5
0.2
Figure 1. ATF-10236 Γo vs. Frequency at Vds = 2 volts and Ids = 25 mA.
An added benefit of using source
inductance is enhanced stability
as evidence by the Rollett stability factor, k. Excessive source
inductance can have an adverse
effect on stability at the higher
frequencies. In the case of the
900 MHz amplifier, zero length
source leads create potential
instability at low frequencies
while longer source lead length
creates potential instability at
high frequencies, i.e. 6.6 GHz. It
is determined that 0.060 inch
source lead length is an optimum
choice based on all parameters.
The optimum source lead length
4
varies with frequency of operation. At 1575 MHz, 0.020 inch
source lead length provides
optimum performance with
unconditional stability up to
12 GHz. Decreasing source lead
length improves stability at
12 GHz while making k<1 at
400 MHz. Adding series resistance in the output matching
circuit increases stability over a
wide frequency range. The effect
of adding a series resistor R10 on
amplifier performance is also
shown in Table 2.
Achieving the associated gain of
which the device is capable is
difficult since the device is not
inherently stable. It is not enough
that the amplifier be stable at the
operating frequency — it must be
stable at all frequencies. Any outof-band oscillation will make the
amplifier unusable.
The simplest technique to ensure
broad-band stability is to resistively load the drain. Resistive
loading produces a constant
impedance on the device over a
wide frequency range. In the
case of the ATF-10236, a 50 Ω
resistor and a small amount of
series inductance is used to load
the output of the device. The
series inductance provides some
impedance matching. This
produces acceptable gain while
ensuring a good output match
and retaining stability over as
wide a bandwidth as possible.
occur from one production run to
another may dictate the need to
change the value of the source
resistor. The calculation to
determine the source resistor RS
is as follows:
RS =
[ Vp (1 –
Id/ Idss ) ]
Id
The amplifier circuit is shown in
Figure 2. For simplicity, the
original LNA used the selfbiasing technique to set the bias
point. The loss in noise figure
associated with the bypassed
source resistor topology is no
greater than 0.1 dB at these
frequencies. The 33 Ω resistor
between the source and ground
sets the drain current while the
100 Ω resistor sets the drain
voltage. There is some interaction, however, between the two
resistors. The supply voltage is
regulated by the TL05 5 volt
regulator U2. The disadvantage
of the self biasing technique is
that variations in Pinchoff
Voltage (Vp) and Saturated
Drain Current (Idss) that may
Assuming typical device
parameters of:
Idss = 130 mA
Vp = -1.3 volts
Id = 25 mA
yields: Rs = 29.2 ohms
This agrees favorably with the
33 Ω resistor used in the actual
amplifier.
The preferred alternative in a
production environment is the
use of an active bias network as
described in Figure 3. Although
active biasing does add cost by
requiring extra components,
including a way of generating a
Table 2. Simulated 900 MHz amplifier performance vs. source lead length , R10 = 0 Ω.
Lead
Length LL1
Noise
Figure
Gain
|S11|2
|S22|2
K at 900
MHz
K at 6.6
GHz
K at 10
GHz
0 inch
0.06 inch
0.2 inch
0.54 dB
0.56 dB
0.62 dB
20.1 dB
18.0 dB
14.4 dB
-4.9 dB
-12.2 dB
-12.7 dB
-8.8 dB
-12.7dB
-18.0 dB
0.58
1.011
1.57
2.94
2.39
0.35
1.65
1.09
1.47
Simulated 900 MHz amplifier performance vs. source lead length , R10 =16 Ω.
Lead
Length LL1
Noise
Figure
Gain
|S11|2
|S22|2
K at 900
MHz
K at 6.6
GHz
K at 7.5
GHz
K at 10
GHz
0 inch
0.06 inch
0.2 inch
0.56 dB
0.58 dB
0.65 dB
18.7 dB
16.6 dB
13.2 dB
-4.6 dB
-11.7 dB
-14.7 dB
-12.0 dB
-13.6 dB
-13.3 dB
0.66
1.22
1.96
4.05
3.36
0.35
3.05
2.50
-0.14
2.15
1.15
1.98
5
Q1
INPUT
C1
C2
L1
Zo
OUTPUT
C3
R10
Zo
Zo
U3
Zo
Zo
Zo
R3
R4
L2
LL1/2
R1
R2
C5
C6
C7
C9
R9
Vcc
U2
C13
COMPONENTS CHANGED WITH FREQUENCY OF AMPLIFIER
FREQUENCY
RFC1
L1
LL1/LL2
900 MHz
0.33 µH
5T #26
0.075" ID
CLOSEWOUND
0.060"
C12
1575 MHz
0.18 µH
2T #28
0.050" ID
CLOSEWOUND
0.020"
COMPONENTS COMMON TO ALL AMPLIFIERS:
R4 = ADJUST FOR DESIRED MMIC CURRENT
C1, C2, C3 = 100 pF CHIP CAPACITOR
R10 = 5 TO 25 Ω CHIP RESISTOR USED TO IMPROVE STABILITY AND OUTPUT RETURN
C5, C6, C7, C9 = 1000 pF CHIP CAPACITOR
LOSS (SEE TEXT)
C12, C13 = 0.1 µF CAPACITOR
U2 = 5 VOLT REGULATOR, TOKO TK11650U
Q1 = HEWLETT-PACKARD ATF-10236 GaAs FET
U3 = HEWLETT-PACKARD MSA-SERIES MMIC
R1 = 100 Ω CHIP RESISTOR
R2 = 27 Ω CHIP RESISTOR
PROVIDES ADDITIONAL GAIN - OPTIONAL
Zo = 50 Ω MICROSTRIPLINE
R3, R9 = 50 Ω CHIP RESISTOR
Figure 2. Schematic of the GaAs FET amplifier using passive biasing. The only change required to modify the
frequency operation is the proper choice of RFC1, L1, and LL1/LL2.
negative voltage, the advantages
generally outweigh the disadvantages. Active biasing offers the
advantage that variations in VP
and Idss won’t necessitate a
change in either the source or
drain resistor value for a given
bias condition. The active bias
network automatically sets Vgs
for the desired drain voltage and
drain current.
The active biasing scheme for
FETs requires that the source
leads be grounded and an additional supply be used to generate
the negative voltage required at
the gate for typical operation.
Direct grounding the FET source
leads has the additional advantage of not requiring bypass
capacitors to bypass a source
resistor that would typically be
5480-2
used for self biasing in a single
supply circuit. When the source
bypass capacitors are removed ,
the source lead length must be
increased as suggested in Figure
3 to offset the decrease in series
inductance. When active biasing
is used, the cold end of the 100
ohm resistor, R1, in the gate
network must be bypassed to
ground with a 1000 pF capacitor
instead of being dc grounded.
This allows gate voltage to be
applied.
Referring to Figure 3, resistors
R6 and R8 provide a regulated
voltage at the base of Q2. The
voltage is increased by 0.7 volts
by virtue of the emitter-base
junction of Q2 and then applied
to the drain of Q1 through
resistor R3. Since R3 is included
in the RF matching circuit, the
voltage drop across R3 must be
taken into account when designing the bias circuitry. Resistor
R9 is connected between two
regulated voltage points and
therefore sets the drain current.
Q1 gate is connected to a voltage
divider consisting of R5 and R7
connected between the collector
of Q2 and a negative voltage
converter. The gate voltage can
then assume a value necessary to
sustain the desired drain voltage
and current as predetermined by
R6, R8, and R9.
Measurements On
Amplifiers
The actual measured performance of the amplifiers compares
favorably to that predicted by the
computer simulation. Table 3
6
Q1
INPUT
C1
C2
L1
Zo
OUTPUT
C3
R10
Zo
Zo
Zo
U3
Zo
Zo
R3
R4
L2
LL1/2
R1
C4
C7
R7
C8
C9
R9
Q2
Vcc
U2
R6
R8
C13
C12
R5
COMPONENTS CHANGED WITH FREQUENCY OF AMPLIFIER
FREQUENCY
RFC1
L1
U1
C10
C11
LL1/LL2
900 MHz
0.33 µH
5T #26
0.075" ID
CLOSEWOUND
0.090"
1575 MHz
0.18 µH
2T #28
0.050" ID
CLOSEWOUND
0.050"
NOTE: LENGTH OF LL1 AND LL2 INCLUDES LENGTH OF
COPPER STRAP USED TO REPLACE C5 AND C6.
COMPONENTS COMMON TO ALL AMPLIFIERS:
R4 = ADJUST FOR DESIRED MMIC CURRENT
C1, C2, C3 = 100 pF CHIP CAPACITOR
R5, R7 = 10K Ω CHIP RESISTOR
C4, C7, C9 = 1000 pF CHIP CAPACITOR
R6 = 1.1K Ω CHIP RESISTOR
C5, C6, = NOT REQUIRED, REPLACE WITH COPPER STRAP
R8 = 1K Ω CHIP RESISTOR
C8, C12, C13 = 0.1 µF CHIP CAPACITOR
R9 = 70 Ω CHIP RESISTOR
C10, C11 = 10 µF CHIP CAPACITOR
R10 = 5 TO 25 Ω CHIP RESISTOR USED TO IMPROVE STABILITY AND OUTPUT RETURN
Q1 = HEWLETT-PACKARD ATF-10236 GaAs FET
LOSS (SEE TEXT)
Q2 = SIEMENS SMBT 2907A PNP TRANSISTOR
U1 = LINEAR TECHNOLOGY LTC1044CS8 VOLTAGE CONVERTER
R1 = 100 Ω CHIP RESISTOR
U2 = 5 VOLT REGULATOR, TOKO TK11650U
R2 = NOT REQUIRED
U3 = HEWLETT-PACKARD MSA-SERIES MMIC
R3 = 50 Ω CHIP RESISTOR
PROVIDES ADDITIONAL GAIN - OPTIONAL
Zo = 50 Ω MICROSTRIPLINE
Figure 3. Schematic of the GaAs FET amplifier using passive biasing. The only change required to modify the
frequency operation is the proper choice of RFC1, L1, and LL1/LL2.
summarizes the gain, noise
figure and VSWR parameters.
The noise figure of the 900 MHz
amplifier is actually a little lower
than the simulation would
predict while the gain is very
comparable. The input return
loss measured 5.6 dB versus 12.2
dB according to the simulation.
The difference between measured and simulated may be due
to the loss of the the RF choke in
the input circuit. The input
VSWR could be improved by
increasing source inductance
while paying careful attention to
stabilty. The last alternative is
to sacrifice some noise figure for
5480-3
sured gain with the output series
resistor R10 measured 12.5 dB
which is within 1.3 dB of the
computer simulation. Gain
increases to 14.3 dB without R10.
Input return loss measured 6.7
The noise figure of the 1575 MHz dB while the output return loss
measured 15 dB which is fairly
amplifier was measured at 0.73
dB which is higher than the 0.45 close to the computer simulation.
Stability is very good with no
dB as predicted. This can be
explained by the dielectric board problems noted when cascading
stages.
losses of the FR-4. The loss of
the FR-4 accounts for 0.3 dB of
The swept gain plots (included in
loss at 1575 MHz. This was
Figures 4-5) show the wide
verified by cutting off the input
bandwidth of these amplifiers.
section of the board , attaching
Low noise figure is also retained
two SMA connectors and careover the bandwidth. The 900
fully measuring the loss. Meaa better input match. Output
return loss measured 13.2 dB
which compares favorably to the
12.7 dB predicted by the computer simulation.
7
Table 3. Measured amplifier performance vs. computer simulation
(* indicates R10 = 16 Ω, ** indicates R10 = 0 Ω)
Noise
Figure
900 MHz
1575 MHz
measured
simulated
measured
simulated
+16.45 dB
+895.00 MHz
20
Gain
0.46 dB
0.56 dB
0.73 dB
16.7 dB
18.0 dB
12.5 dB *
14.3 dB **
0.85 dB
13.8 dB
|S11|2
|S22|2
-5.6 dB
-12.2 dB
-6.7 dB
-13.2 dB
-12.7 dB
-15.0 dB
-8.3 dB
-12.2 dB
10
GAIN (dB)
Frequency
30
0
-10
-20
-30
Using The Design At
Other Frequencies
The basic amplifier design can be
adapted for any frequency in the
400 to 1600 MHz range. Scaling
the input inductor (L1) will allow
operation on a different frequency. The graph shown in
Figure 6 gives some idea of the
relationship of L1 vs. frequency.
Source feedback should be
adjusted accordingly. The ATF10236 has been used successfully
in circuits operating at as low as
150 MHz with similar results.
GaAs FET Demonstration Board
A demonstration board built on
0.031 inch FR-4/G-10 is shown in
Figures 7 and 8. The board as
shown can be set up with either
An additional resistor R10 can
be added in series with capacitor
C2 in order to improve stability
and output return loss of the
first stage. As an example, the
5480-4
use of a 16 Ω resistor at R10 will
decrease gain by about 1 dB
while improving output match
and stability. It may also help
when cascading the FET stage
with the MMIC or another
device.
The measured loss of the input
microstripline excluding the
blocking capacitor C1 is 0.3 dB
at 1600 MHz and 0.15 dB at 900
MHz. Keep this loss in mind
when evaluating the devices as
most computer simulations
appear to be optimistic about
dielectric board losses and
therefore give an optimistic
(lower) prediction of noise figure.
5480-5
100
500
900
1700
1300
FREQUENCY (MHz)
2100
Figure 4. Typical swept gain
performance of 900 MHz amplifier.
30
+14.31 dB
+1.5700 MHz
20
10
GAIN (dB)
At frequencies above 2 GHz, the
ATF-10236 is rated for minimum
noise figure when operated at
Vds of 2 V, and Id of 25 mA. At
frequencies below 2 GHz, it was
empirically determined that an
additional 0.1 dB reduction in
noise figure is possible if the
device is rebiased. At 900 MHz,
1 V gave the lowest noise figure
while 1.5 volts is optimum for
1575 MHz.
passive or active biasing. If
desired an additional gain stage
in the form of a silicon MMIC
such as the Hewlett-Packard
MSA series can be used for
additional gain at a slight
increase in overall cascade noise
figure. A later improved version
of the artwork has the locations
of R1 and L2 reversed. The
mounting pad for R1 and L2 has
an adverse effect on noise figure.
It is therefore better to have the
RF choke first then followed by
the resistor since the impedance
of the choke is higher than that
of the resistor.
0
-10
-20
-30
100
500
900
1700
1300
FREQUENCY (MHz)
2100
Figure 5. Typical swept gain
performance of 1575 MHz amplifier.
150
INDUCTANCE, nH
MHz amplifier has a maximum
of 0.5 dB noise figure between
800 and 1000 MHz. Similarly,
the 1575 MHz amplifier has less
than a 1 dB noise figure from
1200 MHz to 1700 MHz.
100
50
0
0.1
0.5
1.0
1.5
FREQUENCY, GHz
2.0
Figure 6. Inductance L1 vs. optimum
frequency.
8
2.1"
.031 FR-4 GaAsFET DEMO BD
C5
C2
INPUT C1
R1
C6
L2 R2
R3
C8
C7
H
U3 C3 OUTPUT
R4
Vdd
C9
C13
C4
R5
C11
R7
R9
U2
R6
C10
R8
U1
C12
Figure 7. GaAs FET Demo board showing component placement. For best
performance reverse the location of R1 and L2 (see text).
2.1"
.031 FR-4 GaAsFET DEMO BD
C5
INPUT C1
C2
L1
R3
C6
R1
C7 C8
L2 R2
Hhnidqwwertyuiop
U3 C3
R4
Vdd
C9
C13
C4
R5
OUTPUT
C
11 R7
R9
R6
R8
U2
C10
5480-7
U1
C12
Figure 8. GaAs FET Demo board 1X artwork.
In Case Of Difficulty
Generally the noise figure should
be within a couple or three
tenths of a dB of that indicated
in the performance table and
gain within a few dB. If the
noise figure or gain is not as
expected then check the bias
conditions. Is the Vds between 1
and 2 volts and drain current 20
to 25 mA? If the bias voltage
and current is adjustable, does
performance maximize at the
rated bias conditions or at some
other set of values? If so, then
the problem may be excessive
source inductance which is
causing a high frequency oscillation. If a device is oscillating at
any frequency, even an out-of5480-8
the-band frequency, it may be
difficult to obtain rated performance. Shorten the source lead
length or find a source bypass
capacitor with lower parasitic
inductance. The design assumed
0.4 nH of associated inductance
for the 1000 pF source bypass
capacitors. If the problem is
inband stability, then the solution is to add series resistance
between the drain and output
connector. This in series with
blocking capacitor C2. Additional gain reduction can also be
had by decreasing the length of
the shunt output inductor
between R3 and C7. In some
cases, higher than expected noise
figure can be attributed to noise
from the dc to dc converter or the
PNP transistor used for active
biasing. Additional use of 0.1 µF
bypass capacitors at C4 and C8
may be necessary. The use of a
lossy or low Q RF choke for L2
can contribute to an increase in
noise figure. The use of small
molded RF chokes for L2 works
well in prototypes. For surface
mount applications be sure to
choose a high Q wire-wound
choke such as those made by
CoilCraft.
In some instances, the enclosure
can cause undesirable feedback
across the circuit board which
can cause instabilities. This
phenomena is true of any amplifier design. A cross sectional
view of the housing can be
viewed as a piece of waveguide
whose dimensions, both width
and height, determine the band
of frequencies that it may pass
with minimal attenuation. A
combination of the amplifier
response along with the housing
response could contribute to
instabilities if not controlled.
The use of low profile surface
mount components will minimize
this effect. Making sure that the
cover is no closer to the printed
circuit board than is necessary
will minimize coupling from the
cover. It is preferred to have the
cover at least 0.3 to 0.5 inches
above the circuit board. RF
Absorptive material such as
ECOSORB™ can be used on the
cover to minimize reflections if
the cover has to be in close
proximity to the board. The use
of a metal divider hanging down
from the cover is also another
method of breaking up enclosure
effects.
Amplifier Tuning
9
Due to the inherent broad
bandwidth of these amplifiers,
they should require no RF tuning
in production. With the use of
active biasing, the ampifiers
should power up at the proper
bias point without any further
adjustment required. The
frequency at which minimum
input VSWR will occur corresponds automatically with
minimum noise figure and
maximum gain. As shown in the
foregoing data, the noise figure
varies very little over a wide
bandwidth, so it might be advantageous to tune for minimum
input VSWR as opposed to noise
figure. Without the source
inductance, the input VSWR will
be considerably higher.
The use of heavy resistive
loading in the output circuit to
obtain broadband stability can
be expected to limit the power
output capability of the amplifier. Despite the resistive
loading the 900 MHz amplifier
has a measured 1 dB gain
compression point of 5 mW when
biased at a Vds of 1 volt and Ids
of 20 mA.
Operation At Reduced
Current
For applications that are more
current conscious, the 900 MHz
amplifier was tested at various
Table 4. Scattering and Noise Parameters for the Hewlett-Packard
ATF-10236 GaAs FET, Vds = 1 volt and Ids = 10 mA
Freq
Since the noise matching network is low Q it does offer broad
bandwidth and insensitivity to
tolerance on the series inductor.
According to the computer
simulation, varying the input
inductor L1 from its nominal
value of 7 nH to a low of 5 nH to
a high of 10 nH increases the
noise figure less than 0.2 dB. To
minimize cost, the lumped
inductor can be replaced by a
microstripline .010 inches wide
by 0.28 inches long at the expense of 0.2 to 0.3 dB increase in
noise figure. See computer
simulation in the Appendix.
The simple series L/R matching
network in the output circuit
forces a good broadband low
VSWR output match. Due to the
finite amount of reverse isolation
of the device, the output match is
affected by the input match and
vice versa. Therefore the frequency of best output VSWR is
somewhat dependent on where
the input network is optimum.
Power Output
drain currents from 1.5 mA to 25
mA with a constant Vds of 1 volt.
The graph shown in Figure 9
reflects typical performance at
reduced drain current without
readjusting either the input or
output impedance match. The S
Parameters shown in Tables 4, 5,
and 6 can be used to help design
an amplifier for a specific low
current application. The noise
parameters at the nominal bias
(Table 1) will provide a good
starting point for a design at
lower current. The final design
is best optimized on the bench.
S11
S21
S12
S22
GHz
Mag
Ang
Mag
Ang
Mag
Ang
Mag
Ang
0.1
0.5
1.0
2.0
.99
.99
.98
.92
-4
-17
-36
-69
3.66
3.62
3.52
3.25
177
164
147
120
.006
.031
.060
.117
83
80
66
44
.35
.35
.34
.30
-2
-13
-28
-60
Table 5. Scattering and Noise Parameters for the Hewlett-Packard
ATF-10236 GaAs FET, Vds = 1 volt and Ids = 5 mA
Freq
S11
S21
S12
S22
GHz
Mag
Ang
Mag
Ang
Mag
Ang
Mag
Ang
0.1
0.5
1.0
2.0
.99
.99
.99
.95
-3
-15
-32
-62
2.67
2.66
2.63
2.49
178
165
149
123
.004
.033
.068
.128
98
76
68
44
.50
.50
.49
.44
-2
-12
-25
-50
Table 6. Scattering and Noise Parameters for the Hewlett-Packard
ATF-10236 GaAs FET, Vds = 1 volt and Ids = 2 mA
Freq
S11
S21
S12
S22
GHz
Mag
Ang
Mag
Ang
Mag
Ang
Mag
Ang
0.1
0.5
1.0
2.0
.999
.999
.99
.97
-3
-13
-28
-55
1.60
1.60
1.61
1.56
178
167
151
126
.009
.036
.075
.143
76
78
69
48
.68
.67
.66
.62
-3
-11
-22
-43
10
inexpensive epoxy glass dielectric printed circuit board.
2.0
20
GAIN
GAIN (dB)
1.0
10
NF
NOISE FIGURE (dB)
1.5
15
0.5
5
0
0
0
5
10
15
20
DRAIN CURRENT (mA)
25
Figure 9. Noise figure and gain
performance of the 900 MHz LNA at
Vds = 1 volt and various drain
currents.
Other Applications
The basic ATF-10236 amplifier
circuit can also be successfully
used as a mixer for either
upconverting or downconverting
an input signal. As a
downconverter, use the RF input
as the input port and use the RF
output port as the local oscillator
input port. Only a few milliwatts of LO is required at this
port. This is termed a drain
pumped mixer. The IF can be
taken off of the drain bias
decoupling line. The drain bias
decoupling line should be bypassed with a fairly low value of
capacitance such that the normally low frequency of the IF can
still pass through and be coupled
out to the IF port through a low
pass matching network. Mixer
design is covered in more detail
in application note AN-G005.
The single-element input matching network provides very good
performance in this frequency
range and offers the greatest
bandwidth and least sensitivity
to manufacturing tolerances.
The resistive loading in the
output network provides the best
broad-band stable performance
by sacrificing some in-band gain.
The computer simulation programs are instrumental in
analyzing overall amplifier
performance.
References
1. L. Besser, “Stability Considerations of Low Noise Transistor
Amplifiers With Simultaneous
Noise and Power Match,” Low
Noise Microwave Transistors
and Amplifiers, H. Fukui, Editor,
IEEE Press, 1981, pp. 272-274.
2. G.D. Vendelin, “Feedback
Effects on the Noise Performance
of GaAs MESFETs,” Low Noise
Microwave Transistors and
Amplifiers, H. Fukui, Editor,
IEEE Press, 1981, pp.. 294-296.
3. D. Williams, W. Lum, S.
Weinreb, “L-Band Cryogenically
Cooled GaAs FET Amplifiers,”
Microwave Journal, October
1980, p. 73.
4. G. Gonzalez, Microwave
Transistor Amplifiers, PrenticeHall, 1984, pp. 131-133.
5. G.D. Vendelin, Design of
Amplifiers and Oscillators by the
S-Parameter Method, Wiley,
1982, pp. 53-59.
Appendix I.
Low Noise Amplifier for 900 MHz Applications
Using the Hewlett-Packard ATF-10236 Low Noise GaAs FET
Self Biasing Technique
DIM
FREQ GHZ
IND NH
CAP PF
LNG IN
VAR
LL=0.06
CKT
Conclusion
The results show that highfrequency GaAs FETs can be
used very successfully in the
VHF through 1700 MHz frequency range with very simple
circuit techniques. Noise figures
of 0.5 dB at 900 MHz and 0.73
dB at 1575 MHz are possible
using the inexpensive ATF10236 GaAs FET device on an
MSUB ER=4.8 H=.031 T=.0014 RHO=1 RGH=0
MLIN
1
2
W=.060 L=.2
SLC
2
3
L=.25 C=100
MLIN
3
4
W=.06 L=.15
IND
4
5
L\27
MLIN
5
7
W=.02 L=.04
IND
3
6
L=330
RES
6
0
R=100
DEF2P
1
7
NAIN
S2PA
DEF3P
1
1
2
2
3 C:\S_DATA\FET\F102362A.S2P
3 NA2P
11
MLIN
MLIN
SLC
SLC
VIA
VIA
RES
DEF1P
1
1
2
3
4
5
2
1
2
3
4
5
0
0
0
W=.02 L^LL
W=.02 L^LL
L=.4 C=1000
L=.4 C=1000
D1=.03 D2=.03 H=.031 T=.0014
D1=.03 D2=.03 H=.031 T=.0014
R=33
NASER
MLIN
RES
MLIN
SLC
MLIN
SLC
RES
MLIN
DEF2P
1
2
3
4
2
6
7
8
1
2
3
4
0
6
7
8
9
9
W=.060 L=.1
R=50
W=.02 L=.55
L=.4 C=1000
W=.060 L=.1
L=.25 C=100
R=0 !INCREASES STABILITY I.E. 5 TO 25 OHMS
W=.060 L=.1
NAOUT
NAIN
NA2P
NASER
NAOUT
DEF2P
1
2
4
3
1
2
3
4
5
5
AMP
FREQ
!STEP
SWEEP
OUT
AMP
AMP
AMP
AMP
AMP
AMP
AMP
.900
.100
12
.1 GR1
DB[S11]
DB[S21]
DB[S12]
DB[S22] GR1
DB[NF]
K
B1
FREQ
GHz
DB[S11]
AMP
DB[S21]
AMP
DB[S12]
AMP
DB[S22]
AMP
DB[NF]
AMP
K
AMP
B1
AMP
0.10000
0.20000
0.30000
0.40000
0.50000
0.60000
0.70000
0.80000
0.90000
1.00000
1.10000
1.20000
1.30000
1.40000
1.50000
1.60000
1.70000
1.80000
1.90000
-1.838
-0.583
-0.416
-0.500
-0.789
-1.396
-2.734
-5.985
-12.211
-7.497
-3.930
-2.344
-1.553
-1.108
-0.835
-0.656
-0.532
-0.443
-0.377
9.775
10.565
11.235
12.107
13.239
14.593
16.136
17.525
17.959
16.899
14.761
12.597
10.640
8.912
7.386
6.026
4.804
3.694
2.679
-45.329
-42.091
-39.462
-36.918
-34.294
-30.903
-27.619
-24.694
-22.875
-22.667
-23.810
-25.018
-26.056
-26.902
-27.582
-28.130
-28.575
-28.940
-29.243
-12.954
-16.803
-18.601
-20.024
-21.478
-23.388
-21.990
-16.746
-12.677
-10.909
-11.378
-12.085
-12.744
-13.291
-13.723
-14.050
-14.281
-14.421
-14.475
3.145
2.603
2.326
2.090
1.867
1.406
0.976
0.660
0.557
0.752
1.243
1.993
2.913
3.915
4.932
5.922
6.861
7.736
8.540
9.684
2.274
1.182
1.008
1.039
1.029
1.028
1.023
1.011
0.991
1.039
1.067
1.082
1.086
1.084
1.079
1.071
1.063
1.055
1.576
1.838
1.881
1.864
1.802
1.676
1.439
1.061
0.745
0.817
1.121
1.365
1.527
1.632
1.703
1.752
1.787
1.812
1.829
12
FREQ
GHz
DB[S11]
AMP
DB[S21]
AMP
DB[S12]
AMP
DB[S22]
AMP
DB[NF]
AMP
K
AMP
B1
AMP
2.00000
2.10000
2.20000
2.30000
2.40000
2.50000
2.60000
2.70000
2.80000
2.90000
3.00000
3.10000
3.20000
3.30000
3.40000
3.50000
3.60000
3.70000
3.80000
3.90000
4.00000
4.10000
4.20000
4.30000
4.40000
4.50000
4.60000
4.70000
4.80000
4.90000
5.00000
5.10000
5.20000
5.30000
5.40000
5.50000
5.60000
5.70000
5.80000
5.90000
6.00000
6.10000
6.20000
6.30000
6.40000
6.50000
6.60000
6.70000
6.80000
6.90000
7.00000
7.10000
7.20000
7.30000
7.40000
7.50000
7.60000
7.70000
7.80000
-0.327
-0.294
-0.267
-0.244
-0.226
-0.210
-0.197
-0.185
-0.175
-0.165
-0.157
-0.147
-0.139
-0.131
-0.123
-0.116
-0.110
-0.103
-0.098
-0.092
-0.087
-0.082
-0.078
-0.074
-0.070
-0.067
-0.064
-0.062
-0.060
-0.058
-0.057
-0.056
-0.055
-0.054
-0.053
-0.053
-0.053
-0.052
-0.052
-0.052
-0.053
-0.053
-0.054
-0.054
-0.055
-0.056
-0.056
-0.057
-0.058
-0.058
-0.059
-0.059
-0.059
-0.059
-0.058
-0.058
-0.057
-0.056
-0.055
1.741
0.962
0.228
-0.468
-1.131
-1.767
-2.381
-2.977
-3.561
-4.136
-4.704
-5.274
-5.845
-6.417
-6.993
-7.574
-8.159
-8.748
-9.341
-9.936
-10.531
-11.173
-11.807
-12.429
-13.037
-13.629
-14.200
-14.746
-15.262
-15.744
-16.185
-16.628
-17.019
-17.352
-17.619
-17.816
-17.941
-17.992
-17.973
-17.886
-17.737
-17.619
-17.467
-17.288
-17.091
-16.883
-16.674
-16.468
-16.273
-16.093
-15.933
-15.849
-15.794
-15.770
-15.776
-15.811
-15.874
-15.960
-16.068
-29.498
-29.661
-29.803
-29.929
-30.045
-30.156
-30.266
-30.378
-30.497
-30.625
-30.765
-30.975
-31.200
-31.441
-31.701
-31.978
-32.273
-32.585
-32.914
-33.258
-33.614
-33.960
-34.305
-34.645
-34.979
-35.303
-35.613
-35.906
-36.176
-36.419
-36.629
-37.014
-37.322
-37.544
-37.674
-37.708
-37.648
-37.499
-37.267
-36.963
-36.598
-36.353
-36.069
-35.753
-35.414
-35.060
-34.698
-34.337
-33.981
-33.636
-33.307
-33.049
-32.818
-32.614
-32.438
-32.289
-32.164
-32.062
-31.978
-14.445
-14.453
-14.393
-14.266
-14.074
-13.821
-13.515
-13.167
-12.785
-12.382
-11.968
-11.577
-11.192
-10.819
-10.464
-10.133
-9.829
-9.553
-9.307
-9.092
-8.907
-8.782
-8.702
-8.661
-8.660
-8.695
-8.765
-8.870
-9.010
-9.184
-9.394
-9.499
-9.712
-10.034
-10.464
-11.003
-11.650
-12.406
-13.270
-14.242
-15.318
-16.669
-18.276
-20.060
-21.566
-21.763
-20.283
-18.110
-15.992
-14.131
-12.536
-11.182
-10.014
-9.003
-8.128
-7.368
-6.708
-6.136
-5.640
9.269
10.163
11.023
11.848
12.641
13.402
14.134
14.836
15.511
16.157
16.776
17.366
17.926
18.454
18.945
19.397
19.804
20.161
20.462
20.701
20.872
21.366
21.826
22.246
22.621
22.945
23.215
23.428
23.582
23.676
23.715
23.716
23.668
23.576
23.443
23.272
23.066
22.825
22.550
22.241
21.901
22.128
22.293
22.407
22.478
22.513
22.518
22.496
22.453
22.390
22.310
22.219
22.114
21.997
21.869
21.731
21.584
21.429
21.266
1.049
1.046
1.045
1.045
1.048
1.053
1.060
1.070
1.082
1.097
1.114
1.132
1.152
1.175
1.202
1.232
1.267
1.306
1.350
1.399
1.454
1.515
1.584
1.660
1.745
1.838
1.939
2.047
2.160
2.277
2.394
2.573
2.747
2.906
3.037
3.129
3.176
3.172
3.120
3.027
2.900
2.846
2.774
2.688
2.593
2.493
2.391
2.291
2.193
2.100
2.013
1.945
1.883
1.826
1.774
1.727
1.684
1.645
1.609
1.842
1.851
1.858
1.862
1.864
1.864
1.862
1.858
1.852
1.844
1.835
1.826
1.816
1.804
1.792
1.780
1.768
1.756
1.745
1.734
1.725
1.718
1.715
1.713
1.714
1.717
1.722
1.729
1.737
1.747
1.759
1.765
1.776
1.791
1.810
1.831
1.853
1.874
1.895
1.914
1.930
1.946
1.959
1.968
1.974
1.974
1.968
1.956
1.936
1.909
1.875
1.834
1.787
1.735
1.678
1.620
1.560
1.500
1.442
13
FREQ
GHz
DB[S11]
AMP
DB[S21]
AMP
DB[S12]
AMP
DB[S22]
AMP
DB[NF]
AMP
K
AMP
B1
AMP
7.90000
8.00000
8.10000
8.20000
8.30000
8.40000
8.50000
8.60000
8.70000
8.80000
8.90000
9.00000
9.10000
9.20000
9.30000
9.40000
9.50000
9.60000
9.70000
9.80000
9.90000
10.0000
10.1000
10.2000
10.3000
10.4000
10.5000
10.6000
10.7000
10.8000
10.9000
11.0000
11.1000
11.2000
11.3000
11.4000
11.5000
11.6000
11.7000
11.8000
11.9000
12.0000
-0.053
-0.052
-0.050
-0.049
-0.047
-0.046
-0.044
-0.043
-0.042
-0.040
-0.039
-0.038
-0.037
-0.036
-0.035
-0.034
-0.034
-0.033
-0.034
-0.034
-0.036
-0.038
-0.042
-0.047
-0.056
-0.069
-0.089
-0.115
-0.147
-0.176
-0.190
-0.185
-0.167
-0.146
-0.128
-0.112
-0.099
-0.089
-0.080
-0.073
-0.067
-0.062
-16.193
-16.332
-16.443
-16.564
-16.691
-16.820
-16.945
-17.063
-17.169
-17.259
-17.327
-17.369
-17.400
-17.400
-17.366
-17.292
-17.172
-17.001
-16.770
-16.472
-16.096
-15.632
-15.187
-14.651
-14.022
-13.306
-12.534
-11.776
-11.162
-10.869
-11.028
-11.620
-12.394
-13.274
-14.141
-14.944
-15.677
-16.355
-17.007
-17.661
-18.341
-19.064
-31.909
-31.851
-31.876
-31.914
-31.962
-32.015
-32.069
-32.119
-32.162
-32.191
-32.203
-32.192
-32.223
-32.223
-32.188
-32.112
-31.990
-31.815
-31.580
-31.276
-30.895
-30.424
-29.901
-29.286
-28.574
-27.775
-26.918
-26.073
-25.372
-24.990
-25.059
-25.561
-26.403
-27.353
-28.292
-29.171
-29.983
-30.745
-31.487
-32.238
-33.023
-33.861
-5.211
-4.840
-4.466
-4.143
-3.863
-3.621
-3.411
-3.228
-3.069
-2.930
-2.808
-2.700
-2.677
-2.661
-2.651
-2.645
-2.643
-2.644
-2.649
-2.656
-2.668
-2.688
-2.696
-2.720
-2.772
-2.884
-3.114
-3.574
-4.449
-5.998
-8.473
-11.989
-15.470
-16.096
-13.965
-11.622
-9.625
-7.930
-6.474
-5.226
-4.173
-3.308
21.096
20.920
20.937
20.952
20.965
20.976
20.985
20.990
20.992
20.989
20.982
20.968
20.956
20.935
20.905
20.865
20.815
20.754
20.682
20.601
20.511
20.415
20.315
20.218
20.130
20.059
20.018
20.016
20.064
20.172
20.343
20.575
20.854
21.166
21.497
21.835
22.173
22.505
22.830
23.150
23.468
23.787
1.576
1.546
1.515
1.487
1.461
1.436
1.413
1.391
1.369
1.347
1.325
1.302
1.283
1.263
1.241
1.218
1.194
1.169
1.144
1.121
1.101
1.086
1.082
1.086
1.101
1.128
1.171
1.231
1.309
1.405
1.518
1.643
1.766
1.891
2.011
2.119
2.208
2.276
2.320
2.343
2.348
2.339
1.385
1.332
1.273
1.218
1.168
1.121
1.078
1.039
1.004
0.972
0.943
0.917
0.912
0.908
0.905
0.904
0.903
0.904
0.904
0.906
0.908
0.912
0.913
0.917
0.927
0.951
0.999
1.090
1.243
1.452
1.668
1.828
1.903
1.916
1.889
1.836
1.759
1.658
1.532
1.385
1.222
1.056
Low Noise Amplifier for 1575 MHz Applications
Using the Hewlett-Packard ATF-10236 Series Low Noise GaAs FET
Self Biasing Technique
DIM
FREQ
IND
CAP
LNG
GHz
NH
PF
IN
VAR
LL=.02
CKT
MSUB ER=4.8 H=.031 T=.0014 RHO=1 RGH=0
MLIN
1
2
W=.060 L=.2
SLC
2
3
L=.25 C=100
MLIN
3
4
W=.06 L=.15
14
IND
!MLIN
MLIN
IND
RES
DEF2P
4
4
5
3
6
1
5
5
7
6
0
7
L\7
W=.01 L=.28 !OPTIONAL INPUT NOISE MATCH
W=.02 L=.03
L=100
R=100
NAIN
S2PA
DEF3P
1
1
2
2
3 C:\S_DATA\FET\F102362A.S2P
3 NA2P
MLIN
MLIN
SLC
SLC
VIA
VIA
RES
DEF1P
1
1
2
3
4
5
2
1
2
3
4
5
0
0
0
W=.02 L^LL
W=.02 L^LL
L=.4 C=1000
L=.4 C=1000
D1=.03 D2=.03 H=.031 T=.0014
D1=.03 D2=.03 H=.031 T=.0014
R=33
NASER
MLIN
RES
MLIN
SLC
MLIN
SLC
RES
MLIN
DEF2P
1
2
3
4
2
6
7
8
1
2
3
4
0
6
7
8
9
9
W=.060 L=.1
R=50
W=.02 L=.55
L=.4 C=1000
W=.060 L=.1
L=.25 C=100
R=22 !IMPROVES STABILITY I.E. 5 TO 25 OHMS
W=.060 L=.1
NAOUT
NAIN
NA2P
NASER
NAOUT
DEF2P
1
2
4
3
1
2
3
4
5
5
AMP
FREQ
STEP
SWEEP
1.575
.100
12
0.1 GR1
OUT
AMP
AMP
AMP
AMP
AMP
AMP
AMP
DB[S11]
DB[S21]
DB[S12]
DB[S22] GR1
DB[NF]
K
B1
FREQ
GHz
DB[S11]
AMP
DB[S21]
AMP
DB[S12]
AMP
DB[S22]
AMP
DB[NF]
AMP
K
AMP
B1
AMP
0.10000
0.20000
0.30000
0.40000
0.50000
0.60000
0.70000
0.80000
0.90000
1.00000
1.10000
1.20000
-7.200
-3.584
-2.070
-1.394
-1.090
-0.982
-0.995
-1.107
-1.318
-1.649
-2.333
-3.281
5.838
7.134
8.039
8.741
9.376
9.988
10.626
11.299
12.001
12.716
13.255
13.708
-49.265
-45.530
-42.699
-40.388
-38.350
-35.792
-33.511
-31.407
-29.428
-27.555
-26.139
-24.842
-15.944
-20.339
-20.303
-18.855
-17.249
-15.797
-14.503
-13.355
-12.347
-11.482
-10.898
-10.548
4.854
3.891
3.351
3.038
2.832
2.550
2.287
2.037
1.797
1.571
1.363
1.179
58.464
23.125
10.170
5.218
3.118
2.001
1.474
1.200
1.047
0.955
1.002
1.043
1.161
1.425
1.605
1.701
1.740
1.742
1.718
1.671
1.602
1.512
1.383
1.245
15
FREQ
GHz
DB[S11]
AMP
DB[S21]
AMP
DB[S12]
AMP
DB[S22]
AMP
DB[NF]
AMP
K
AMP
B1
AMP
1.30000
1.40000
1.50000
1.57500
1.60000
1.70000
1.80000
1.90000
2.00000
2.10000
2.20000
2.30000
2.40000
2.50000
2.60000
2.70000
2.80000
2.90000
3.00000
3.10000
3.20000
3.30000
3.40000
3.50000
3.60000
3.70000
3.80000
3.90000
4.00000
4.10000
4.20000
4.30000
4.40000
4.50000
4.60000
4.70000
4.80000
4.90000
5.00000
5.10000
5.20000
5.30000
5.40000
5.50000
5.60000
5.70000
5.80000
5.90000
6.00000
6.10000
6.20000
6.30000
6.40000
6.50000
6.60000
6.70000
6.80000
6.90000
7.00000
-4.539
-6.064
-7.568
-8.310
-8.428
-8.226
-7.343
-6.331
-5.438
-4.819
-4.307
-3.886
-3.541
-3.253
-3.010
-2.800
-2.616
-2.449
-2.295
-2.130
-1.973
-1.822
-1.677
-1.538
-1.405
-1.280
-1.162
-1.053
-0.953
-0.849
-0.759
-0.680
-0.611
-0.551
-0.499
-0.455
-0.417
-0.384
-0.356
-0.330
-0.307
-0.288
-0.273
-0.260
-0.251
-0.243
-0.238
-0.235
-0.234
-0.234
-0.235
-0.237
-0.239
-0.241
-0.243
-0.245
-0.245
-0.244
-0.243
14.023
14.149
14.058
13.848
13.753
13.268
12.656
11.965
11.237
10.564
9.896
9.242
8.605
7.987
7.385
6.798
6.222
5.654
5.089
4.538
3.979
3.410
2.829
2.236
1.630
1.013
0.387
-0.247
-0.885
-1.581
-2.270
-2.946
-3.608
-4.249
-4.866
-5.455
-6.011
-6.528
-7.001
-7.492
-7.921
-8.284
-8.575
-8.792
-8.933
-8.999
-8.995
-8.924
-8.797
-8.703
-8.580
-8.436
-8.281
-8.123
-7.971
-7.830
-7.708
-7.607
-7.530
-23.717
-22.809
-22.148
-21.811
-21.727
-21.512
-21.450
-21.490
-21.591
-21.696
-21.814
-21.937
-22.060
-22.181
-22.301
-22.423
-22.548
-22.681
-22.824
-23.029
-23.253
-23.498
-23.766
-24.057
-24.371
-24.706
-25.061
-25.433
-25.818
-26.234
-26.647
-27.052
-27.447
-27.827
-28.189
-28.527
-28.837
-29.114
-29.352
-29.743
-30.053
-30.276
-30.409
-30.451
-30.405
-30.275
-30.070
-29.800
-29.476
-29.257
-29.003
-28.724
-28.429
-28.128
-27.827
-27.534
-27.254
-26.992
-26.751
-10.487
-10.773
-11.448
-12.216
-12.521
-13.967
-15.709
-17.571
-19.158
-19.940
-19.964
-19.389
-18.540
-17.654
-16.845
-16.153
-15.589
-15.151
-14.837
-14.637
-14.565
-14.619
-14.802
-15.115
-15.564
-16.158
-16.912
-17.845
-18.991
-20.202
-21.715
-23.648
-26.229
-29.975
-36.417
-41.416
-32.676
-27.911
-24.846
-22.982
-21.551
-20.378
-19.366
-18.457
-17.626
-16.862
-16.165
-15.543
-15.005
-14.772
-14.633
-14.588
-14.643
-14.802
-15.076
-15.473
-16.008
-16.698
-17.565
1.028
0.920
0.861
0.852
0.857
0.911
1.021
1.184
1.389
1.675
2.015
2.406
2.840
3.312
3.812
4.334
4.870
5.415
5.962
6.505
7.038
7.557
8.054
8.525
8.962
9.358
9.708
10.005
10.242
10.763
11.257
11.717
12.138
12.514
12.840
13.113
13.330
13.492
13.599
13.679
13.710
13.695
13.636
13.535
13.394
13.213
12.996
12.743
12.460
12.655
12.803
12.914
12.996
13.053
13.092
13.114
13.124
13.124
13.115
1.078
1.109
1.137
1.158
1.164
1.191
1.218
1.247
1.278
1.310
1.344
1.381
1.420
1.461
1.504
1.550
1.598
1.648
1.700
1.750
1.801
1.853
1.906
1.959
2.012
2.066
2.119
2.172
2.225
2.269
2.312
2.354
2.393
2.430
2.463
2.491
2.513
2.528
2.536
2.588
2.618
2.623
2.600
2.549
2.474
2.378
2.267
2.148
2.025
1.959
1.890
1.820
1.752
1.685
1.621
1.559
1.501
1.445
1.391
1.115
1.014
0.960
0.954
0.958
0.998
1.061
1.132
1.200
1.252
1.298
1.337
1.369
1.396
1.418
1.438
1.457
1.475
1.493
1.515
1.538
1.564
1.592
1.621
1.651
1.681
1.712
1.741
1.769
1.796
1.821
1.842
1.861
1.877
1.890
1.900
1.908
1.913
1.916
1.919
1.920
1.920
1.918
1.916
1.912
1.907
1.901
1.894
1.887
1.884
1.881
1.880
1.880
1.882
1.885
1.890
1.896
1.903
1.911
hH
FREQ
GHz
DB[S11]
AMP
DB[S21]
AMP
DB[S12]
AMP
DB[S22]
AMP
DB[NF]
AMP
K
AMP
B1
AMP
7.10000
7.20000
7.30000
7.40000
7.50000
7.60000
7.70000
7.80000
7.90000
8.00000
8.10000
8.20000
8.30000
8.40000
8.50000
8.60000
8.70000
8.80000
8.90000
9.00000
9.10000
9.20000
9.30000
9.40000
9.50000
9.60000
9.70000
9.80000
9.90000
10.0000
10.1000
10.2000
10.3000
10.4000
10.5000
10.6000
10.7000
10.8000
10.9000
11.0000
11.1000
11.2000
11.3000
11.4000
11.5000
11.6000
11.7000
11.8000
11.9000
12.0000
-0.238
-0.232
-0.225
-0.216
-0.208
-0.198
-0.188
-0.179
-0.169
-0.160
-0.153
-0.146
-0.139
-0.134
-0.129
-0.125
-0.122
-0.120
-0.119
-0.119
-0.118
-0.119
-0.121
-0.125
-0.130
-0.138
-0.148
-0.160
-0.175
-0.194
-0.214
-0.237
-0.264
-0.296
-0.332
-0.374
-0.421
-0.474
-0.532
-0.593
-0.664
-0.734
-0.801
-0.861
-0.912
-0.951
-0.977
-0.986
-0.974
-0.939
-7.529
-7.559
-7.621
-7.711
-7.829
-7.972
-8.137
-8.320
-8.520
-8.732
-8.889
-9.054
-9.225
-9.398
-9.572
-9.743
-9.910
-10.070
-10.221
-10.359
-10.466
-10.559
-10.637
-10.699
-10.742
-10.765
-10.766
-10.744
-10.696
-10.621
-10.602
-10.557
-10.485
-10.384
-10.254
-10.095
-9.905
-9.688
-9.445
-9.178
-8.760
-8.349
-7.954
-7.584
-7.243
-6.941
-6.690
-6.506
-6.412
-6.434
-26.586
-26.449
-26.340
-26.256
-26.196
-26.157
-26.135
-26.129
-26.134
-26.148
-26.211
-26.285
-26.365
-26.450
-26.536
-26.621
-26.702
-26.777
-26.844
-26.900
-26.995
-27.076
-27.140
-27.188
-27.216
-27.222
-27.206
-27.165
-27.097
-27.001
-26.894
-26.758
-26.591
-26.393
-26.161
-25.896
-25.596
-25.265
-24.902
-24.511
-24.129
-23.754
-23.394
-23.058
-22.750
-22.480
-22.259
-22.104
-22.038
-22.084
-18.743
-20.226
-22.094
-24.419
-27.033
-28.577
-27.202
-24.493
-21.992
-19.928
-18.052
-16.506
-15.205
-14.092
-13.126
-12.276
-11.523
-10.848
-10.239
-9.688
-9.234
-8.819
-8.438
-8.087
-7.763
-7.465
-7.189
-6.936
-6.702
-6.489
-6.292
-6.111
-5.948
-5.801
-5.673
-5.564
-5.477
-5.416
-5.385
-5.391
-5.457
-5.600
-5.846
-6.225
-6.786
-7.594
-8.754
-10.434
-12.928
-16.702
13.108
13.096
13.081
13.063
13.043
13.023
13.002
12.981
12.961
12.943
13.040
13.136
13.228
13.318
13.403
13.484
13.559
13.628
13.689
13.743
13.779
13.805
13.821
13.827
13.822
13.805
13.776
13.734
13.678
13.609
13.547
13.472
13.383
13.281
13.165
13.037
12.899
12.752
12.599
12.444
12.270
12.112
11.971
11.851
11.752
11.675
11.622
11.593
11.592
11.619
1.349
1.309
1.271
1.237
1.204
1.175
1.147
1.123
1.102
1.083
1.071
1.063
1.057
1.056
1.058
1.064
1.075
1.089
1.109
1.133
1.153
1.180
1.213
1.255
1.305
1.364
1.433
1.511
1.601
1.700
1.802
1.910
2.023
2.140
2.257
2.370
2.476
2.570
2.649
2.708
2.739
2.752
2.747
2.727
2.695
2.655
2.611
2.567
2.525
2.488
1.920
1.929
1.937
1.943
1.948
1.951
1.952
1.950
1.947
1.941
1.931
1.919
1.905
1.888
1.869
1.849
1.827
1.804
1.780
1.754
1.731
1.707
1.682
1.657
1.632
1.607
1.582
1.557
1.531
1.506
1.482
1.458
1.435
1.412
1.390
1.369
1.350
1.333
1.320
1.310
1.306
1.313
1.331
1.363
1.410
1.472
1.547
1.629
1.707
1.771
For more information call:
United States*
Europe*
* Call your local HP sales office listed
in your telephone directory. Ask for a
Components representative.
Far East/Australasia: (65) 290-6305
Data Subject to Change
Copyright © 1995 Hewlett-Packard Co.
Canada: (416) 206-4725
Printed in U.S.A. 5963-3780E (3/95)
Japan: (813) 3331-6111