Dmitri VINNIKOV1, Ryszard STRZELECKI2
Tallinn University of Technology (1), Gdynia Maritime University (2)
Comparison of 2- and 3-level Half-Bridge DC/DC Converters
for High-Voltage High-Power Applications
Abstract. This paper is focused on the high-voltage (> 2 kV) high-power (> 20 kW) isolated DC/DC converters. The 3.3 kV IGBT based three-level
half-bridge inverter topology was analyzed as an alternative to the two-level half-bridge with 6.5 kV IGBTs. The properties of primary switches, theirs
selection procedure as well as inverter loss distribution, design challenges, costs of semiconductors and passive components of both concurrent
topologies were evaluated and compared. The overall feasibility of two- and three-level inverter topologies was compared for the selected application
and final recommendations are given.
Streszczenie. Artykuł dotyczy wysokonapięciowych (>2 kV) izolowanych przekształtników DC/DC dużej mocy (>20 kW). Dokonano analizy topologii
półmostkowej, trójpoziomowej z tranzystorami IGBT 3,3 kV, która stanowi alternatywę dla topologii dwupoziomowej z tranzystorami IGBT 6,5 kV.
Zbadano i porównano obydwie topologie pod względem właściwości łączników, metody doboru, rozkładu strat, wyzwań projektowe oraz kosztu
półprzewodników i elementów pasywnych. Porównano wykonalność obydwu rodzajów przekształtników oraz podano końcowe rekomendacje.
(Porównanie dwu- i trójpoziomowych półmostkowych przekształtników DC/DC w aplikacjach dużej mocy i wysokiego napięcia)
Keywords: high-voltage IGBT, DC/DC converter, efficiency, rolling stock.
Słowa kluczowe: wysokonapięciowe IGBT, przekształtnik DC/DC, sprawność, tabor kolejowy.
Introduction
With the latest advancements in power electronic
components and technologies further optimization
possibilities in high-voltage (> 2 kV) high-power (> 20 kW)
IGBT converters became available. For instance, with the
introduction of state-of-the-art 6.5 kV IGBTs, the simple and
reliable two-level voltage-source inverter (VSI) topologies
provide a reasonable choice for the rolling stock auxiliary
power units (APU) with the output power up to several MW.
Investigations have shown, that an experimental converter
based on very simple half-bridge topology with two
200 A/6.5 kV IGBTs is capable of providing an outstanding
performance within the whole range of supply voltage
(2.2...4.0 kV) and load variations [1].
applications a diode-clamped three-level VSI is the most
widespread solution. In a half-bridge configuration only two
additional transistors and two clamping diodes are required
in contrast to the two-level half-bridge counterpart (Fig. 2).
The three-level topology can be easily derived from the twolevel topology with series connected transistors by the
introduction of clamping diodes, which balance out voltage
sharing between series connected top and bottom group
transistors.
Fig. 2. Three-level half-bridge 3.3 kV IGBT based APU
Fig. 1. Two-level half-bridge 6.5 kV IGBT based APU
The most serious problem reported is a high power
dissipation and limited switching frequency of 6.5 kV IGBTs.
According to several researchers [2-5], to reach effective
solutions, multilevel VSI topologies can be implemented.
Three basic topologies have been proposed for multilevel
inverters: diode-clamped (neutral point clamped), capacitorclamped (flying capacitor) and cascaded multicell topology.
All of these three topologies are very popular in such
DC/AC applications as high-voltage AC drives, flexible AC
transmission lines, wind power engineering, etc. In DC/DC
This paper first addresses the three-level half-bridge
diode-clamped topology in a high-voltage high-power
application such as a rolling stock APU. The topology
presented will be comprehensively compared with the twolevel half-bridge topology implemented today. Focus is on
the analysis and discussion of different high-voltage IGBT
properties and features (3.3 kV IGBT for three-level and
6.5 kV for two-level design), inverter switch loss distribution,
passive component ratings as well as some feasibility
problems, like expense of power semiconductors, converter
weight and volume minimization possibilities.
PRZEGLĄD ELEKTROTECHNICZNY (Electrical Review), ISSN 0033-2097, R. 85 NR 10/2009
217
Table 1. Basic data of the reference inverter
3.6 kV DC
0.25
100 kW
Supply voltage of the inverter, UIN
Inverter switch duty ratio, D
Converter rated power, PO
Maximum junction temperature of
semiconductors, Tj,max
0
125 C
In both of the evaluated topologies the rms collector
current of switching transistors will be the same:
(1)
I Crms
P
1
= O ⋅
= 55.5 (A).
U IN
D
The secondary part (isolation transformer and rectifierfilter stack) of the reference converter is represented by the
equivalent load resistance Rekv (represents the secondary
part of the converter referred to the primary), the value of
which can be obtained from (2).
(2)
Rekv =
2
U TRrms
,
PO
where: UTRrms - rms voltage of the isolation transformer, PO converter output power. The rms voltage of the isolation
transformer can be defined by (3).
(3)
U TRrms = U TR 2 ⋅ D ,
where: UTR - amplitude value of the isolation transformer
voltage. In the analyzed operation point the equivalent load
resistance is 16.2 Ω.
Comparison of semiconductor properties and losses
Semiconductor losses are a central evaluation criterion
for a topology due to the direct correlation with virtually all
other electrical parameters of the converter [6]. As it was
mentioned before, the switching losses of high-voltage
IGBTs are one of the limiting factors during power converter
design with such devices. It finally leads to more powerful
and complicated heatsinks to be implemented for ensuring
the proper junction temperature of semiconductors or
limiting of the switching frequency. Both of these measures
result in a negative impact on space-weight parameters,
which are essential during the design of modern power
electronic converters.
Comparison of different high-voltage IGBTs
In the application discussed the switching devices in the
three-level half-bridge configuration require half the blocking
voltage of that in a two-level half-bridge, i.e. instead of the
single 6.5 kV IGBT two series connected 3.3 kV IGBT
transistors with the same collector current rating could be
implemented with confidence. To provide a better
comparability, the high-voltage IGBTs from one vendor
(Eupec-Infineon) are compared with the datasheet values
available from the Internet. The analysis below is based on
the following IGBTs:
• Eupec
3.3 kV/200 A
dual
IGBT
module
FF200R33KF2C,
218
•
Infineon 6.5 kV/200 A single IGBT module
FZ200R65KF1.
As shown in Table 2, the switching loss energies (Eon,
Eoff, Erec) of 3.3 kV IGBT transistors for the same current are
2-5 times smaller than those of 6.5 kV IGBTs. Theoretically
it means that a series connection of two 3.3 kV IGBTs could
provide a 30-40 % reduction of switching losses as
compared to a single 6.5 kV IGBT module for the same
switching frequency and transferred power of the inverter.
On-state voltage drop UCEsat of a 3.3 kV IGBT makes about
80% from that of 6.5 kV counterparts, but the resulting
voltage drop of two 3.3 kV modules connected in series will
be 62% higher.
Table 2. Typical values of different investigated IGBTs
IGBT type
Conditions
UCE=1800 V,
IC=200 A,
Tj=125 0C
UCE=3600 V,
IC=200 A,
Tj=125 0C
3.3 kV/200 A
FF200R33KF2C
6.5 kV/200 A
FZ200R65KF1
Eon,
mJ
Eoff,
mJ
Erec,
mJ
UCEsat,
V
365
255
255
4.3
1900
1200
550
5.3
Comparison of the high-voltage (HV) IGBT switching
properties investigated (Fig. 3) shows, that the dynamics of
the 6.5 kV IGBT is considerably decreased than that of the
3.3 kV counterpart, especially in terms of turn-off delay time
(td,off). The simplified method of the evaluation of the
maximum switching frequency of IGBT in hard switching
applications shows that the series connection of 3.3 kV
IGBTs could theoretically provide more than 2 times higher
switching frequency for the same transferred power as
compared to 6.5 kV IGBTs.
f max,lim =
(4)
t d ,on
a
,
+ t r + t d ,off +t f
where: a - switching period limitation on the total switching
time (conventionally 5% of the switching period), td,on - turnon delay time, td,off - turn-off delay time, tr - current rise time
and tf - current fall time (last four values are always
presented in the datasheets).
6
6.0
5
Time [us]
Definition of evaluation criteria
In both of the topologies to be analyzed (Figs. 1 and 2),
the isolation transformer TX sees only the half of input
voltage UIN. Two equal capacitors C1 and C2 are connected
in series across the DC input voltage source, providing a
constant potential of one-half UIN at their junction. To
simplify further discussion it was assumed that the isolation
transformer and the rectifier-filter assemblies of both
topologies are identical and lossless. Table 1 shows the
basic data of the reference converter and conditions for the
comparison.
4
3
1.7
2
1
0
0.28
0.72
tdon
td,on
0.2 0.4
ttrr
0.2
tdoff
td,off
0.5
tftf
FF200R33KF2C (Uce=1800V, Ic=200A, Tj=125 °C)
FZ200R65KF1 (Uce=3600V, Ic=200A, Tj=125 °C)
Fig. 3. Side-by-side comparison of switching properties of different
investigated IGBTs
Derivation of upper switching frequency
Fig. 4 interprets the practical switching frequency limits,
which can be achieved by each HV IGBT technology in
hard switching mode and in the discussed operation
conditions (see Table 1). The rms collector current in given
application is 55.5 A (28% from the modules’ nominal). As it
was predicted in the previous section, the three-level halfbridge topology with 3.3 kV IGBTs has a strong benefit in
terms of doubling the switching frequency in contrast to the
6.5 kV IGBT based two-level counterpart.
PRZEGLĄD ELEKTROTECHNICZNY (Electrical Review), ISSN 0033-2097, R. 85 NR 10/2009
In practice, the switching frequency limit fup of IGBT is
mostly determined by the thermal management system of
the IGBT module. The thermal limit to frequency was
derived by (5).
f up
(5)
TJC
− Pcond
Z thJC
,
=
Eon + Eoff
Total inverter losses [W]
125
100
75
2150 Hz
55A
50
1050 Hz
25
1600
1200
800
400
0
Three-level inverter with
FF200R33KF2C (Uce=1800V,
Ic=55.5A, fsw=2.15kHz)
Two-level inverter with
FZ200R65KF1(Uce=3600V,
Ic=55.5A, fsw=1.05kHz)
Pdyn,FWD [W]
507,2
267,2
Pstat,FWD [W]
1,6
1,2
Pdyn,IGBT [W]
666,0
813,2
Pstat,IGBT [W]
272,0
180,0
0
500
1000
1500
2000
2500
3000
Switching frequency [Hz]
FF200R33KF2C (Uce=1800V, Tj=125 °C)
FZ200R65KF1(Uce=3600V, Tj=125 °C)
Fig. 4. Side-by-side comparison of switching properties of different
investigated IGBTs
Average switch losses [W]
Analysis of inverter losses
As a result of the analysis of average switch losses in
two- and three-level topologies for the discussed operation
point and with the maximum allowable switching frequency
it was found that those in three-level topology with the
3.3 kV IGBTs are 43% smaller than in the two-level design
(362 W vs. 631 W). As shown in Fig. 5, a huge part of the
power dissipation of the 6.5 kV IGBT module (more than
60%) is formed by the dynamic losses of the IGBT. In the
case of 3.3 kV IGBT modules the dynamic losses of the
IGBT transistor and the integrated freewheeling diode
(FWD) are almost equalized.
Fig. 6. Breakdown of total inverter losses in different inverter
configurations and with different HV IGBTs
Total inverter losses [W]
Maximum collector rms
current [A]
where: TJC - junction to case temperature, ZthJC - junction to
case thermal impedance, Pcond - conduction power loss, Eon
and Eoff - switching loss energies.
It should be pointed out that the thermal limitation
should be considered only for the IGBT transistors, whereas
no thermal limitations exist with regard to the freewheeling
diodes for both investigated modules.
resulting power dissipation of the inverter will be 14% higher
than in the two-level design (Fig. 6). Such differences in
losses are mostly caused by the IGBT static (conduction)
losses, which in the case of series connected 3.3 kV is 51%
higher than in a single 6.5 kV IGBT. The dynamic losses
content is almost on the same level in both of the
investigated topologies.
Otherwise, for the same switching frequency as in a
two-level inverter (1.05 kHz) the three-level topology could
markedly raise the efficiency, which is mostly caused by the
33% decreased total inverter power dissipation (Fig. 7), as it
was predicted in the previous section. While the switching
loss content was considerably reduced with the frequency,
the percentage of IGBT conduction losses was increased
and formed more than 30% from the total power dissipation
of three-level inverter topology.
1600
1200
800
400
0
Three-level inverter with
FF200R33KF2C (Uce=1800V,
Ic=55.5A, fsw=1.05kHz)
Two-level inverter with
FZ200R65KF1(Uce=3600V,
Ic=55.5A, fsw=1.05kHz)
Pdyn,FWD [W]
247,6
267,2
Pstat,FWD [W]
1,6
1,2
Pdyn,IGBT [W]
325,2
813,2
Pstat,IGBT [W]
272,0
180,0
800
Fig. 7. Breakdown of total inverter losses for the same switching
frequency in different inverter configurations and with different HV
IGBTs
600
400
200
0
FF200R33KF2C (Uce=1800V,
Ic=55.5A, fsw=2.15kHz)
FZ200R65KF1(Uce=3600V,
Ic=55.5A, fsw=1.05kHz)
Pdyn,FWD [W]
126,8
133,6
Pstat,FWD [W]
0,4
0,6
Pdyn,IGBT [W]
166,5
406,6
Pstat,IGBT [W]
68,0
90,0
Fig. 5. Breakdown of single switch losses in different inverter
configurations and with different HV IGBTs
Due to better switching properties, the three-level
inverters with 3.3 kV IGBT can operate with at least doubled
switching frequency in contrast to the 6.5 kV IGBT based
two-level converters. However, despite the 43% smaller
per-switch power dissipation of the three-level topology, the
Comparison of passive components
As previous analysis has shown, the three-level
topology in the current application ensures either a twofold
increased switching frequency or 33% reduced total inverter
power dissipation. From the designer’s point of view any
improvements in the switching frequency lead to the
minimization of passive components, like filter capacitors
and inductors as well as isolation transformers. The most
critical passive components in the given application are the
half-bridge capacitors (C1 and C2) and the isolation
transformer TX.
Half-bridge capacitors have to withstand high voltage
and current ripple and in the current design could occupy
more than 25% of the total converter volume, which can
change proportional to the switching frequency. The value
of the required capacitance is calculated from the known
primary current and operating frequency:
PRZEGLĄD ELEKTROTECHNICZNY (Electrical Review), ISSN 0033-2097, R. 85 NR 10/2009
219
I IN ⋅ Δt
P
=
,
2
ΔU C
2 ⋅ U IN ⋅ f sw ⋅ ΔU C
Toroidal Core GM14DC
Temperature rise Δ T=50 °C; Operating Flux Density BAC=0.4T; Window
Utilization KU =0.25
where: IIN - input current, P - inverter total power, UIN - input
voltage, ΔUC - capacitor voltage ripple and fsw - switching
frequency.
Fig. 8 presents the comparison of minimum capacitance
required for the half-bridge capacitors C1 and C2 estimated
for different voltage ripple factors. With the switching
frequency rising, the required capacitance value is
decreasing exponentially. The result is that the three-level
half-bridge inverter operated with doubled switching
frequency features 50% smaller input capacitors than the
two-level topology (Fig. 9). The same weight-volume
optimization progress could be achieved with the output
filter components as well.
Required capacitance
for C1 and C2 [uF]
400
300
200
100
0
1000
1500
2000
2500
3000
Switching frequency [Hz]
Ripple Uc=1%
Ripple Uc=2%
Ripple Uc=4%
Fig. 8. Comparison of capacitance requirements for the switching
frequency range of 1000...3000 Hz and for different ripple factors
20000
3
and C2 [cm ]
Volume of capacitors C1
Volumes are estimated for high voltage capacitor with organic film dielectric
ELKOD K75-80-4kV-100uF
15000
10000
5000
0
1000
1500
2000
2500
3000
Switching frequency [Hz]
Ripple Uc=1%
Ripple Uc=2%
Ripple Uc=4%
Fig. 9. Comparison of half-bridge capacitor volumes for the
switching frequency range of 1000...3000 Hz and for different ripple
factors
Isolation transformer TX is another bulky component in
voltage converters with such power range. The pulse
transformer core volume Vm should be selected to meet the
power requirements and temperature rise for the selected
operating frequency:
(7)
Vm = 1.5
A ⋅ k ADD ⋅ kT
⋅
kU
P
1
4
,
f sw ⋅ ΔT
where: P - transformer rated power, kT - temperature
coefficient of winding resistance, kU - window utilization
factor, kADD - added loss factor (increase of winding
resistance with frequency due to skin and proximity effects),
ΔT - transformer temperature rise. Increasing the switching
frequency the magnetic core volume could be effectively
reduced by a factor of 1.8 (Fig. 10).
220
Required core volume
3
3
[cm (x10 )]
C=
(6)
15
12
9
6
3
0
1000
1500
2000
2500
3000
Switching frequency [Hz]
Fig. 10. Minimal required core volume as a function of operating
frequency for the 100 kVA isolation transformer with the GM14DC
magnetic core
Comparison of overall feasibility
Packaging
Despite the increased number of transistors in the threelevel topology the resulting installation area requirements
remain the same (Table 3). Thanks to the dual-transistor
modules available for 3.3 kV IGBTs the space-weight
constraints imposed by the two-level topology will not be
broken. Moreover, due to the 20% reduced height of 3.3 kV
IGBT modules the overall height of the inverter stack could
be decreased.
Table 3. Comparison of physical parameters of the investigated
IGBTs
IGBT type
3.3 kV/200 A
FF200R33KF2C
6.5 kV/200 A
FZ200R65KF1
Weight,
g
Length,
mm
Height,
mm
Width,
mm
500
140
38
73
500
140
48
73
It is evident that the clamping diodes Dcl1 and Dcl2 are
additional components in three-level topology. Although
they must withstand high voltage during the positive and
negative freewheeling modes, when the freewheeling path
for the magnetizing current of the isolation transformer is
created by Dcl1 and T2, and Dcl2 and T3, respectively.
However, the operating current in these modes is relatively
low, so the fast recovery diode modules in ISOTOP
packages (38 x 25 mm) could be implemented with
confidence and with no serious impact on the inverter’s
dimensions and weight.
The total size and weight of the converter with the threelevel inverter topology could be decreased progressively
either by the minimized passive components at maximum
switching frequency or by the 30% minimized heatsink
requirements in the case of the switching frequency parity
for both topologies.
Control and protection
Although the number of controlled IGBTs is increased
by a factor of two, the complexity of the control algorithm
and the control system as a whole will not change
significantly. The control signals for two additional IGBTs
can be derived within the hardware simply by inverting two
present PWM signals by the logic element (74HC04 Hex
inverter). The only difference is that in contrast to the twolevel topology, the control system should process twice the
number of error feedbacks from power transistors.
Control of dual-IGBT modules can be performed by a
single driver core (for example, 2SD315AI-33 from CTConcept), which gives additional benefits in terms of
reliability and system complexity.
PRZEGLĄD ELEKTROTECHNICZNY (Electrical Review), ISSN 0033-2097, R. 85 NR 10/2009
The number and position of sensors remains unchanged
(input and output voltages and currents, central point
asymmetry, etc.).
Semiconductor price evaluation
Semiconductor price, in particular in high-voltage
applications, is one of the essential aspects from the
designer’s point of view, because the competitiveness of
the device produced is always price-dependent. In the case
of high-voltage IGBTs the specially designed dedicated
gate drives (plug-and-play drivers) should be used. The
price of high-voltage semiconductors due to their relative
novelty and recently high manufacturing costs is relatively
high and the feasibility of both investigated topologies
seems to be a front row question.
Fig. 11 shows a comparison of total prices for both of
the investigated inverter topologies. As compared to a twolevel topology, the three-level topology with 3.3 kV IGBTs
has a remarkable benefit of over 40% inverter price
reduction.
Total price evaluation [%]
100
80
60
100 %
40
53 %
20
0
2-level with 6.5 kV IGBTs
3-level with 3.3 kV IGBTs
Investigated topologies
which in particular kills all the presented benefits of these
transistors.
The obvious solution to a problem is to use the lower
voltage IGBTs in series connection to provide the same
voltage blocking capability as in the case of 6.5 kV IGBT.
The most technically feasible design is to use in series
connection two dual 3.3 kV IGBT modules with the same
housings as 6.5 kV IGBT (IHV 73 mm). Further, by some
small modifications (introduction of two additional clamping
diodes) the three-level topology could be derived from the
two-level one. Clamping diodes provides the proper voltage
sharing between the series-connected transistors thus
optimizing their operation conditions.
The analysis and comparison presented in the paper
shows that 3.3 kV IGBT based three-level diode-clamped
inverter topology is a very attractive alternative to a recently
popular two-level 6.5 kV IGBT based inverter configuration.
For the same transferred power and switching frequency
the three-level topology features more than 30%
minimization of power losses. Otherwise, for the same loss
level the IGBTs in the three-level topology could be
operated with a double switching frequency, thus achieving
strong progress in the minimization of passive components
(input and output filters, isolation transformer).
From the viewpoint of overall feasibility, by the
implementation of the three-level topology instead of the
two-level the inverter component costs can be reduced by
50%, which is especially relevant in the design of highvoltage high-power DC/DC converters.
Authors thank Estonian Science Foundation (Grant
ETF7425 “Research of Dynamic Performance of HighVoltage IGBTs”) for financial support of this study.
Fig. 11. Comparison of semiconductor price of two- and three-level
topologies
Such a huge difference is mostly caused by the over
50% price difference between 6.5 kV and 3.3 kV IGBT
modules with a collector current of 200 A. As the analysis
shows (Fig. 12), namely the price of IGBTs accounts for
over 70% of the total inverter price in such high-voltage
applications and is the final price determinative factor. Even
if two additional clamping diodes are used in the three-level
topology, no serious impact on the overall competibility of
the proposed solution is achieved.
IGBTs
81%
Drivers
19%
(a)
Drivers
23%
Clamping
diodes
5%
IGBTs
72%
(b)
Fig. 12. Distribution of basic component prices in the two-level
topology with 6.5 kV IGBTs (a) and three-level topology with
3.3. kV IGBTs (b)
Conclusions
In terms of recent developments, industry and traction
demand higher power equipment, which has already
crossed a megawatt level. The implementation of highvoltage IGBTs (6.5 kV) in high-power converters gives an
attractive possibility of using simple and robust two-level
inverter topologies for the input DC voltages up to 4.2 kV.
Such inverters, especially as an integral part of isolated
DC/DC converters, are very simple in control and
protection, have reduced component count and provide
better reliability. But the determinative factors here are the
low switching dynamics and high price of 6.5 kV IGBTs,
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Authors: Dr. Sc. techn. Dmitri Vinnikov, Senior Researcher,
Tallinn University of Technology, Department of Electrical Drives
and Power Electronics, Ehitajate str. 5, 19086 Tallinn, Estonia,
E-mail: dm.vin@mail.ee; D.Sc. Ryszard Strzelecki, Professor,
Department of Ship Automation, Gdynia Maritime University, 81-87
Morska Str., 81-225 Gdynia,Poland, E-mail: rstrzele@am.gdynia.pl.
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