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This paper demonstrates the possibility of performing radio frequency (RF) measurements in the IEEE 1149.4 environment using a selected combination of mixing and sub-sampling as the method to down-convert the RF signal into a low frequency suitable for the analogue busses specified by the 1149.4 standard. Signals generated by RF signal generators (2 GHz) and a commercial VCO (3 GHz) were used to demonstrate the operation of the developed RF-to-LF circuitry in cooperation with National’s SCANSTA400 IEEE 1149.4
Analog Test Access Device. The measured repeatability of RF power measurement through the whole system, i.e. from the RF input to the output of SCANSTA400, is ±0.15 dB and the measurement uncertainty of the RF frequency due to measurement instruments and signal sources is
±350 Hz.
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below RF signals). However, if the standard could be used for RF testing, the speed (e.g. BIST) and cost (lowfrequency low-cost equipment may be used) improvements could be considerable. In addition, this would allow the use of standard (1149.4) test equipment available in the future.
In order to allow the measurement of RF signals, some kind of signal processing which extracts the information one is measuring from the RF signal and to present the result as a low-frequency signal (i.e. below 100 kHz) further processed by structures specified in the standard
(1149.4) must be used.
As the first step towards the application of the IEEE
1149.4 architecture for RF testing, embedded RF test circuitry inside RF ASICs and low-cost measurement instrument are used to replace the conventional and expensive RF testing equipment. Whether this is cost effective when the whole test system is considered depends on integration costs. However, at least RF interfacing and probing problems can be avoided during testing.
Reliable, accurate, fast and low-cost testing of products is one of the corner stones of modern electronics industry.
Here the term testing covers production testing, calibration, tuning and maintenance testing. Especially the radio frequency (RF) circuitry presents a tough challenge in the testing of electrical products (e.g. cellular phones, base stations). Testing and tuning of high-frequency parts of electrical systems is extremely time-consuming and requires special, often expensive, test equipment. Furthermore, RF testing will continue to be a major challenge in the future because of new standards (e.g. Bluetooth and IEEE 802.11) and multiband/multi-function products.
The emerging testing standard IEEE 1149.4 [1] is mainly targeted for low-frequency testing, wherefore it is not directly applicable to the testing of high-frequency circuitry. The key problem here is the limited frequency range accommodated by the standard (several decades
In the future cost-effective Life-Cycle-Testing (LCT) with high fault coverage and advanced diagnostics may be achieved with totally embedded testing, which provides test reusability during the entire product life cycle. The embedded test architecture may be based on the IEEE 1149.4 architecture with added RF capability, which provides integrated access, stimulus generation and measurement instruments (BIST, Build-In Self-
Test).
In developing the embedded RF testing the first step is to investigate the possible frequency down-conversion techniques in cooperation with the 1149.4 low-frequency architecture. The accuracy of the measurement of the translated RF signal property and complexity of the required circuitry are the principal issues when evaluating the usability of different RF test structures for further use, e.g. for ASIC realizations.
Therefore, the specific aim of this project was to demonstrate that it is

f i
(t) f c f o
(t) f i
(t) f o
(t)
T
τ t
τ
T t
Figure 1 The sub-sampling process possible to perform simple high-frequency measurements by suitable RF analogue circuit structures (i.e. RF-to-LF signal processing) in cooperation with the IEEE 1149.4 specific structures from target RF frequencies from 2 to 3
GHz.
There are several ways to translate a high-frequency signal down to low frequencies, out of which frequency mixing [2] and sub-sampling [3] are the ones most often employed. In the design reported here a combination of these methods was used. Mixing alone would certainly work for us, but the combination of the two adds freedom in system design. Additionally, sub-sampling from intermediate frequencies may even be directly performed by utilizing the circuit structures specified in the standard as demonstrated in the literature [4]. However, subsampling alone is not conveniently done from frequencies above 1 GHz. Therefore, linear mixing was used to shift the input signal to the 100 MHz range, where sampling is more convenient as far as noise aliasing and the speed of
|F(
ω
)| |G(
ω
)| f c
2f c
3f c f c
=nf c f
RF
+
∆ f f
∆ f ∆ f
∆ f
∆ f
∆ f
∆ f
∆ f
Figure 2 Sub-sampling in the frequency domain f the sampler are considered. Due to its familiarity, frequency mixing is not treated theoretically here (e.g.
[2]).
2.1 Sub-sampling
( ≈ f vco
2 GHz )
LRPS-2-25J
( ≈ f
LO
1.9 GHz )
≈ band
2 GHz
pass pulse shaper
T
( ≈ f
IF
100 MHz
3 dB
)
≈ 4 band
GHz
stop
φ
S/H
φ
VCOOUT
(to SCANSTA 400)
( ≈ f out
50 kHz ) f clk
( ≈ 9.995 MHz)
Figure 3 Structure of the RF-to-LF processing unit
Figure 4 The RF-to-LF measurement unit
If a signal is sampled by a rate, which is less than the frequency of the signal, the signal is said to be subsampled [3]. In this case, if the signal bandwidth is small, a copy of the original signal is produced at low frequencies by the well-known process of aliasing. The sub-sampling process is shown in Fig. 1 and can be presented mathematically as [3]
G (
ω
)
= dF (
ω
)
+ d n n
=
≠
∞
−∞
0
(sin( n
π d )) /( n
π d ) F (
ω − n
ω c
)
, (1) where G(
ω
) = Fourier transform of
Fourier transform of pulse, T = 1/ f c f i f o
(t) , d =
τ
/T , F(
ω
) =
(t) ,
τ
= width of the sampling
= time between samples and
ω c
= 2
π
/ T .
The sub-sampling process can now be straightforwardly presented in the frequency domain using equation (1). As shown in Fig. 2, a copy of the narrowband signal at f
RF
is produced around multiples of the sampling frequency.
Now, a low-pass filter, for example, may be used to extract only the desired harmonic for further processing.
2.2 Realization
The structure of the RF-to-LF processing unit is shown in
Fig. 3 and the actual in-a-box realization in Fig. 4. To allow reference measurements by traditional RF measuring equipment, the input RF signal (f vco
) is first split into two by a power splitter (LRPS-2-25J [5]). The input signal is then converted to an intermediate frequency by a mixer (MBA-15L [6]) and finally subsampled to the proper output frequency by a S/H circuit
(constructed around MAX4619 [7]). The recommended frequencies for the different signals in Fig. 3 when f vco
≈
110
100
90
80
1
0
-1 fclk=9.985050 MHz fclk=9.988390 MHz fclk=9.991011 MHz
70 -2
60
-3
50
40 fclk=9.985050 MHz fclk=9.988390 MHz fclk=9.991011 MHz
-4
30 -5
2 2.01
2.02 2.03 2.04
2.05 2.06
2.07 2.08
2.09
2.1
f vco
/GHz
(a)
2 2.02
2.04
2.06
f vco
/GHz
(b)
2.08
2.1
Figure 5 (a) Measured fout vs. fvco and (b) measurement error

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-34
-35
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-38
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-40
-41
2.0GHz
2.02GHz
2.04GHz
2.06GHz
2.08GHz
2.1GHz
2.01GHz
2.03GHz
2.05GHz
2.07GHz
2.09GHz
0
-10
-20
-30
-40
-50
-60
-70
-2 -1 0 1 2 3 -80
0 10000 20000 30000
P in
/dBm
(a) fout/Hz
(b)
Figure 6 (a) P out
vs. P in
and (b) P out
vs. f out
40000 fmix=1.9GHz
fmix=1.95GHz
fmix=1.99GHz
50000 60000
2.0 GHz are: f
LO
≈
1.9 GHz
⇒
f
IF
≈
9.995 MHz
⇒
f out
≈
100 MHz, f clk
≈
50 kHz (i.e. suitable for structures found in National’s SCANSTA400 IEEE 1149.4 Analog
Test Access Device [8]).
3.1 Characterization
The RF-to-LF unit was characterized by measuring the output signal of a laboratory RF signal generator
(Marconi Instruments 2024). Agilent 331220A was used as the LO generator and HP 4396A Spectrum analyzer to measure the reference and the LF output.
The measured final LF output frequency as the function of the input frequency ( f vco
) is shown in Fig. 5 (a). Given y = 70.895x - 25.471, Tuning sensitivity is 70.895 kHz/V
110
100
90
80
70
60
50
40
30
20
10
0.5
measured
LMS
0.7
0.9
1.1
Vtune/V
1.3
1.5
1.7
1.9
(a)
Tuning linearity (non-linearity is +/-0.05543kHz).
0.06
0.05
0.04
0.03
0.02
0.01
0
-0.01
-0.02
-0.03
-0.04
0.5
0.7
0.9
1.1
1.3
1.5
1.7
1.9
Vtune/V
(b)
Figure 7 (a) VCO frequency tuning characteristic and (b) tuning linearity the signals shown in Fig. 3, the output frequency can be calculated using the following equation:
f out
= (f vco
– f
LO
) – nf clk
, where n = 10,11,…20 (2)
Equation (2) can also be used to calculate the original RF input frequency from the measured LF output frequency.
Thus, the calculated error between the reference measurement and the calculated output frequency is shown in Fig. 5 (b). The error caused by the RF-to-LF unit is in the kHz range, and is directly caused by the inaccuracies of the LO and clock frequencies. In other words, according to the measurements, RF frequency may be accurately measured, after the f
LO
and f clk
are precisely known.
RF-to-LF power measurement results are shown in Fig.
6. The high attenuation evident in these figures can readily be linked to the attenuation caused by the components used in the RF-to-LF signal path.
3.2 VCO measurements
After the initial measurements, which provided a calibration database for real RF-to-LF measurements, some VCO specific measurements were made. In these measurements the Marconi Instruments 2024 was used as
-36.5
-37
-37.5
71.4
71.2
71
70.8
70.6
70.4
70.2
70
0.5
0.7
0.9
1.1
1.3
Vtune/V
1.5
1.7
(a)
Output power (output power variation is -36.12 dBm-(-38.1 dBm)=1.98dB).
-36
1.9
-38
-38.5
0.5
0.7
0.9
1.1
1.3
Vtune/V
1.5
1.7
1.9
(b)
Figure 8 (a) Delta modulation sensitivity and (b) P out
vs. V tune and output power variation

FAN
2500
VCO box: -3 dB attenuator (LAT-3)
CLK box: 74F04 hex inverter
Figure 9 VCO and 26 MHz crystal clock (for the sub-sampler) a VCO by controlling the output frequency with an external analogue voltage.
Fig. 7 (a) shows the behaviour of the VCO output frequency as a function of the VCO control voltage
(V tune
). The subtraction of the measured data from the best-fit line, i.e. tuning linearity, is shown in Fig. 7 (b) along with the measured tuning non-linearity of ± 0.06 kHz. The delta modulation sensitivity (i.e. df out
/dV tune
) is shown in Fig. 8 (a) and the measured behaviour of the
RF output power vs. the tuning voltage in Fig. 8 (b). The measured modulation bandwidth, which was measured via the modulation index method (see http://www.minicircuits.com/appnote/an95004.pdf) was
450 kHz.
After the VCO-type measurements made with the
Marconi Instruments 2024, test circuits shown in Fig. 9 were built around a real life VCO operating around 3.6
GHz and a 26 MHz crystal oscillator to allow more realistic RF device measurements. Due to the higher
3.44
3.4
3.36
3.32
3.28
3.24
y = 0.2378x + 3.1093
measured
LMS
3.2
0.4
0.6
0.8
Vtune/V
(a)
1 1.2
-100
-200
-300
300
200
100
0
-400
0.6
0.7
0.8
0.9
1 1.1
1.2
1.3
Vtune/V
(b)
Figure 10 (a) f
VCO(L)
(f
VCO
measured from the LF output) vs. V tune and (b) tuning linearity
-28
-30
-32
-34
-36
-38
-40
-42
-44
0.6
input frequency, some of the components (e.g. mixer) in the RF-to-LF unit had to be replaced by higher frequency equivalents.
The VCO frequency vs. VCO tuning voltage characteristic, which was measured using the RF-to-LF circuitry, i. e. measurement taken from the LF output, is shown in Fig. 10 (a). The measured tuning sensitivity of
237.8 MHz/V matches the one given in the data sheets of the VCO (250+/-3 MHz/V). The measured tuning linearity is shown in Fig. 10 (b).
Fig. 11 (a) gives the attenuation caused by the RF-to-LF circuitry, i.e. the difference between the RF input power and LF output power of the circuitry, as a function of the
VCO tuning voltage in three measurement runs. As shown, at a given tuning voltage, the error in power measurement is roughly ±1.5 dB. Similarly, in Fig. 11
(b) the frequency measurement error is shown to be around 26 ppm. Note, however, that the measurements of the real-life VCO are hard to make in this kind of a system due to the inherent instability of the VCO output signal. Random changes in frequency and output power during measurement make accurate measurements hard to perform. Therefore, further analysis of the repeatability of measurements was made using a stable reference signal.
0.7
0.8
1.1
1.2
1.3
0.9
Vt/V
1
(a)
26
24
22
20
18
16
0.6
0.7
0.8
0.9
1 1.1
1.2
1.3
Vtune/V
(b)
Figure 11 (a) Attenuation caused by the RF-to-LF circuitry and
(b) frequency error of the measurement

Signal generator f in
≈ 3 GHz…3.1 GHz
Signal generator f mix
≈ 2.9 GHz…3 GHz
T
MBA-35L
-3 dB pulse shaper
φ
S/H
φ
φ
SCANSTA
400
f
CLK
(26.00001) MHz)
Figure 12 Structure of repeatability measurements
3.3 Repeatability of measurements
Since the output signal of a VCO is not very stable, it is hard to estimate the repeatability of measurements when using a VCO as the DUT. Frequency drift and power fluctuation of the VCO itself would mask the frequency and power errors of the RF-to-LF conversion, the repeatability of which is of interest here. Therefore, as shown in Fig. 12, in the following measurements an RF generator was used as the signal source once more. Due to the frequency limitations of the signal generators f in was limited to between 3.0 GHz and 3.1 GHz. A spectrum analyzer was used to measure the LF signals
-24
-25
-26
-27
-28
-29
3 3.05
fin/GHz
(a)
3.1
-56
-56.5
-57
-57.5
-58
-58.5
3 3.05
fin/GHz
3.1
(b)
Figure 13 (a) Attenuation of RF-to-LF circuitry and (b) output power of SCANSTA400
-24
-24.5
-25
-25.5
-26
-26.5
-27
-27.5
-28
-28.5
average maximum minimum average scansta maximum scansta minimum scansta attennuation of box (Pin=0dBm)
-29
3 3.02
3.04
3.06
fin/GHz
3.08
3.1
Figure 14 Attenuation of the RF-to-LF circuitry and output power of SCANSTA400 after the RF-to-LF device and SCANSTA400 [8].
Attenuation produced by the RF-to-LF conversion and
SCANSTA400 and the frequency measurement errors could readily be calculated from the known f out
, f clk
, f mix and f in
and input and output power levels.
A more detailed description of the low-frequency
SCANSTA measurements are discussed in a separate paper presented in ITC 2003 by T. Saikkonen et al.
In repeatability measurements the input frequency was swept ten times from 3 GHz to 3.1 GHz. The attenuation of RF-to-LF conversion and the output power from
SCANSTA400 are shown in Fig.
13.
SCANSTA400 attenuates the incoming LF signal by 30 dB due to the resistivity of the MOS switches used to connect the signal to the internal IEEE 1149.4 analogue busses. This attenuation is removed from the measurements, when necessary, to make the results more comparable.
Fig. 14 shows the maximum, minimum and mean attenuation of RF-to-LF circuitry and the maximum, minimum and mean output power of SCANSTA400 (+30 dB), as calculated from the results of the 10 measurements mentioned above. As shown, the attenuation of the RF-to-LF circuitry and the output power of SCANSTA400 vary about ±0.125 dB and ±0.15 dB, respectively. The apparent difference in the slopes of these curves, which can be calibrated out from the actual measurement results, is caused by the different RF input power used during measurements (i.e. needed to overcome the 30 dB attenuation produced by
SCANSTA400). The slope of the attenuation curves of the RF-to-LF circuitry was found out to be the same as the slope of the output power of SCANSTA400, when 0 dBm input was applied in both measurements (see curve labelled as ‘attenuation of box (Pin=0dBm)’ in Fig. 14).
According to these observations, it seems that some kind

5.1
5
4.9
4.8
4.7
4.6
4.5
4.4
4.3
4.2
2.98
3 3.02
3.04
3.06
3.08
3.1
3.12
f in
/GHz
Figure 15 Frequency measurement error of the RF-to-LF circuitry of clipping occurs inside the RF-to-LF circuitry when the input RF-power approaches 0 dBm.
The frequency measurement error of the RF-to-LF circuitry is shown figure 15. The frequency measurement error was calculated by equation f error
= f in
- f calc
, where f calc
= f out
+ 4f clk
+ f mix
. The almost constant frequency measurement error produced by frequency offsets of the signal generators ranges from 4.3 kHz to 5kHz.
According to the figure, the measurement uncertainty is roughly ±350 Hz.
The use of sub-sampling causes many problems in the
RF-to-LF circuitry. The input RF signal must be mixed to quite a low frequency (100 MHz) to allow the use of a typical not-so-fast switch, which requires a very high LO frequency. With a more advanced high-frequency switch, the mixer could be used for coarse frequency tuning of the input signal, or it would not be needed at all. In any case, sub-sampling has its problems, similar to those that have prevented its wide use in any radio receiver (speed and noise, for example).
Positioning the LF output signal down to the 0-100 kHz band is also problematic. The correct mixing frequency can only be calculated if we knew the frequency of the input signal beforehand. Otherwise the frequency band of interest must be swept by tuning the mixing frequency.
Searching the signal by manual tuning is quite laborious, wherefore an automatic sweep-and-measure system will be needed in practical applications. This may slow down the entire measurement to an unacceptable extent.
Attenuation of the RF-to-LF circuitry (
≈
26 dB) and
SCANSTA400 (
≈
30 dB) cause the signal level to drop very low (i.e. close to the noise level of the measurement instruments). For example, during the VCO measurements the attenuation had to be compensated for by an LF-amplifier, which was connected between the
RF-to-LF circuitry and SCANSTA400. To overcome the attenuation of the RF-to-LF circuitry the sampling circuit should be redesigned for more broadband operation. If the sampler were fast enough and, thus, there were no need for a mixer, the attenuation would certainly drop to a more manageable level.
Accurate VCO measurements are hard to make due to the inherent instability of any unlocked VCO. Several measurements are needed to average results to a meaningful accuracy, which makes measurements slow to perform.
However, due to the many promising results, the next goal of the ongoing project is to design a selection of RFto-LF analogue circuit structures needed to produce IEEE
1149.4 compatible RF Analog Boundary Modules (RF-
ABMs) for basic RF measurements and to realize them as an integrated circuit.
It seems, according to the measurements, that it is possible to make simple RF measurements using a combination of mixing and sub-sampling as the method to down-convert the RF signal into a low frequency suitable for the analogue busses specified in the 1149.4 standard. However, for accurate measurements the RF-to-
LF unit has to be carefully characterized and the gathered information thoroughly used to de-embed the unit from the measurement system. The repeatability of RF power measurement through the whole system, i.e. from the RF input to the output of SCANSTA400, is ±0.15 dB. The measurement uncertainty of RF frequency caused by measurement instruments and signal sources is ±350 Hz.
This work was carried out in the project “Development of the testability of Mixed-Signal circuits” funded by the
Technology Development Center of Finland (TEKES),
Nokia and Elektrobit Group.
[1] IEEE Std 1149.4-1999, “Standard for a Mixed
Signal Test Bus”, IEEE, USA, 2000
[2] H. L. Krauss, C. W. Bostian, F. H. Raab, ”Solid
State Radio Engineering”, John Wiley & Sons, Inc,
USA, 1980
[3] R. Mason, S. Ma, “Analog DFT using an undersampling technique”, IEEE Design & Test of
Computers, Oct-Dec 1999, pp. 84-88.
[4] S. Sunter, K. Filliter, J. Woo, P. McHugh, “A
General Purpose 1149.4 IC with HF Analog Test
Capabilities”, Proceedings IEEE International Test
Conference , 2001, pp. 38-45

[5] “http://www.minicircuits.com/cgi-bin/spec?cat= splitter&model=LRPS-2-25J&pix=qqq569.gif&bv=
4”, Mini-Circuit’s online data sheets, May 2003
[6] http://www.minicircuits.com/cgibin/spec?cat=mixer &model=MBA-
15L&pix=sm2bc.gif&bv=4, Mini-Circuit’s online datasheets, May 2003
[7] “http://www.maxim-ic.com/quick_view2.cfm/ qv_pk/2064”, Maxim’s online data sheets, May
2003
[8] “SCANSTA400 IEEE 1149.4 Analog Test Access
Device”, National Semiconductor Advanced
Information, August 2000