A FREQUENCY MIXING AND SUB-SAMPLING BASED RF-MEASUREMENT
APPARATUS FOR IEEE 1149.4
Juha Hakkinen1, Pekka Syri1, Juha-Veikko Voutilainen2 and Markku Moilanen2
Department of Electrical and Information Engineering, Electronics Laboratory1, Optoelectronics and
Measurement Techniques Laboratory2, University of Oulu, Finland
Abstract
This paper demonstrates the possibility of performing
radio frequency (RF) measurements in the IEEE 1149.4
environment using a selected combination of mixing and
sub-sampling as the method to down-convert the RF
signal into a low frequency suitable for the analogue
busses specified by the 1149.4 standard. Signals
generated by RF signal generators (2 GHz) and a
commercial VCO (3 GHz) were used to demonstrate the
operation of the developed RF-to-LF circuitry in
cooperation with National’s SCANSTA400 IEEE 1149.4
Analog Test Access Device. The measured repeatability
of RF power measurement through the whole system, i.e.
from the RF input to the output of SCANSTA400, is ±0.15
dB and the measurement uncertainty of the RF frequency
due to measurement instruments and signal sources is
±350 Hz. Interconnect measurements at 2.1 Ghz were
also demonstrated. Simple faults such as shorts, opens
and missing components may readily be detected at the
low-frequency output of the RF-to-LF circuitry.
1.
Introduction
Reliable, accurate, fast and low-cost testing of products is
one of the cornerstones of modern electronics industry.
Here the term testing covers production testing,
calibration, tuning and maintenance testing. Especially
the radio frequency (RF) circuitry presents a tough
challenge in the testing of electrical products (e.g.
cellular phones, base stations). Testing and tuning of
high-frequency parts of electrical systems is extremely
time-consuming and requires special, often expensive,
test equipment. Furthermore, RF testing will continue to
be a major challenge in the future because of new
standards (e.g. Bluetooth and IEEE 802.11) and multiband/multi-function products.
The emerging testing standard IEEE 1149.4 [1], which
extends the well-known digital boundary scan
architecture (i.e. JTAG) by adding analogue
measurement access to the boundary scan cells, is mainly
targeted for low-frequency testing of PCB interconnects
and measuring discrete component values, which has
previously been demonstrated, for example, in [2] where
methods for the calculation of passive component values
in some simple situations has been established. However,
due to the limited frequency range accommodated by the
standard (several decades below RF signals), the standard
is not directly applicable to the testing of high-frequency
circuitry. However, if the standard could be used for RF
testing, the speed (e.g. BIST) and cost (low-frequency
low-cost equipment may be used) improvements could be
considerable. In addition, this would allow the use of
standard (1149.4) test equipment available in the future.
In order to allow the measurement of RF signals, some
kind of signal processing which extracts the information
one is measuring from the RF signal and to present the
result as a low-frequency signal (i.e. below 100 kHz)
further processed by structures specified in the standard
(1149.4) must be used.
As the first step towards the application of the IEEE
1149.4 architecture for RF testing, embedded RF test
circuitry inside RF ASICs and low-cost measurement
instrument are used to replace the conventional and
expensive RF testing equipment. Whether this is cost
effective when the whole test system is considered
depends on integration costs. However, at least RF
interfacing and probing problems can be avoided during
testing.
In the future cost-effective Life-Cycle-Testing (LCT)
with high fault coverage and advanced diagnostics may
be achieved with totally embedded testing, which
provides test reusability during the entire product life
cycle. The embedded test architecture may be based on
the IEEE 1149.4 architecture with added RF capability,
which provides integrated access, stimulus generation
and measurement instruments (BIST, Build-In SelfTest).
ITC INTERNATIONAL TEST CONFERENCE
0-7803-8580-2/04 $20.00 Copyright 2004 IEEE
Paper 19.2
551
fi (t)
fc
τ
t
T
fo(t)
fi(t)
fo(t)
τ
t
T
Figure 1 The sub-sampling process
In developing the embedded RF testing the first step is to
investigate the possible frequency down-conversion
techniques in cooperation with the 1149.4 low-frequency
architecture. The accuracy of the measurement of the
translated RF signal property and complexity of the
required circuitry are the principal issues when
evaluating the usability of different RF test structures for
further use, e.g. for ASIC realizations. Therefore, the
specific aim of this project was to demonstrate that it is
possible to perform simple high-frequency measurements
by suitable RF analogue circuit structures (i.e. RF-to-LF
signal processing) in cooperation with the IEEE 1149.4
specific structures from target RF frequencies from 2 to 3
GHz.
The Apparatus
There are several ways to translate a high-frequency
signal down to low frequencies, out of which frequency
mixing [3] and sub-sampling [4] are the ones most often
employed. In the design reported here a combination of
these methods was used. Mixing alone would certainly
|F(ω)|
f
∆f
fc
2fc
3fc
∆f ∆f
∆f ∆f
∆f ∆f
work for us, but the combination of the two adds freedom
in system design. Additionally, sub-sampling from
intermediate frequencies may even be directly performed
by utilizing the circuit structures specified in the standard
as demonstrated in the literature [5]. However, sub-
The sub-sampling process can now be straightforwardly
presented in the frequency domain using equation (1). As
shown in Fig. 2, a copy of the narrowband signal at fRF is
produced around multiples of the sampling frequency.
Now, a low-pass filter, for example, may be used to
extract only the desired harmonic for further processing.
110
1
100
fLO
(≈ 1.9 GHz )
φ
-3 dB
T
φ
S/H
φ
fout
(to SCANSTA 400)
( ≈ 50 kHz )
80
70
60
50
fclk=9.985050 MHz
fclk=9.988390 MHz
fclk=9.991011 MHz
40
2.01 2.02 2.03 2.04 2.05 2.06 2.07 2.08 2.09
f vco/GHz
f clk (≈ 9.995 MHz)
552
-2
-3
-5
2
Figure 3 Structure of the RF-to-LF processing unit
-1
-4
30
Paper 19.2
fclk=9.985050 MHz
fclk=9.988390 MHz
fclk=9.991011 MHz
0
90
fout /kHz
≈ 4 GHz
band -stop
fIF
( ≈ 100 MHz )
(1)
n = −∞
n ≠0
≈ 2 GHz
band -pass
pulse shaper
∞
∑ (sin(nπd )) /(nπd )F (ω − nωc ) ,
2.2
Realization
The structure of the RF-to-LF processing unit is shown in
VCOOUT
fvco
( ≈ 2 GHz )
G (ω ) = dF (ω ) + d
f
Figure 2 Sub-sampling in the frequency domain
LRPS-2-25J
2.1
Sub-sampling
If a signal is sampled by a rate, which is less than the
frequency of the signal, the signal is said to be subsampled [4]. In this case, if the signal bandwidth is small,
a copy of the original signal is produced at low
frequencies by the well-known process of aliasing. The
sub-sampling process is shown in Fig. 1 and can be
presented mathematically as [4]
where G(ω) = Fourier transform of fo(t), d = τ/T, F(ω) =
Fourier transform of fi (t), τ = width of the sampling
pulse, T = 1/fc = time between samples and ωc = 2π/T.
{
{
fRF
=nfc + ∆f
{
{
fc
{
{
|G(ω)|
sampling alone is not conveniently done from frequencies
above 1 GHz. Therefore, linear mixing was used to shift
the input signal to the 100 MHz range, where sampling is
more convenient as far as noise aliasing and the speed of
the sampler are considered. Due to its familiarity,
frequency mixing is not treated theoretically here (e.g.
[3]).
∆fvco/kHz
2.
Figure 4 The RF-to-LF measurement unit
(a)
2.1
2
2.02
2.04
2.06
2.08
2.1
fvco/GHz
(b)
Figure 5 (a) Measured fout vs. fvco and (b) measurement error
3.
Experimental Results
3.1
Characterization
The RF-to-LF unit was characterized by measuring the
output signal of a laboratory RF signal generator
(Marconi Instruments 2024). Agilent 331220A was used
as the LO generator and HP 4396A Spectrum analyzer to
measure the reference and the LF output.
The measured final LF output frequency as the function
of the input frequency (fvco) is shown in Fig. 5 (a). Given
-31
-32
-33
-34
-35
-36
-37
-38
-39
-40
-41
2.0GHz
2.02GHz
2.04GHz
2.06GHz
2.08GHz
2.1GHz
0
2.01GHz
2.03GHz
2.05GHz
2.07GHz
2.09GHz
-10
-20
Pout/dBm
Pout /dBm
Fig. 3 and the actual in-a-box realization in Fig. 4. To
allow reference measurements by traditional RF
measuring equipment, the input RF signal (fvco) is first
split into two by a power splitter (LRPS-2-25J [6]). The
input signal is then converted to an intermediate
frequency by a mixer (MBA-15L [7]) and finally subsampled to the proper output frequency by a S/H circuit
(constructed around MAX4619 [8]). The recommended
frequencies for the different signals in Fig. 3 when fvco ≈
2.0 GHz are: fLO ≈ 1.9 GHz ⇒ fIF ≈ 100 MHz, fclk
≈ 9.995 MHz ⇒ fout ≈ 50 kHz (i.e. suitable for structures
found in National’s SCANSTA400 IEEE 1149.4 Analog
Test Access Device [9]).
-30
-40
fmix=1.9GHz
-50
fmix=1.95GHz
-60
fmix=1.99GHz
-70
-2
-1
0
1
2
3
-80
0
10000
20000
30000
Pin/dBm
40000
50000
60000
fout/Hz
(a)
Figure 6 (a) Pout vs. Pin and (b) Pout vs. fout
(b)
the signals shown in Fig. 3, the output frequency can be
calculated using the following equation:
fout = (fvco – fLO) – nfclk, where n = 10,11,…20
(2)
Equation (2) can also be used to calculate the original RF
input frequency from the measured LF output frequency.
Thus, the calculated error between the reference
measurement and the calculated output frequency is
shown in Fig. 5 (b). The error caused by the RF-to-LF
unit is in the kHz range, and is directly caused by the
inaccuracies of the LO and clock frequencies. In other
words, according to the measurements, RF frequency
may be accurately measured, after the fLO and fclk are
precisely known.
RF-to-LF power measurement results are shown in Fig.
6. The high attenuation evident in these figures can
y = 70.895x - 25.471, Tuning sensitivity is 70.895 kHz/V
71.4
110
100
71.2
(dfout/dVtune)/kHz/V
90
fout/kHz
80
70
60
50
measured
40
LMS
30
20
71
70.8
70.6
70.4
70.2
10
0.5
0.7
0.9
1.1
1.3
1.5
1.7
1.9
70
0.5
Vtune/V
0.7
0.9
(a)
1.1
1.3
Vtune/V
1.5
1.7
1.9
(a)
Output power (output power variation is -36.12 dBm-(-38.1
dBm)=1.98dB).
Tuning linearity (non-linearity is +/-0.05543kHz).
0.06
-36
0.05
-36.5
0.03
Pout/dBm
fout-ftrend/kHz
0.04
0.02
0.01
0
-37
-37.5
-0.01
-0.02
-38
-0.03
-0.04
0.5
0.7
0.9
1.1
1.3
1.5
1.7
1.9
Vtune/V
Figure 7
(b)
(a) VCO frequency tuning characteristic and (b)
tuning linearity
-38.5
0.5
0.7
0.9
1.1
1.3
Vtune/V
1.5
1.7
1.9
(b)
Figure 8 (a) Delta modulation sensitivity and (b) Pout vs. Vtune
and output power variation
Paper 19.2
553
3.2V
2.7V
FAN
2500
Vtune
fvco
VCO
0.7…1.2V
via
the
modulation
index
method
(see
http://www.minicircuits.com/appnote/an95004.pdf) was
450 kHz.
3.23…3.98 GHz
VCO box: -3 dB attenuator (LAT-3)
CLK box: 74F04 hex inverter
Figure 9 VCO and 26 MHz crystal clock (for the sub-sampler)
readily be linked to the attenuation caused by the
components used in the RF-to-LF signal path.
3.2
VCO measurements
After the initial measurements, which provided a
calibration database for real RF-to-LF measurements,
some VCO specific measurements were made. In these
measurements the Marconi Instruments 2024 was used as
a VCO by controlling the output frequency with an
external analogue voltage.
Fig. 7 (a) shows the behaviour of the VCO output
frequency as a function of the VCO control voltage
(Vtune). The subtraction of the measured data from the
best-fit line, i.e. tuning linearity, is shown in Fig. 7 (b)
along with the measured tuning non-linearity of ± 0.06
kHz. The delta modulation sensitivity (i.e. dfout/dVtune) is
shown in Fig. 8 (a) and the measured behaviour of the
RF output power vs. the tuning voltage in Fig. 8 (b). The
measured modulation bandwidth, which was measured
After the VCO-type measurements made with the
Marconi Instruments 2024, test circuits shown in Fig. 9
were built around a real life VCO operating around 3.6
GHz and a 26 MHz crystal oscillator to allow more
realistic RF device measurements. Due to the higher
input frequency, some of the components (e.g. mixer) in
the RF-to-LF unit had to be replaced by higher frequency
equivalents.
The VCO frequency vs. VCO tuning voltage
characteristic, which was measured using the RF-to-LF
circuitry, i. e. measurement taken from the LF output, is
shown in Fig. 10 (a). The measured tuning sensitivity of
237.8 MHz/V matches the one given in the data sheets of
the VCO (250+/-3 MHz/V). The measured tuning
linearity is shown in Fig. 10 (b).
Fig. 11 (a) gives the attenuation caused by the RF-to-LF
circuitry, i.e. the difference between the RF input power
and LF output power of the circuitry, as a function of the
VCO tuning voltage in three measurement runs. As
shown, at a given tuning voltage, the error in power
measurement is roughly ±1.5 dB. Similarly, in Fig. 11
(b) the frequency measurement error is shown to be
3.44
-28
-30
3.4
Attenuation/dB
fVCO(L)/GHz
-32
3.36
y = 0.2378x + 3.1093
3.32
measured
3.28
LMS
-34
-36
-38
-40
3.24
-42
3.2
0.4
0.6
0.8
1
-44
1.2
0.6
0.7
0.8
0.9
1
Vtune/V
Vt/V
(a)
(a)
1.1
1.2
1.3
300
26
100
24
ferror/ppm
fVCO(L)-fLMS/kHz
200
0
-100
22
20
-200
-300
18
-400
0.6
0.7
0.8
0.9
1
1.1
1.2
1.3
Vtune/V
16
0.6
0.7
0.8
0.9
1
1.1
1.2
1.3
Vtune/V
(b)
Figure 10 (a) fVCO(L) (fVCO measured from the LF output) vs. Vtune
and (b) tuning linearity
Paper 19.2
554
(b)
Figure 11 (a) Attenuation caused by the RF-to-LF circuitry and
(b) frequency error of the measurement
-24
-24.5
Signal generator
fin ≈ 3 GHz…3.1 GHz
-25
MBA-35L
Signal generator
-3 dB
φ
fmix ≈ 2.9 GHz…3 GHz
S/H
T
SCANSTA
400
φ
φ
Attennuation/dB
-25.5
-26
-26.5
-27
average
maximum
minimum
average scansta
maximum scansta
minimum scansta
attennuation of box (Pin=0dBm)
-27.5
pulse shaper
-28
-28.5
fCLK (26.00001) MHz)
-29
Figure 12 Structure of repeatability measurements
3
3.02
3.04
3.06
3.08
3.1
fin/GHz
around 26 ppm. Note, however, that the measurements of
the real-life VCO are hard to make in this kind of a
system due to the inherent instability of the VCO output
signal. Random changes in frequency and output power
during measurement make accurate measurements hard
to perform. Therefore, further analysis of the
repeatability of measurements was made using a stable
reference signal.
3.3 Repeatability of measurements
Since the output signal of a VCO is not very stable, it is
hard to estimate the repeatability of measurements when
using a VCO as the DUT. Frequency drift and power
Attennuation/dB
-24
Attenuation of the RF-to-LF circuitry and output
power of SCANSTA400
fluctuation of the VCO itself would mask the frequency
and power errors of the RF-to-LF conversion, the
repeatability of which is of interest here. Therefore, as
shown in Fig. 12, in the following measurements an RF
generator was used as the signal source once more. Due
to the frequency limitations of the signal generators fin
was limited to between 3.0 GHz and 3.1 GHz. A
spectrum analyzer was used to measure the LF signals
after the RF-to-LF device and SCANSTA400 [9].
Attenuation produced by the RF-to-LF conversion and
SCANSTA400 and the frequency measurement errors
could readily be calculated from the known fout, fclk, fmix
and fin and input and output power levels.
In repeatability measurements the input frequency was
swept ten times from 3 GHz to 3.1 GHz. The attenuation
of RF-to-LF conversion and the output power from
SCANSTA400 are shown in Fig. 13.
-25
-26
-27
-28
-29
3
3.05
fin/GHz
3.1
(a)
-56
-56.5
P/dBm
Figure 14
-57
-57.5
-58
-58.5
3
3.05
fin/GHz
3.1
(b)
Figure 13 (a) Attenuation of RF-to-LF circuitry and (b) output
power of SCANSTA400
SCANSTA400 attenuates the incoming LF signal by 30
dB due to the resistivity of the MOS switches used to
connect the signal to the internal IEEE 1149.4 analogue
busses. This attenuation is removed from the
measurements, when necessary, to make the results more
comparable.
Fig. 14 shows the maximum, minimum and mean
attenuation of RF-to-LF circuitry and the maximum,
minimum and mean output power of SCANSTA400 (+30
dB), as calculated from the results of the 10
measurements mentioned above. As shown, the
attenuation of the RF-to-LF circuitry and the output
power of SCANSTA400 vary about ±0.125 dB and ±0.15
dB, respectively. The apparent difference in the slopes of
these curves, which can be calibrated out from the actual
measurement results, is caused by the different RF input
power used during measurements (i.e. needed to
overcome the 30 dB attenuation produced by
SCANSTA400). The slope of the attenuation curves of
the RF-to-LF circuitry was found out to be the same as
Paper 19.2
555
4.
5.1
5
4.9
f error/kHz
4.8
4.7
4.6
4.5
4.4
4.3
4.2
2.98
3
3.02
3.04
3.06
3.08
3.1
3.12
fin/GHz
Figure 15 Frequency measurement error of the RF-to-LF
circuitry
the slope of the output power of SCANSTA400, when 0
dBm input was applied in both measurements (see curve
labelled as ‘attenuation of box (Pin=0dBm)’ in Fig. 14).
According to these observations, it seems that some kind
of clipping occurs inside the RF-to-LF circuitry when the
input RF-power approaches 0 dBm.
The frequency measurement error of the RF-to-LF
circuitry is shown figure 15. The frequency measurement
error was calculated by equation ferror = fin - fcalc, where
fcalc = fout + 4fclk + fmix. The almost constant frequency
measurement error produced by frequency offsets of the
signal generators ranges from 4.3 kHz to 5kHz.
According to the figure, the measurement uncertainty is
roughly ±350 Hz.
Interconnect measurements
At this point it was recognized, that the measurements so
far did not quite fulfill the basic premises of the IEEE
1149.4 standard – namely interconnect measurements.
The low-frequency interconnect measurements proposed
by the standard are for the most part meant for detecting
faulty opens and shorts on the printed circuit assembly
(PCA), which arise from faulty soldering and problems in
the manufacturing process, for example. However, these
low-frequency measurements cannot be used for detecting
faulty short and opens in RF interconnects, where DC
blocking (i.e. series capacitors) is often used.
Additionally, the standard does not lend itself at all to the
measurement of the actual RF properties of the
interconnect transmission lines and RF components, i.e.
characteristic impedances and insertion loss, for example.
In the following paragraphs, the RF-to-LF circuitry will
be used to demonstrate the detection of faulty shorts and
opens in an RF transmission line operating at 2.1 GHz.
There are several ways to detect a faulty open or short at
the end of an RF transmission line. All the basic methods
investigated here are based on the detection amplitude
and/or phase of the signal reflected back from the load
(i.e. open or short) as shown in Fig. 16. Simple amplitude
detection is especially useful, when the intended, nonfaulty, load provides a good impedance match, i.e. the
difference between the magnitude of the reflected wave
between the non-faulty (i.e. ideally a zero reflected wave
Z0
Z0
Z0
Z0
(a)
(b)
∆φ
Z0
Z0
Z0
Z0
(c)
Figure 16
Paper 19.2
556
(d)
Different reflection cases. (a) Perfectly matched load, (b) faulty open, (c) faulty short and (d) faulty open, with nonmatched load.
RF source
50 Ω TL
Coupler
0.05
0.04
LO source
RF
LF
G + LP
STA400
LF-out/V
lock
open
short
0.03
Reflected
signal
CLK source
0.02
0.01
50Ω
0
-0.01
-0.02
-0.03
-0.04
REF source
-0.02
(a)
-0.01
0
0.01
0.02
0.03
0.02
0.03
time/s
(a)
0.05
0.04
short
long
LF-out/V
0.03
0.02
0.01
0
-0.01
-0.02
-0.03
-0.04
-0.02
-0.01
0
(b)
Figure 17
Measurement
setup
for
the
reflection
measurements. (a) Schematic and (b) laboratory
setup.
as in Fig. 16. (a)) and faulty (ideally the reflected wave
has the same magnitude as the input wave Fig. 16. (b)
and (c)) cases is large. However, in situations where the
original load is poorly matched to the transmission line
impedance, i.e. the reflected wave does not change its
magnitude whether there is a fault or not, the only way to
detect the fault is through measurements of changes in
the electrical length of the transmission line, which may
be accomplished by measuring the phase difference ∆φ of
the reflected waves as in Fig. (d), for example. This
simple measurement may, however, miss the fault if ∆φ
happens to equal 360° due to the different complex
reflection coefficients of the load and the fault or the
exact position of the fault. On the other hand, the
possibility of a miss is relatively small.
All reflection measurements were made using the
measurement setup of Fig. 17. The 2.1 GHz test signal is
connected to the test transmission line through a
directional coupler (Coupler), which connects only the
reflected wave to the RF-to-LF apparatus. Since phase
measurements can only be made relative to some phase
0.01
time/s
(b)
Figure 18
Signals at the output of the LF amplifier/low-pass
filter, when (a) the termination of a 50 Ω
transmission line is 50 Ω, ∞ Ω (open) and 0 Ω
(short) and (b) when the length of an open-ended
transmission line is changed by approximately ¼
of the wavelength.
reference, all signal sources (RF, LO, CLK and REF) are
frequency locked, out of which REF source provides the
reference signal against which the 100 Hz output of the
RF-to-LF circuitry is compared using an oscilloscope.
Due to the high level of attenuation caused by the RF-toLF circuitry and the SCANSTA 400 PCB (STA400), an
LF amplifier low-pass-filter combination (G + LP) was
used to boost the LF signal of interest. Furthermore, the
low-pass filter reduces the high-frequency components of
the RF-to-LF circuitry and thus makes the output more
readable from the screen of the oscilloscope.
As shown in Fig. 18. (a) the low-frequency output may be
used to detect differences in the magnitude of the
reflected wave in case of a good and poor matches.
Additionally, the phase of the output waveform clearly
indicates the type of load, i.e. open or short. Since it is
easy make out the difference between the amplitudes of
the well matched 50 Ω load case and the two cases
Paper 19.2
557
mimicking most common failure mechanisms (i.e.
electrical short and missing or poorly soldered component
at the end of the transmission line), this test seems quite
practical and useful.
In Fig. 18. (b) an open-ended transmission line was first
measured and the output signal then compared to that of
a shorter transmission line. The different electrical length
clearly manifests itself in the LF output of the RF-to-LF
circuitry. In this case, the difference between the lengths
of the transmission lines was roughly ¼ of the
wavelength, thus producing approximately 90° phase
difference. The signals in Fig. 18. (b) are quite noisy due
to the harmonics of the sampling frequency, which would
suggest that the detection of small differences in
transmission line lengths would be hard to detect.
However, the signals could very easily be filtered to those
superimposed on the measurement results with a higher
order low-pass filter than used during measurements.
Even so, the detection of phase differences is by far more
difficult than the detection of amplitude differences.
However, unmatched terminations with reflection
coefficient magnitudes of one are quite rare in practice,
thus, the amplitude detection may often be used in
conjunction with the phase measurement for increased
reliability of fault detection.
5.
Discussion
The use of sub-sampling causes many problems in the
RF-to-LF circuitry. The input RF signal must be mixed to
quite a low frequency (100 MHz) to allow the use of a
typical not-so-fast switch, which requires a very high LO
frequency. With a more advanced high-frequency switch,
the mixer could be used for coarse frequency tuning of
the input signal, or it would not be needed at all. In any
case, sub-sampling has its problems, similar to those that
have prevented its wide use in any radio receiver (speed
and noise, for example).
Positioning the LF output signal down to the 0-100 kHz
band is also problematic. The correct mixing frequency
can only be calculated if we knew the frequency of the
input signal beforehand. Otherwise the frequency band of
interest must be swept by tuning the mixing frequency.
Searching the signal by manual tuning is quite laborious,
wherefore an automatic sweep-and-measure system will
be needed in practical applications. This may slow down
the entire measurement to an unacceptable extent.
Overall, the testing time using the described circuitry is
at the moment comparable to that of using conventional
dedicated RF test instruments. In other words, attaching
measurement cables (or probes) and producing/reading
the measurement result is roughly the same. The savings
in testing time come into play only after a measurement
Paper 19.2
558
circuit similar to the one proposed here is embedded into
the DUT itself. In this case, the test access is always
available without any special test arrangement or ATE,
for example. Additional savings in time can be achieved
through optimized test software, which may be partially
implemented in hardware for further operational speed.
Therefore, the integration of the RF-to-LF circuitry as
part of the equipment under test is the road to test cost
savings during production testing and even more so in
field tests, where the production ATE and test equipment
may not be readily available. However, even if the
presented circuit in its current form does not provide any
savings in testing time, the cost savings compared to
conventional pieces of equipment is considerable. The
total cost of the described circuit is in the $100-or-less
range, which is from one upto two decades below that of
the combined cost of a commercial RF power and
frequency meter. Naturally, savings due to the simplicity
of the circuit come at the expense of reduced accuracy.
Attenuation of the RF-to-LF circuitry (≈ 26 dB) and
SCANSTA400 (≈ 30 dB) cause the signal level to drop
very low (i.e. close to the noise level of the measurement
instruments). For example, during the VCO
measurements the attenuation had to be compensated for
by an LF-amplifier, which was connected between the
RF-to-LF circuitry and SCANSTA400. To overcome the
attenuation of the RF-to-LF circuitry the sampling circuit
should be redesigned for more broadband operation. If
the sampler were fast enough and, thus, there were no
need for a mixer, the attenuation would certainly drop to
a more manageable level.
Accurate VCO measurements are hard to make due to the
inherent instability of any unlocked VCO. Several
measurements are needed to average results to a
meaningful accuracy, which makes measurements slow
to perform.
Interconnect measurements at RF may be performed
using the proposed apparatus. Simple faults, such as
shorts, opens and missing components may readily be
detected at the low-frequency output of the RF-to-LF
circuitry. Here, only simple reflection measurements at a
constant frequency were briefly investigated. Interconnect
faults, which change the magnitude of the reflected wave
are easily detected, since they don’t require a phase
reference. However, if some kind of low-frequency
reference signal is available, faults creating phase
changes in the reflected wave, e.g. shorts, opens or wrong
termination impedances, may be detected through phase
measurements. In our case, the phase reference was
generated by an auxiliary low-frequency generator and by
carefully frequency locking all signal generators, which
may not be possible in a real-life implementation.
Due to the many promising results, the next goal of the
ongoing project is to design a selection of RF-to-LF
analogue circuit structures needed to produce IEEE
1149.4 compatible RF Analog Boundary Modules (RFABMs) for basic RF measurements and to realize them as
an integrated circuit.
6.
Conclusions
It seems, according to the measurements, that it is
possible to make simple RF measurements using a
combination of mixing and sub-sampling as the method
to down-convert the RF signal into a low frequency
suitable for the analogue busses specified in the 1149.4
standard. However, for accurate measurements the RF-toLF unit has to be carefully characterized and the gathered
information thoroughly used to de-embed the unit from
the measurement system. The repeatability of RF power
measurement through the whole system, i.e. from the RF
input to the output of SCANSTA400, is ±0.15 dB. The
measurement uncertainty of RF frequency caused by
measurement instruments and signal sources is ±350 Hz.
Interconnect measurements at RF may also be performed
using the proposed apparatus. Simple faults, such as
shorts, opens and missing components may readily be
detected at the low-frequency output of the RF-to-LF
circuitry.
7.
[1]
[3]
[4]
[5]
[6]
[7]
[8]
Acknowledgements
This work was carried out in the project “Development of
the testability of Mixed-Signal circuits” funded by the
Technology Development Center of Finland (TEKES),
Nokia and Elektrobit Group.
8.
[2]
[9]
T. Saikkonen, J-V. Voutilainen, M. Moilanen,
“Some Methods to Calculate the Values of Passive
Components from the Measurements Made with an
1149.4 Compliant Device”, 2nd IEEE International
Board Test Workshop, October 2-3, 2003, Charlotte
USA, 10p
H. L. Krauss, C. W. Bostian, F. H. Raab, ”Solid
State Radio Engineering”, John Wiley & Sons, Inc,
USA, 1980
R. Mason, S. Ma, “Analog DFT using an
undersampling technique”, IEEE Design & Test of
Computers, Oct-Dec 1999, pp. 84-88.
S. Sunter, K. Filliter, J. Woo, P. McHugh, “A
General Purpose 1149.4 IC with HF Analog Test
Capabilities”, Proceedings IEEE International Test
Conference, 2001, pp. 38-45
Mini-Circuits, “LRPS-2-25J Power Splitter”, http://
www.minicircuits.com/cgi-bin/spec?cat=splitter&
model=LRPS-2-25J&pix=qqq569.gif&bv=4, MiniCircuit’s online data sheets, May 2003
Mini-Circuits, “MBA-15L Frequency Mixer”,
http://www.minicircuits.com/cgibin/spec?cat=mixer
&model=MBA15L&pix=sm2bc.gif&bv=4, Mini-Circuit’s online
datasheets, May 2003
Maxim, “MAX4617, MAX4618, MAX4619 HighSpeed, Low-Voltage, CMOS Analog Multiplexers/
Switches“,http://www.maxim-ic.com/quick_view2.
cfm/qv_pk/2064, Maxim’s online data sheets, May
2003
“SCANSTA400 IEEE 1149.4 Analog Test Access
Device”, National Semiconductor Advanced
Information, August 2000
References
IEEE Std 1149.4-1999, “Standard for a Mixed
Signal Test Bus”, IEEE, USA, 2000
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