Low-Power Low-Voltage Chopped
Amplier with a New Class AB Output
Stage for Mixed Level Applications
M.A.T. Sanduleanu1 , A.J.M. van Tuijl1;2, R.F. Wassenaar1 , M.C. Lammers1 and H. Wallinga1
1
University of Twente,
Department of Electrical Engineering
P.O.Box 217, 7500 AE Enschede, The Netherlands
tel. +31 (0)53 489 4007
fax. +31 (0)53 4892 799
2
Philips Semiconductors, Nijmegen, The Netherlands
m.a.t.sanduleanu@el.utwente.nl
Abstract | This paper describes the principle and namic range [1]. The solution would be to keep the
the design of a 0.5m CMOS, low-power, low- largest possible swing but things get worse because
voltage, chopped amplier for noise and oset re- of the conict between large swings and cascoding
duction in mixed analogue digital applications. The needed to get high gains. In conventional chopper
operation is based on chopping and dynamic eleopamps [2], [3], [4] for 1/f noise and oset
ment matching to reduce noise and oset, without stabilized
reduction,
dierential ampliers are being used and
excessive increase of the charge injection residual
oset. It consists of a chopped transconductance bandwidth is limited to few tens of kHz. Switching
stage and a new class AB stage capable of working at the dierential output will introduce most of the
at 1.3V supply voltage. The main goal is to achieve switching noise and residual oset [1]. Other solutions
low residual osets by chopping at high frequencies for oset reduction like ping-pong techniques have the
reducing at the same time the 1/f noise of the am- disadvantage of high power consumption and linearplier. Loaded with a heavy load 32
jj 300pF it ity problems [5]. Rail to rail input stages have ohas a 91dB open loop gain and a GBW of 1.8MHz. set and linearity problems [6]. This paper presents a
Simulations show a THD of -90dB for a 1k
load
and -83dB for a 32
load and 1KHz input signal. chopped amplier which circumvents the above menThe simulated static oset is 1.67mV. The simu- tioned drawbacks. The chopping frequency can reach
lated residual oset is 450V at 10MHz chopping. MHz range without excessive increase in residual oAt 1KHz chopping, the oset becomes lower than set. The new class AB output stage allows large volt1V.
age swings, low osets and high linearity. The opamp
can be used wherever oset and noise specications
are important. However, the chopped amplier is priI. Introduction
marily meant for use as an amplier capable of driving
In a CMOS technology for mixed signal applications, headphones in portable digital audio. The 1 bit outlowering the supply voltage is a must, being dictated put of the D/A converter has to be ltered in order
by digital signal processing functions. The process, to get less out of band noise and to eliminate the reptuned towards digital needs becomes less and less com- etitions of the spectra due to the sampling process.
patible with analog requirements. As a consequence, The analog lter has to do the job without increasing
for analog functions built in digital CMOS, the sig- the in-band noise and with high linearity. Besides,
nal swing reduces with power supply and so does the it has to be integrated on the same chip with the
dynamic range. 1/f noise properties are worsened for D/A converter and the sampled-data FIR lter, in
surface PMOST's in comparison to buried transistors a pure digital CMOS technology. The input signal for
and this, under the presence of substrate bounce due the chopped opamp, consists of a baseband spectrum
to low ohmic substrates which reduces further the dy- and the attenuated aliases centered around multiples
451
Proceedings of the ProRISC Workshop on Circuits, Systems and Signal Processing 1997
of the sampling frequency. In this particular design,
64 times oversampling is used, such that the eective
sample frequency is 2.8MHz. Chopping at multiples
of sampling frequency is an advantage, because, the
spectrum is quiet, due to the sin(x)=x type of dips
from sampling process. Chopping at high frequency
will produce residual oset which has to be withstand
by the headphone causing dc currents. The paper
focuses on the design and the realization of an amplier with a class AB output stage capable of chopping
up to 10MHZ, with low noise, high linearity and low
residual oset.
Noise at output (dBuV/root(Hz))
452
Chopamp
40
30
20
10
0
-10
-20
-30
-40
-50
10
Vout(f)
m(t)
M(f)
1/n
1
2fchop
fchop
-1
6fchop
Fig. 2. Chopped amplier spectrum
5fchop
3fchop
m(t)
Vin(t)
4fchop
99.43 988.6 9829 97724 1E+0 9.66E 9.61E
6
+06 +07
Frequency (Hz)
m(t)
A
Vout(t)
Snoise(f)
Vnoise+offset
Sthermal
fchop
3fchop
5fchop
Fig. 1. Chopping principle
II. Chopping principle
The chopping principle is depicted in g.1. Here, the
input signal is multiplied with a rectangular signal
m(t) with unity amplitude and 50% duty-cycle. As a
result, the signal is modulated at odd harmonics of the
chopper frequency because, the chopping signal has
spectral contributions only at odd harmonics. The
signal will be amplied and modulated back, leaving
spectral contributions at even harmonics of the chopper frequency. The strength of the modulation signal
decreases with 1=n where n is the harmonic number.
Therefore, chopping is not producing important
foldover eects as in the sampling case. Oset and
1/f noise are modulated at odd harmonics leaving the
baseband free of 1/f noise. In the ideal chopping case,
the bandwidth of the amplier should be innity. As
long as this is true, multiplying the signal twice with
m(t) will reconstruct the input signal ideally. If the
bandwidth of the amplier is limited, the result is a
high frequency residue centered around the even harmonics and the signal in the baseband is attenuated.
The eect of chopping on 1/f noise can be seen from
g.2 where the measured low frequency spectrum of
a chopped amplier is shown. Here, the chopping frequency is 1MHz. Two situations are considered: an
unchopped version and a chopped one. The residual
noise, left at low frequencies, is just the thermal noise.
A necessary condition to cancel out the 1/f noise can
be derived from g.3. This condition would be:
fchop BWsignal + fcorner
(1)
Snoise(f)
Sthermal
fchop
fchop-fcorner
BW signal
f
Fig. 3. Baseband spectrum
The eect of chopping on white and 1/f noise can be
quantied by considering the power spectral density
of the noise at the output as shown in g.4. A nite
amplier bandwidth has been assumed. The normalized power spectral density of the white noise remains
unchanged, as long as, the bandwidth of the amplier
is about ten times larger than the chopping frequency.
Under this condition, the normalized power spectral
density of the 1/f noise remains nite at low frequencies.
In g.4 the constant kF is the 1/f noise constant and
depends on the process. As a conclusion, if we want
to chop at high frequencies, we need a large amplier
bandwidth compared to chopping frequency in order
to reduce 1/f noise and to let the thermal noise unchanged.
Low-Power Low-Voltage Chopped Amplier with a New Class AB Output Stage
453
III. Basic Circuit
VDD
In conventional choppers, the signal is being transposed at the input of the dierential pair, amplied
and demodulated back at the output nodes as shown
in g.5.
Switching at high impedance nodes would be disadvantageous due to limited bandwidth of the amplier.
In this approach, high frequency chopping is not possible and therefore, this method is limited to few tens
of KHz. Besides, the switching noise is directly coupled to the output. Because we have switches in the
middle of the supply voltage, for low voltage applications charge pumps are needed. This is to ensure that
all switches are rmly open and/or closed. The basic
circuit, shown in g.6, comprises an input modulator, a PMOS dierential pair, current sources and a
low voltage, high bandwidth cascoded mirror, to perform a dierential to single ended conversion. The second chopper would transpose again the signal at low
impedance nodes and demodulates back the signal,
canceling out the oset of the bottom transistors. The
oset and noise from the current sources would be canceled out by the third chopper which matches dynamically the two transistors on top. There are no consequences on the signal due to the third chopper. The
benet of chopping at low impedance nodes comes
from the unlimited bandwidth of the basic amplier.
Therefore, we can chop at much higher frequencies
where the only limitation would be the charge injection residual oset. In plus, the cascode transistors
would provide low-pass ltering for the high frequency
spectral contributions coming from chopping.
In this approach, charge pumps are not needed if the
common mode voltage is well chosen and switching
is close to the supply rails. The output node, used
for Miller compensation, lters out the undesired high
frequency spectral components from switching, delivering to the output stage an oset/noise free voltage.
S1/F /k
T
F
Stherm al/S0
1
1.2
0.861
1.1
0.5
0.9
BW/fchop 0.826
0.0
0
5
10
0
f/fchop
5
10
Fig. 4. Noise PSD after chopping
CM FEEDBACK
VB2
OUT
IN
VB3
VB4
VSS
Fig. 5. Conventional choppers
VDD
VB1
VB2
OUT
IN
VB3
VSS
Fig. 6. Basic principle
IV. The Class AB output stage
The chopped amplier is divided in two parts: the
gain stage and the output stage capable of driving
low-ohmic loads. The output stage is shown in g.7
and consists of an input common-mode current source
of the pair M14 and M15, with active load, and the
class AB control.
The output transistors M35, M36 are driven in phase
from high impedance nodes. To control the output
currents, a feedback control has been chosen. A scaled
copy of the current in the output devices ows through
M33 and M37. The two copies are forced in the transistors M34 and M32. The feedback loop will enforce
the condition:
VGS34 + VGS32 = Ebiasn
(2)
If the transistors M34 and M32 are in weak inversion
and forward saturated (VDS UT ) then, the same
454
Proceedings of the ProRISC Workshop on Circuits, Systems and Signal Processing 1997
1.6m
VDD
Ibias
ID_M36
1.4m
M36
M37
1.2m
M10
M9
ID_M35
1.0m
Ebiasp
800.0u
VOUT
Ebiasn
M15
M14
600.0u
Rload
M34
400.0u
200.0u
IMIN
M32
Iin
M33
M35
IMIN
VSS
Fig. 7. Class AB output stage
condition can be rewritten as:
(mI36 ? IMIN )(mI36 ? IMIN ) =
KW2 U (E ? 2V ) 2 (3)
T
n T biasn
0.0
-60.0m
-40.0m
-20.0m
0.0
20.0m
40.0m
60.0m
V OUT
Fig. 8. The class AB control of the output transistors
the condition (2) can be rewritten as:
s
IMIN +
Ebiasn ? 2VT = 2 (mI36 ?
s
2 (mI35 ? IMIN
(5)
where n is the slope factor, UT is the thermal voltage,
V. Circuit priciple
represents the gain factor, m the scaling factor and
KW is a constant. In most cases, the constant KW The circuit consists of a chopped transconductance
stage and a class AB output stage. The nal version
can be approximated with:
of the circuit is depicted in g. 9. The dierential
KW = 20 (n ? 1)
(4) input signal is being transposed by the input chopper M3, M4, M5, M6, modulated and amplied by a
PMOS dierential pair M1 and M2 working in weak
The transistors M32 and M34 are considered identical inversion. The large bandwidth mirror M7, M8, M13
and at the same temperature. One can see that for a and M14,15 used for subtraction, is being chopped at
given temperature, the product of the left hand side low impedance nodes. This operation demodulates
terms is constant. The class AB behavior has been back the signal and modulates the noise and oset at
simulated, as shown in g.8. If only the product rule odd multiples of the chopper frequency. The cascoded
would have been implemented, the residual current is bias sources on top M43, M44, M52 and M24,25 are
not limited. Hence, if one of the transistors would dynamically matched by using a third chopper out of
conduct a lot of current, the other one works at very the signal path. The single ended signal current acts
low currents and the gate-source voltage associated as a common-mode current of the dierential pair M14
with it becomes low. This is undesirable as long as the and M15 being splitted in two branches. Due to the
circuit that drives the gates of the output transistors equal splitting, the output transistors M36 and M35
should have some 500div600mV voltage room to keep are driven in phase. The only sources of oset and
all of the transistors in saturation.
noise remains the output stage and mismatches beIn this approach there is a well dened residual cur- tween the two signal branches. There are few reasons
rent in the output transistors. The two extra cur- to desire a high gain for the transconductance stage.
rent sources IMIN subtract a small current from the For low ohmic loads, the output stage attenuates the
copied output currents coming from M33 and M37. signal up to -12dB. Hence, the open loop gain of the
This causes in all situations a shift upwards for the amplier drops down. If the transconductance stage
current owing in the output transistors. The quies- has large gain, the oset and 1/f noise generated in
cent current in the output stage is about 300A. If the output stage can be neglected. However, in modthe transistors M34 and M32 are working in strong ern processes, the output resistance of the transistors
inversion, there is also a class AB control, this time, is very low. Simple cascoding does not oer a solu-
Low-Power Low-Voltage Chopped Amplier with a New Class AB Output Stage
120/1.6
M47
120/1.6
60/1.6
M46
4/0.5
M44
4/0.5
1/10
120/1.6
120/1.6
M45
M43
M49
455
1/10
M42
M41
1/10
M40
25/1
25/1
160/0.5
M39
M38
M37
M36
2720/0.5
Fold=60
4/0.5
M50
M48
4/0.5
M51
120/1.6
60/1.6
M53
60/1.6
M24
350/1
Fold=6
M52
M0
30uA
120/1.6
120/1.6
M25
350/1
Fold=6
M22
M23
120/1.6
M20
300/1
Fold=8
300/1
Fold=8
M1
120/1.6
M21
C1
5nA
5nA
40nA
40nA
5pF
M2
M4
2/0.5
M3
2/0.5
M5
2.5pF
2/0.5
C2
2/0.5
M6
M18
M19
70/1
70/1
10uA
M17
M16
50/1
M54
50/1
M15
M14
M13
70/1
35/1
2/0.5
35/1
M10
27.6/15
M34
M9
2/0.5
10uA
M11
2/0.5
2/0.5
160/0.5
Fold=6
M32
M12
M7
105/1
M26
M8
105/1
80/1
M27
M28
10/1
10/1
M31 160/0.5 M33
Fold=6
M29 M30
80/1
M35
1000/0.5
Fold=26
80/0.5
50/150/1
Fig. 9. Circuit diagram
tion in this case. Besides, we want large swings at the
output nodes of the transconductance stage in order
VI. Simulations
to ensure that the output transistors are driven out of
saturation. A gain boosting circuit M16, M17, M22 The circuit has been simulated and the results are
and M23 has been added to the cascode transistors shown below. For a 32
load, the maximum outM18, M19, M20 and M21. It's current consumption put voltage is close to the rails within 360 millivolts.
is limited to few nanoamperes and the noise and oset of the gain boosting transistors is limited by the
large areas used. At the output node, we can go as
low as VGS 16 + VGS 18 ? VT . Because the gain boosting
1.8MHz
transistor works at low current, in weak inversion, the
gate-source voltage of M16 is close to VT and therefore
we can go as low as 600mV from the VSS rail. The
same applies to the PMOS counterparts. The gate of
the transistor M14 is connected to a constant voltage
and the dierential amplier will enforce a constant
voltage at the gate of M15. The gain boosting circuit
limits the minimum supply voltage at about 1.4V :
100.0 (LIN)
50.0
0.0
DB(Vout)
-50.0
CL =5pF
-100.0
CL =300p
-150.0
100.0m
10.0
1.0
1.0k
100.0
100.0k
10.0k
10.0M
1.0M
(LOG)
VDD;min = VGS16 + VDSsat;18 +
jVGS22 j + VDSsat;20 j
Fig. 10. Open loop gain
1.0G
100.0M
F
The open loop gain of the circuit is presented in
456
Proceedings of the ProRISC Workshop on Circuits, Systems and Signal Processing 1997
g. 10 under the condition of a heavy load of 32
jj5pF
and 32
jj300pF load capacitance, respectively. The
opamp has a gain bandwidth product of 1.8 MHz,
91 dB low frequency gain and the phase margin is 87
degree and 83 degree respectively. However, by steering the output transistors in triode region, to have a
close to rail output, instability can occur. In order
to prove that the circuit is stable in all possible situations, a transient analysis has been performed for
dierent amplitudes at the output as shown in g.11.
2.0
1.5
1.0
500.0m
VOUT
0.0
-500.0m
-1.0
-1.5
-2.0
40.0u
80.0u
120.0u
160.0u
200.0u
60.0u
100.0u
140.0u
180.0u
220.0u
T
Fig. 11. Transient response
VII. Conclusions
A chopped amplier with a new class AB stage has
been presented. It's principle is based on chopping
and dynamic element matching to reduce 1/f noise
and oset without excessive increase of the charge injection residual oset. The main goal of the circuit it
is to achieve low noise and residual osets by chopping at high frequencies. It can drive low ohmic loads
without stability problems with high gain and high
linearity, meeting specications for high end audio applications. The circuit has been designed in a 0.5m
digital CMOS process. A summary of performance
based on simulations is presented below.
AV 0Lj32
jj300pF
91 dB
GBW
1:8 MHz
THDj1k
;1kHz
> 90 dB
THDj32
;1kHz
> 83 dB
VOS;STATIC
< 1:67 mV
VOS jfchop = 10MHz < 450 V
VOS jfchop = 1kHz
< 1 Vp
VNOISE;THERMAL
37 nV/ Hz
SUPPLY
3:3 V 10%
VDD;MIN
1:4 V
P
1:56 mW
Technology
0:5m, 2PS,3AL,
CMOS
Harmonic distortion depends on the load and frequency. This has been presented in g. 12. At 1 kHz
frequency, the THD can be lower than -90 dB for a
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power,
low-voltage
chopped transconductance amplier for
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-50
-60
)Bd( DHT
1 kΩ load
32 Ω load
-70
-80
-90
-100
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
Output amplitude (Volt)
Fig. 12. THD vs. Amplitude
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[6] W.C. Wu, W.J. Helms, J.A. Kuhn and B.E. Byrkett,
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