IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 41, NO. 9, SEPTEMBER 2006
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A Low-Voltage Broadband Feedforward-Linearized
BJT Mixer
Su-Tarn Lim, Student Member, IEEE, and John R. Long, Member, IEEE
Abstract—A low-voltage, feedforward-linearized bipolar mixer
realizes an input IP3 of 14.3 dBm and an input IP2 of 54.5 dBm
at 2.4 GHz. Conversion (power) gain over the 1–6 GHz RF input
range is 12.4 0.35 dB, while the input IP3 is 13.6 1.8 dBm
over the same frequency range. The broadband mixer’s RF input
impedance varies from 60.3-j7.1 at 2.4 GHz to 57.4-j16.6
at
5.8 GHz. Measured SSB (50 ) noise figure is 18.6 dB at 2.4 GHz.
No on-chip inductors are used in the design, and the 0.14 mm2
(active area) mixer dissipates 7.2 mW from a (minimum) 1.2 V
supply.
+
+
Index Terms—Active mixer, BiCMOS, distortion cancellation,
feedforward linearization, SiGe-HBT, silicon radio frequency integrated circuits, third-order input intercept point (IIP3 ).
I. INTRODUCTION
HE popularity of cellular communications and wireless
local-area networking (WLAN) is increasing the number
of users sharing the same frequency spectrum, leading to
greater interference levels and lower data throughput because
of bit-error limitations. The number of frequency channels
available for reuse in a geographic area can be increased by
reducing cell sizes. However, this also increases the number
of base-stations, and may not be possible when users and
their wireless devices share the same office space and expect
a reasonable quality of service within 10–100 m of the same
base-station. The problem is exacerbated by new devices
using standards that share existing frequency bands and interfere with each other (e.g., IEEE-802.11a and 802.15.3a
multiband-OFDM ultrawideband). Radio frequency integrated
circuits with wide dynamic range—and in particular, higher
linearity—as well as low power consumption are needed to
satisfy the constraints imposed by new and evolving portable
wireless systems.
Fig. 1 illustrates a receiver front-end which supports
the 802.11b/g, WCDMA, and Group A of the 802.15.3a
ultrawideband (UWB) standard. Each standard is provided
with its own antenna interface, preselect bandpass filter and
low-noise amplifier (LNA) to simplify antenna interfacing and
minimize radio-frequency (RF) signal loss between the antenna
and LNA input. When all standards share the same mixer and
baseband processing circuits, the receiver circuit complexity,
parts count and current consumption are all reduced. However,
T
Manuscript received January 5, 2006; revised May 12, 2006. This work was
supported by the Philips Associated Center at DIMES (PACD).
The authors are with the Electronics Research Laboratory/DIMES, Delft University of Technology, 2628 CD Delft, The Netherlands (e-mail: s.t.lim@ewi.
tudelft.nl).
Digital Object Identifier 10.1109/JSSC.2006.880384
the mixer must be capable of operating over a wide range of
frequencies and meeting the diverse requirements imposed by
multiple standards.
The linearity of a radio receiver such as the one shown in
Fig. 1 is typically limited by the intermodulation (IM) distortion generated by the first downconverting mixer. Therefore, the
mixer must be designed so that its linearity is as high as possible given the supply current/power consumption budget (as
linearity and current consumption are usually related). On the
other hand, the noise figure target for the mixer may be relaxed
when it has sufficient RF-IF gain to suppress baseband noise.
This is becoming more feasible as the noise performance of integrated LNAs continues to improve. For example, a mixer with
12 dB conversion gain and a double sideband noise figure of
16.5 dB (or better) can meet an overall noise figure target for
the receiver of 7 dB (e.g., for 3G-WCDMA [1]) when the LNA
gain and noise figure are 18 dB and 1.5 dB, respectively. This
estimate assumes 3 dB loss both before and after the LNA from
bandselect filters and other interfacing components.
Considering other design aspects, the mixer should maintain
robust IM performance at low supply voltages (approaching 1 V
for production CMOS). On-chip inductors, which are often used
to improve linearity in a mixer, should be avoided if possible because the die area they consume (which may be on the order
of 100–1000 transistor equivalents) increases the cost of the
radio. Moreover, the mixer should be robust to processing variations and operating temperature shifts. The mixer’s RF input
impedance should also be close to 50 (broadband) to reduce
the components and cost required for a broadband impedance
matching network. However, higher impedance levels may be
chosen in applications where filtering between the LNA and
downconverting mixer—and hence a 50 RF interface—is not
required (e.g., low-IF or direct conversion receiver).
The most widely used mixer in silicon IC technology is the
Gilbert modulator [2], which consists of a differential input amplifier cascoded by a 4-transistor commutating (or quad) circuit. The RF input to the mixer is converted to current by the
input pair and then fed to the quad, which is driven by the local
oscillator signal. The absolute minimum supply voltage for a
bipolar implementation is approximately 2.7 V (two cascode
stages biased by tail current sources), making it incompatible
with the 0.13 m digital CMOS used in production today which
is powered by a 1.2 V supply.
IM distortion in the Gilbert modulator is typically limited by
the RF input transconductance stage. Improving linearity compromises the conversion gain, current consumption, or noise
figure of the mixer. For example, degeneration of the input pair
reduces the mixer conversion gain, and complicates impedance
matching by increasing the input impedance well above 50 .
0018-9200/$20.00 © 2006 IEEE
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IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 41, NO. 9, SEPTEMBER 2006
Fig. 1. Multi-standard receiver front-end block diagram.
Other input stages such as the Class-AB transconductor [3],
[4] and multi-tanh input stages [5], as well as out-of-band terminations [6], [7] for harmonic currents have been developed
to reduce the distortion and improve the dynamic range of
the mixer. However, these alternatives suffer from drawbacks
ranging from processing variations and mismatch resulting in
sub-optimal linearity (for harmonic terminations), to bandwidth
limitations giving rise to asymmetry and even-order distortion
for Class-AB and multi-tanh stages.
A new low-voltage balanced mixer circuit implemented in
BiCMOS-7HP technology [8] is presented
IBM’s 120 GHz
in this paper. Linearity (as measured by the third-order input
intercept point, or IIP ) is improved by 4 dB over previous circuit realizations through the use of feedforward cancellation of
intermodulation distortion. The circuit is inherently broadband,
so it can be applied up to 6 GHz, and beyond. The complete
circuit (including bias sources) cascodes only two transistors
between the supply rails, hence operation supply voltages as
low as 1.2 V is possible in most bipolar technologies. Inductors or transformers are not employed, resulting in a very compact design that requires little chip area. A mixer RF-IF conversion gain greater than 12 dB is realized using a modest IF
load impedance of 470 (at each IF output) and bias current
of 6 mA. In addition, the broadband RF input impedance is
close to 50 (single-ended), simplifying impedance matching
to off-chip components. The main disadvantage of the circuit is
its relatively high noise figure (17–18 dB SSB-50 ), which is
a limitation for receiver, but not for transmitter applications.
Section II of this paper briefly reviews the theory and sources
of IM distortion in bipolar amplifiers and mixers necessary
to understand the feedforward-linearized design. Section III
presents the feedforward-linearized mixer circuit, which consists of parallel path mixers with distortion equal in magnitude
but opposite in phase resulting in cancellation of the net IM
distortion at the IF output. Experimental results for a prototype
feedforward-linearized mixer are presented in Section IV,
followed by a brief summary.
II. BACKGROUND
In the conventional Gilbert mixer [2], IM distortion is generated by both the input stage (typically a differential pair) and the
mixing quad. Ignoring parasitic capacitances, the third-order intermodulation distortion present in the collector current ( ) of
the input pair depends upon the square of the differential input
voltage ( ). The intercept point (IIP ) can be expressed in
terms of input voltage (without emitter degeneration) as
, where
is the BJT thermal voltage (
), which is 104
mV at room temperature [9]. This is equivalent to an IIP of
12.9 dBm across a matched (100 ) differential input, which
and
is too low for most radio receiver applications. Since
are related, the intercept point may also be expressed as the
, which makes
ratio of AC and DC collector currents as
it clear that lowering IM distortion requires an increase in bias
current and consequently power consumption. The analysis of
distortion in a complete mixer requires computer simulation, as
the relationships between circuit variables in closed-form expressions are normally obscured by their complexity [10].
A. Feedback and Feedforward Amplifiers
Using feedback, it is also possible to design amplifier circuits
that have higher linearity, use less bias current, and maintain
acceptable noise figure by exploiting the phenomenon of distortion cancellation [11]. It can be shown that third-order IM
distortion in a common-emitter amplifier with emitter degeneration is cancelled when the product of transistor transconductance ( ) and degeneration resistance ( ) equals one-half,
or
. However, perfect cancellation is difficult to
realize because of processing variations and component mismatches, even when all devices are fabricated on the same IC
chip. Moreover, as the frequency of operation increases, other
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LIM AND LONG: A LOW-VOLTAGE BROADBAND FEEDFORWARD-LINEARIZED BJT MIXER
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Fig. 3. Schematic of modified balanced mixer switching quad.
Fig. 2. Schematic of cascade compensated (CasComp) transconductor [13].
sources of distortion (such as bias-dependent capacitors) begin
to dominate, and make the local feedback ineffective in reducing
the overall IM distortion.
Distortion cancellation can also be implemented with a
feedforward circuit topology. In the feedforward linearization
method, distortion produced by a main and second error amplifier are designed to be equal in magnitude but opposite in
phase so that they cancel each other when summed, producing
a “distortion-free” output [12]. Another advantage of the feedforward technique for RF applications is its inherent broadband
response [12].
An example of a feedforward-linearized amplifier is shown in
Fig. 2. The cascode compensation (CasComp) transconductor
[13] uses degenerated common-emitter and common-base
stages operating as a parallel-path amplifier. Distortion produced by the base-emitter junctions of input pair
is replicated across the base-emitter junctions of
.
senses the signal across
Compensating transistor pair
, amplifies it, and then add a compensating anti-phase
. The distortion magnisignal to the collector currents of
tudes are matched by adjusting the gains of the two amplifiers.
This requires appropriate selection of both bias currents and
and
feedback resistance values (e.g., 25 /20 mA for
from [13]). Sensitivity to processing
17 /10 mA for
variations and component mismatch in a CasComp stage is
relatively low, because it depends upon the ratio of the amplifier
gains and not on absolute component values.
B. Local Feedback and Mixer Linearization
Typically, an LO input many times the thermal voltage (e.g.,
200–500 mV-pk differential) is used to drive the switching quad
of a balanced mixer (e.g., see Fig. 3). This reduces the period
– ) are conducting,
of time when all four transistors (i.e.,
and as a result the quad behaves like a common-base amplifier
between the RF (input) and IF (output) over most of the LO
cycle. Therefore, one might expect intuitively that local feedback from emitter resistor would null or cancel the IM distortion in this mixer, similar to the distortion nulling observed in
a common-base stage with local feedback. This is indeed possible, however, the IM distortion generated by the quad is a function of LO amplitude, transistor bias current and the transistor’s
emitter area in general [14].
As an example, the simulated third-order intermodulation distortion (IM3) phase (relative to the fundamental components)
and IIP of the IF output current are plotted in Fig. 4 for the circuit of Fig. 3 as the feedback resistor varies. The transistors
in the switching quad are biased at 4 mA and driven by a 235
mV-pk sinusoidal LO for the simulations. The results show that
as the feedback increases (i.e., an increase in ), the IM3 components fall to zero (or null), thereby maximizing the mixer IIP .
For example, the peak IIP is 9.8 dBm at an RF input frequency
of 250 MHz. The behavior at higher frequencies is similar, but
parasitic capacitances lower the maximum IIP and change the
conditions required to null the IM distortion (e.g., peak IIP occurs at 163 instead of 172 when the frequency increases
from 0.25 to 5.75 GHz).
The effects of this process variation in the mixer IIP were
analyzed using Monte Carlo analysis in Cadence Spectre™.
mA,
The nominal operating conditions of
, and
mV-pk (single-ended) for the mixer of
Fig. 3 produced the desired distortion cancellation and an IIP
of 7.93 dBm. The IIP (for constant
) results obtained from
the Monte Carlo simulations are plotted as a histogram in Fig. 5.
IIP for the linearized mixer follows a Gaussian distribution
with a mean of 8.30 dBm and standard deviation of 0.46 dB.
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IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 41, NO. 9, SEPTEMBER 2006
Fig. 5. Variation in IIP due to process variation (i.e., fast, normal, slow
process) and mismatch from 500 Monte Carlo simulations of a single mixer
with distortion nulling.
Fig. 4. Phase of IM3 distortion and IIP versus degeneration resistance for the
= 4 mA and V = 235 mV-pk).
circuit of Fig. 3 (I
III. FEEDFORWARD DISTORTION COMPENSATION
Given the previous example of the CasComp stage and its
utility as a linear broadband transconductor, feedforward-linearization of a balanced mixer was attempted in an effort to
reduce IM distortion and circuit sensitivity to processing variations and component mismatch. A feedforward scheme used
to linearize an amplifier cannot be adapted directly to a mixer
because of the frequency translation in the signal path between
RF input and IF output. In a classical feedforward linearization
scheme, the error is derived from the output current (e.g., from
and
in Fig. 2). If this were atthe collector current of
tempted in a mixer (i.e., at the IF output), then the original signal
(i.e., from the RF input) could not be used to derive an estimate
of the distortion without doing another frequency translation.
This problem is avoided when the compensating signal is developed independent of the output by using a second, parallel
signal path. The feedforward-linearized mixer of Fig. 6 uses two
switching quads operating in parallel, where each mixer is producing equal amplitude—but anti-phase—IM distortion components. Local feedback does not affect the phase of the fundamental at the IF, so when the outputs from the two switching
quads are connected in parallel, the fundamental components
add in-phase giving a larger output. However, cancellation of
the third-order IM distortion components occurs when the IM3
products produced by each mixer are anti-phase. This condition
is set by appropriate selection of the bias currents and degenerand
in Fig. 6) for each mixer.
ation resistances (e.g.,
Degeneration resistors add thermal noise to the RF input
giving the mixer a relatively high noise figure. Inductors could
be substituted to provide the necessary degeneration impedance
because they do not (ideally) add thermal noise. However,
inductive degeneration restricts the range of possible bias currents and device sizes where the phase shift between the two RF
paths can be matched for IM cancellation, thereby restricting
the RF input frequency range. In addition, resistors consume
significantly less area than on-chip inductors. In an integrated
receiver, the output impedance of the driving stage could also
be used to degenerate the mixers (see Section V). Since the
emphasis in this design was proof of concept for a broadband
mixer with improved linearity, degeneration resistors are used
in the following design.
For validation of the feedforward-compensation concept, all
transistors in the mixer of Fig. 6 were chosen to be 5 0.2 m
and the total bias current limited to 10 mA. Degeneration resistors were restricted to between 50 and 150 to limit the
thermal noise contribution and voltage drop across them. Both
mixers are (nominally) driven by a sinusoidal LO of 166 mV-pk
at each base terminal, which ensures that the quads switch completely using an LO that is easily sourced from an on-chip oscillator in a wireless application.
Feedforward compensation is realized by matching the IM3
magnitudes for mixers A and B. In simulation it was found that
and
, respectively, at a total bias
75 and 120 for
current of 3 mA for mixer A and 5 mA for mixer B set this
condition. This selection of degeneration resistors and bias currents gives approximately the same (50 ) input impedance for
the two mixers. A differential IF load of 470 and a 2.2 V
supply were used for these simulations. The fundamental and
IM3 output powers versus the input power for the feedforwardlinearized mixer as well as each mixer individually are plotted
GHz and IF
MHz). There is an
in Fig. 7 (RF
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LIM AND LONG: A LOW-VOLTAGE BROADBAND FEEDFORWARD-LINEARIZED BJT MIXER
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Fig. 6. Feedforward-linearized mixer schematic.
Fig. 7. Simulated RF to IF output versus input power for fundamental and third= 5:75 GHz, f = 5:60 GHz, 470 order intermodulation distortion (f
IF load, and 2.2 V supply).
improvement of more than 7 dB in IIP for the feedforward-lindBm) compared to either
earized mixer (overall IIP
mixer operated stand-alone. These simulations also show that
the individual mixers and the feedforward-linearized mixer have
approximately the same gain compression point. This is expected, because local feedback only affects distortion generated
by the nonlinear transconductance of transistor. It does not affect distortion generated by waveform clipping, which is typically the source of gain compression.
Linearity and conversion gain for the feedforward-linearized
mixer from a simulation over the RF input frequency range
Fig. 8. Simulated conversion gain and third-order intercept point versus frequency (470 IF load, 8 mA total bias current at 2.2 V supply).
from 1–10 GHz are plotted in Fig. 8. The conversion gain is
approximately 12.3 dB at 1 GHz, and decreases gradually with
increasing frequency to 11.7 dB at 10 GHz. This slight drop
in conversion gain is expected as parasitic capacitance dominates the high frequency response. When operated at a lower
voltage supply, voltage/current clipping at the IF output lowers
dB compression point. However, intermodulation distorthe
tion (dominated by nonlinearity of the active devices) remains
essentially the same, so an IIP similar to what is achieved at
higher supply voltages is expected.
Similarly, the third-order output-referred intercept point
(OIP ) is highest at lower frequencies; it is 31.6 dBm at 1
GHz and drops to 24.3 dBm at 10 GHz. Simulations predict
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Fig. 9. Variation in IIP due to process (i.e., fast, normal, slow) and mismatch
from 500 Monte Carlo simulations of the feedforward-linearized mixer.
that gain and phase mismatch in IM3 components generated
by the two mixers are 0.2 dB and 1 at 1 GHz, but increase to
1.7 dB and 13 at 10 GHz. Distortion cancellation is affected
as parasitic capacitances play a more significant role in the
mixer’s response at higher frequencies.
A single Gilbert mixer operating at 8 mA bias current (3 V
supply) is also simulated for comparison. The same LO swing
(166 mV-pk) and IF load (470 ) are used. The conversion gain
of the mixer is adjusted via a degeneration inductor to equal
the feedforward mixer at 5.75 GHz (RF). Simulation predicts
that the single mixer with a 10 nH degeneration inductor has a
conversion gain and IIP of 12.2 dB and 11.56 dBm (
GHz), respectively. Linearity is still 3 dB lower than the
GHz, and detefeedforward-linearized mixer at
riorates further as the RF decreases below 5.75 GHz. For example, at 2 GHz, simulation predicts a conversion gain and IIP
of 20.4 dB and 1.7 dBm, respectively, for the single mixer
at 8 mA with the same 10 nH degeneration. On the other hand,
the simulated IIP for the feedforward-linearized mixer is better
than 12.6 dBm for frequencies ranging from 1–10 GHz.
Results of Monte Carlo simulation of the complete feedformA,
ward-linearized mixer are plotted in Fig. 9 (
and
mA). The mean and standard deviation are
14.4 dBm and 0.75 dB, respectively. Over 78% of the simulated results for IIP fall between 14 and 15 dBm. Unlike the
mixer with feedback distortion nulling (see Fig. 5), the distribution is not Gaussian. This suggests that although the highest IIP
cannot be obtained for all samples, high linearity (i.e., within
1–2 dB of the max.) is still achieved for the majority of the simulation cases.
The second-order intermodulation distortion (IM2) of the
feedforward-linearized mixer (important in low/zero-IF applications) is primary determined by component matching and
the physical layout of the circuit. Monte Carlo simulation
results for the mixer IIP (refer to Fig. 10) are Gaussian with
a mean and standard deviation of 67.15 dBm and 7.48 dB,
respectively. A similar analysis on the circuit of Fig. 3 designed for IM3 nulling gave similar results with a mean IIP of
Fig. 10. Variation in IIP due to process (i.e., fast, normal, slow) and mismatch
from 500 Monte Carlo simulations of the feedforward-linearized mixer compared to a single doubly balanced mixer with distortion nulling.
66.13 dBm and a standard deviation of 5.94 dB, respectively.
Fewer components give a slightly smaller spread for a single
mixer. Even-order distortion and IIP may be reduced further
by correcting for even-order mismatches using a scheme such
as that proposed in [6].
IV. EXPERIMENTAL RESULTS
The prototype mixer was characterized over the frequency
range from 1 to 6 GHz. This covers the spectrum currently occupied by commercial wireless applications (including Groups
A and B of IEEE 802.15.3a UWB). The experimental test set-up
is shown in Fig. 11. Differential RF and LO signals are derived
from single-ended generators using microwave power dividers
and 0 /180 hybrid baluns. At the RF input, two differential signals are used such that the individual mixers can be biased individually. The mixer was configured as a downconverter for
testing, so 5 pF filtering capacitors (included on-chip) are used
to filter the unwanted (upper) sideband at the IF output. The
)
mixer IF is AC coupled to two 470 single-ended loads (
and balun-coupled to single-ended test instruments (e.g., a spectrum analyzer). The prototype mixer including the I/O pads occupies 1.10 1.24 mm while the active circuitry of the mixer
occupies just 80 180 m , as seen from the chip micrograph
at the center of Fig. 11. The supply voltage was varied between
1.2 V and 2.2 V in testing, although most of the characterization
work was performed using the (nominal) 2.2 V supply. The measured IM3 performance is expected to improve when the mixer
is used in an integrated transceiver design (i.e., RF integrated
with LO and IF/baseband stages), because gain and phase mismatch in the RF, LO and IF paths inherent in the experimental
setup can be minimized with an integrated solution.
The bias and LO input power that resulted in the highest linearity (IM3) was first determined experimentally from a conventional two-tone test. In each case, 2.4 GHz RF signals spaced
100 Hz apart and with an input amplitude of 13 dBm were
downconverted to a 50 kHz IF when the mixer was driven by a
differential LO power of 1.6 dBm (note: these are the powers
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LIM AND LONG: A LOW-VOLTAGE BROADBAND FEEDFORWARD-LINEARIZED BJT MIXER
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Fig. 11. Test setup for experimental characterization.
Fig. 12. Measured difference between fundamental tones and intermodulation
,I
) currents.
distortion ( IM ) at various (I
1 3
at the mixer input ports corrected for cable losses, etc.). Close
spacing of the RF input tones was used so that the tones could be
resolved more accurately at the IF. The largest measured dB-difference between the fundamental and third-order intermodulation distortion products (i.e., the IM ) is plotted in Fig. 12 for
various combinations of bias current in each quad. The ordered
pair of numbers next to each data point represents the bias currents flowing in mixers A and B (from Fig. 6), respectively.
From these results, it can be seen that higher IM is consistently achieved by increasing the bias current (as for other
active mixers), and that the IIP improves by 11 dB (note that
IM ) as the total bias current increases from
IIP
3 mA to 10 mA. However, for total bias current combinations
smaller than 4 mA and larger than 10 mA, each mixer’s IM3
distortion tends to add in-phase at the IF rather than cancel, resulting in a lower IM . For example, the bias current combination (3 mA, 3 mA) in Fig. 12 has a IM of 47 dB that is
comparable to higher bias current combinations such as (3 mA,
4 mA) and (5 mA, 3 mA), but with almost identical IF power at
the fundamental (i.e., between 4 and 4.2 dBm). Thus, there is an
optimum partitioning of the total bias current for the best IM3
performance, and operation close to the shaded area indicated
in Fig. 12 is desirable because less current (and lower power) is
consumed while having a relatively large IM .
For portable applications, both low power dissipation and
high IM is desired. The bias combination (3 mA, 3 mA)
from a 2.2 V supply was therefore selected as a compromise
between low intermodulation distortion and total current consumption for further characterization. At this bias current, the
measured fundamental and IM3 output powers at the IF versus
kHz and
the RF input power are plotted in Fig. 13 (IF
RF
GHz). An LO input power of 2.6 dBm was used for
the measurements. The measured conversion gain and IIP are
12.6 dBm and 14.3 dB, respectively. In this case, the measured
output-referred intercept point (OIP ) is over 10 dB higher than
we previously reported in [15], where a 50 IF load was used.
It is likely that clipping of the IF waveform (i.e., current limiting) is responsible for the rise in distortion observed for low IF
load impedances. The conversion gain can be easily adjusted by
in Fig. 11), however, its
modifying the IF load impedance (
effect on the linearity and IF bandwidth must also be considered
in any design choice. As seen from the transfer curves of Fig. 13,
the IF output of the feedforward mixer is linear up to an RF
input power of approximately 4 dBm, while the IM3 distortion power expands for input powers larger than about 5 dBm.
The measured conversion gain, IIP and OIP of the
feedforward-linearized mixer in the frequency ranges from
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IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 41, NO. 9, SEPTEMBER 2006
Fig. 13. Measured fundamental and third-order IM distortion versus input
power.
Fig. 14. Measured conversion gain and IP from 1–6 GHz (V
I
= 3 mA, and I
= 3 mA).
= 2:2 V ,
1.25–2.4 GHz and 4–6 GHz are compared in Fig. 14 (note
that frequencies between 2.5 and 4 GHz could not be covered
with the present test set-up). The measured values compare
favorably with those predicted from simulation. For example,
the simulated conversion gain and IIP at 5.7 GHz are 12.1 dB
and 14.4 dBm, respectively, compared to the measured values
of 12.6 dB and 13.3 dBm. The conversion gain is relatively
constant at approximately 12.5 dB between 1 and 6 GHz. The
IIP is better than 13 dBm for operating frequencies between
2.1 and 5.8 GHz. IIP varies by 3.5 dB across the measured
dBm).
frequency range and is lowest at 1.25 GHz (IIP
Similar performance can be expected over the entire 1–10 GHz
RF operating range.
Matching between the various single-ended paths in the test
set-up is difficult to achieve in-practice. For example, matching
the amplitudes of all four mixer RF inputs and ensuring 180
phase difference between signals at each RF input pair over
Fig. 15. Measured conversion gain and IIP
= 3 mA and I
= 3 mA).
(I
at various supply voltages
frequency, would require extensive calibration and characterization work and extensive trimming. However, the effects of
amplitude mismatch, phase error and phase mismatch are easily
studied in simulation. In this way, the differences observed between the measurements and the nominal simulation results may
be accounted for.
The measured amplitude and phase errors for the external
passive components used in the experimental set-up (e.g., power
splitters, 0 /180 hybrids, cables, etc.) are frequency dependent,
dB difference in amplitude, and up to
and can cause up to
deviation from the nominal phase condition. Simulations
predict that such errors cause the conversion gain to vary by
dB, so the conversion gain is relatively insensitive to
up to
input amplitude and phase errors. Linearity (enhanced by IM3
feedforward cancellation) is more sensitive to amplitude and/or
phase variations at the RF inputs. The simulations show that
GHz and up to 8 dB at
IIP degrades by up to 5 dB at
GHz. More consistent performance (i.e., similar to that
predicted in Fig. 8) could therefore be expected when the mixer
is integrated with a low-noise preamplifier and LO synthesizer
on the same chip. This would reduce the amplitude variation
and phase mismatch with frequency at the RF and LO inputs introduced by the external components used to test the mixer (as
predicted by simulation of such variation and mismatch).
With only the mixing quads and bias sources connected between the supply rails (i.e., two transistors in series), the feedforward mixer is well-suited to low-voltage operation. The conversion gain and IIP are plotted in Fig. 15 when operating at
2.4 GHz RF from a 1.2 V (minimum) supply. Again, the bias
current combination of (3 mA, 3 mA) was used for these measurements. The variation in gain and distortion is minimal due
to the decrease in supply voltage. It should also be noted that
lower voltage operation (down to approx. 1 V) is possible if the
bias current sources were removed and the supply current regulated using the LO common-mode (i.e., bias) voltage. The IIP
varies by just 0.6 dB as the supply voltage drops from 2.2 V
to 1.2 V.
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LIM AND LONG: A LOW-VOLTAGE BROADBAND FEEDFORWARD-LINEARIZED BJT MIXER
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TABLE I
PERFORMANCE SUMMARY AND COMPARISON
The even-order distortion (i.e., IIP ) was also measured for
one sample at 2.4, 5.3, and 5.8 GHz, where it is 54.5, 36.6, and
36.3 dBm, respectively. These values are lower than predicted
by simulations (i.e., 67 dBm at 5.8 GHz). It is likely that the
difference is caused by mismatching in amplitude and phase of
the signals in the RF and LO paths at the inputs to the mixer test
circuit (mostly determined by mismatch in the external hybrid
baluns and cables). Again, better performance could be expected
from a fully integrated solution, where these sources of error are
lower and are more tightly controlled.
Table I summarizes the measured and simulated results. The
data indicates that optimal linearity is achieved when higher LO
power and lower bias current are used compared to simulations
(i.e., 3 dB lower LO power and 2 mA less current). The measured conversion gain is also slightly larger than predicted by
simulation (by 0.8 dB). The highest linearity observed in the
measurements and simulations are comparable, but occur at different LO input powers and bias currents as noted above, which
is likely due to processing variations and imperfections in the
experimental test setup (e.g., phase inaccuracy of the 0 /180
hybrids).
Two other bipolar mixers reported in the recent literature, one
developed for receiver base-stations [16] and a second mixer
that also uses feedforward-linearization [17], but only in the
RF input stage (i.e., the input diff pair of a Gilbert mixer) are
included in Table I. Linearity of the mixer developed in this
work (i.e., output third-order intercept point) is comparable to
the receiver base-station mixer from [16]. Comparing output
intercept points (OIP ) removes the effect of conversion gain,
which varies widely among the mixers listed in the table. The
LT5527 mixer described in [16] uses on-chip transformers at
the RF and LO inputs for input coupling and matching purposes. It is designed for use in 3G cellular telecommunication
base-stations, where the electronics are not portable or powered
from a battery, so current consumption and supply voltage are
not important design constraints. As shown in Table I, the feedforward-linearized mixer offers comparable linearity (from the
OIP ) but with a current consumption of 6 mA for the core of
the mixer compared to 78 mA (total, including LO buffers) from
a 5 V supply for the LT5527. The feedforward-linearized design brings higher linearity to portable applications where current consumption and operating time on a fresh battery charge
are important. Further power savings are possible for the feedforward design if only one of the two mixers is activated in response to less demanding operating conditions using an adaptive
power management scheme.
The noise figure of the feedforward design presented in this
paper is compromised by the use of resistive degeneration at
the RF input, which adds thermal noise. However, this also allows the mixer to work over a wide range of frequencies (from
baseband up to 6 GHz), whereas the narrowband RF and LO inputs of the LT5527 chip restrict its operating range to between
400 MHz and 3.7 GHz. The (total) RF input impedance of the
mixer is close to 50 (57.4-j16.6 at 5.8 GHz) across most of
the 1–6 GHz band, thereby simplifying impedance matching to
other components (e.g., an image-reject filter between LNA and
mixer). The deleterious effects of high noise figure for the feedforward-linearized mixer are mitigated somewhat by the higher
conversion gain of the active mixer, as described in Section I
of this paper. However, the feedforward mixer would require
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2186
IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 41, NO. 9, SEPTEMBER 2006
VI. CONCLUSION
Fig. 16. RF input degeneration using impedances from the preceding stage.
A broadband feedforward-linearized double-balanced
bipolar mixer topology was demonstrated. Outputs from two
modified switching quads are connected such that fundamental
components add but IM3 distortion in their output currents cancels, thereby resulting in a higher mixer IIP . Measured input
14.3 dBm and an input IP of 54.5 dBm at 2.4 GHz.
IP
Conversion (power) gain over the 1–6 GHz RF input range is
12.4 0.35 dB with an input impedance close to 50 . On-chip
inductors are not used in the design, and the 0.14 mm (active
area) mixer draws just 6 mA from a (min.) 1.2 V supply, while
exhibiting an IIP better than 13 dBm across the frequency
bands commonly used for wireless telephony and high-speed
local-area networking applications.
ACKNOWLEDGMENT
greater noise suppression from a low-noise preamplifier in order
to realize a given overall receiver noise figure compared to that
of the other two mixers listed in Table I.
Degradation of the distortion cancellation and IM3 performance due to minor variation in bias current and LO power
was also investigated. Each mixer’s bias current was varied by
0.5 mA ( 15%), which resulted in a maximal drop of 3.6 dB
in IIP . Similarly, the maximum change observed in IIP was
6.7 dB for a 2 dB variation in the LO input power.
When compared with the mixer reported in [17], which uses
a feedforward transconductor preceding the mixing quad, the
feedforward-linearized mixer described in this paper demonstrates 4 dB improvement in IIP (8 dB improvement in OIP )
and 18 mW lower power dissipation. Moreover, feedforward
cancellation using resistive degeneration does not require the
phase-shifter implemented in [17] and requires less chip area to
implement.
V. LOWERING THE NOISE FIGURE
The relatively high noise figure of the feedforward-linearized
mixer is an impediment to its use in many RF receiver applications. Resistive degeneration at the mixer RF inputs is needed
to implement feedforward cancellation of IM distortion over a
wide range of frequency. However, thermal noise produced by
and
in Fig. 6) is among the
the degeneration resistors (
largest contributors to the total mixer noise. In order to improve
the noise figure, these resistors should be either reduced or eliminated entirely.
One way to achieve this is to use the output impedance of the
RF driving stage to degenerate the mixers. As seen in Fig. 16,
the RF sources for the mixers may be represented by Thevenin
) with a series output resisequivalent voltage sources (
tance (
). The desired values for
and
can be realized in numerous ways, with one possibility
at RF being the use of transformers with different turn ratios
(for narrowband applications) to reflect the desired impedance
levels at the mixer inputs. A driving stage having a constant
output impedance value over frequency (e.g., amplifier with
shunt output feedback) can also be used to synthesized the
and
for broadband applications. In both
desired
these cases the noise figure is improved, as
and
are now included as a part of the source impedance.
The authors acknowledge technical support for testing provided by L. van Schie at TU Delft. Fabrication and technology
access was facilitated by A. Joseph and D. Harame at IBM
Microelectronics, Burlington, VT.
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[11] K. Mayaram and D. O. Pederson, Analog Integrated Circuits for Communication. Norwell, MA: Kluwer, 1991.
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LIM AND LONG: A LOW-VOLTAGE BROADBAND FEEDFORWARD-LINEARIZED BJT MIXER
[15] S. T. Lim and J. R. Long, “A low-voltage feedforward-linearized
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[16] 400 MHz to 3.7 GHz High Signal Downconverting Mixer, Linear Technology, LT5527 [Online]. Available: http://www.linear.com/pc/productDetail.do?navId=H0,C1,C1011,C1097,P10772
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+
Su-Tarn Lim (S’05) received the B.Sc. and M.Sc.
degrees in electrical engineering from the University
of Alberta, Canada, in 1999 and 2001, respectively.
Currently, he is pursuing the Ph.D. degree at the Delft
University of Technology, Delft, The Netherlands.
His interests are in the areas of analog and RF circuit design.
2187
John R. Long (S’77–A’78–M’83) received the B.Sc.
degree in electrical engineering from the University
of Calgary, Calgary, AB, Canada, in 1984, and the
M.Eng. and Ph.D. degrees in electronics from Carleton University, Ottawa, ON, Canada, in 1992 and
1996, respectively.
He was employed for 10 years by Bell-Northern
Research, Ottawa (now Nortel) involved in the design
of ASICs for Gbit/s fibre-optic transmission systems,
and for 5 years as an Assistant and then Associate
Professor at the University of Toronto, Canada. He
joined the faculty at the Delft University of Technology, Delft, The Netherlands,
in January 2002 as Chair of the Electronics Research Laboratory. His current
research interests include low-power transceiver circuitry for highly integrated
wireless applications, and electronics design for high-speed data communications systems.
Prof. Long currently serves on the Program Committees of the IEEE International Solid-State Circuits Conference (ISSCC), the European Solid-State Circuits Conference (ESSCIRC), the IEEE Bipolar/BiCMOS Circuits and Technology Meeting (BCTM), and the European Microwave Conference. He is a
former Associate Editor of the IEEE JOURNAL OF SOLID-STATE CIRCUITS. He
received the NSERC Doctoral Prize and Douglas R. Colton and Governor General’s Medals for research excellence, and Best Paper Awards from ISSCC 2000
and IEEE BCTM 2003.
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