A WIRELESS ELECTRICAL STIMULATION SYSTEM
FOR WOUND HEALING THERAPY
WITH BIPHASIC HIGH-VOLTAGE PULSED CURRENT OUTPUT
by
DANIEL STEVEN HOWE
Submitted in partial fulfillment of the requirements
For the degree of Doctor of Philosophy
Dissertation Adviser: Dr. Steven Garverick
Department of Electrical Engineering and Computer Science
CASE WESTERN RESERVE UNIVERSITY
May, 2013
CASE WESTERN RESERVE UNIVERSITY
SCHOOL OF GRADUATE STUDIES
We hereby approve the thesis/dissertation of
Daniel Steven Howe
candidate for the
Doctor of Philosophy
(signed)
Steven L. Garverick
degree *.
(chair of the committee)
Kath M. Bogie
Pedram Mohseni
Christian A. Zorman
(date)
03/15/2013
*We also certify that written approval has been obtained for any
proprietary material contained therein.
ii
To my wife and family.
iii
Contents
List of Tables .............................................................................................................. 10 List of Figures ............................................................................................................. 11 Acknowledgements ..................................................................................................... 16 List of Abbreviations and Common Signal Names .................................................... 19 Abstract ....................................................................................................................... 21 1 Introduction ........................................................................................................ 23 1.1 Clinical Motivation ..................................................................................... 23 1.2 Electrotherapy and Wound Healing ............................................................ 24 1.3 Previous Wound Healing Electrotherapy Studies ....................................... 25 1.3.1 Current research limitations .................................................................... 26 1.4 Existing electrical stimulation therapies for wound care ............................ 27 1.4.1 Microcurrent devices ............................................................................... 27 1.4.2 TENS/ pain management devices ........................................................... 28 1.4.3 Research Device: DS4 ............................................................................. 28 1.5 Previous work ............................................................................................. 30 1.5.1 Surface Stimulation Device (SSD) .......................................................... 30 1.6 2 Research Objective ..................................................................................... 31 Modular Surface Stimulator for Rodent Studies (MSS) .................................... 33 2.1 Research Study Overview ........................................................................... 33 iv
2.1.1 Battery review ......................................................................................... 36 2.1.2 VDD Step-up converter ........................................................................... 37 2.1.3 Microcontroller........................................................................................ 38 2.2 Boost Converter Circuit .............................................................................. 38 2.2.1 Battery Limitations .................................................................................. 39 2.2.2 Operational Considerations ..................................................................... 39 2.2.3 Critical Component Selection ................................................................. 40 2.2.4 Output capability ..................................................................................... 41 2.2.5 Controller firmware ................................................................................. 42 2.3 Stimulator Circuits ...................................................................................... 45 2.4 Experimental Results .................................................................................. 46 2.4.1 Modular Surface Stimulator Version 1(MSS) ......................................... 47 2.4.2 Stimulator Performance........................................................................... 48 2.4.3 Boost converter efficiency ...................................................................... 50 2.5 3 Summary ..................................................................................................... 54 Application-Specific Integrated Circuit ............................................................. 55 3.1 Functional Requirements ............................................................................ 55 3.2 Foundry Process Features ........................................................................... 56 3.3 High-Voltage Boost Converter ................................................................... 56 3.3.1 Step-up Topology .................................................................................... 57 v
3.3.2 Control loop operation ............................................................................ 59 3.3.3 Inductor current sense ............................................................................. 61 3.3.4 Current sense amplifier ........................................................................... 64 3.3.5 Current sense comparator ........................................................................ 65 3.3.6 NMOS Switch ......................................................................................... 69 3.3.7 Output Rectifier ....................................................................................... 69 3.3.8 Output capacitor ...................................................................................... 70 3.3.9 Output Voltage Sense .............................................................................. 71 3.4 Stimulator .................................................................................................... 72 3.4.1 Architecture ............................................................................................. 72 3.4.2 High-side switches .................................................................................. 73 3.4.3 Low-side DAC pair ................................................................................. 77 3.5 Support circuits ........................................................................................... 80 3.5.1 Bias generation and distribution .............................................................. 80 3.5.2 SPI Serial Port ......................................................................................... 81 3.5.3 Pad Frame, ESD Protection, and Guard rings ......................................... 82 3.6 ASIC Test Results ....................................................................................... 83 3.6.1 Introduction ............................................................................................. 83 3.6.2 MSS ASIC Test Setup ............................................................................. 85 3.6.3 Integrated Circuit Edits Using Focused Ion Beam .................................. 85 vi
3.6.4 Bias circuit............................................................................................... 86 3.6.5 Boost converter operation ....................................................................... 87 3.6.6 Stimulator ................................................................................................ 89 3.6.7 Summary ................................................................................................. 95 4 Wireless Stimulation Bandage and System for a Large Animal Model ............ 97 4.1 Large Infected Wound Study ...................................................................... 97 4.2 Bandage Substrate ..................................................................................... 100 4.2.1 Bandage Electrode characterization ...................................................... 101 4.3 Battery Selection ....................................................................................... 103 4.4 Stimulator Module PCB ............................................................................ 104 4.4.1 Wireless communication protocol and processor selection .................. 104 4.5 Firmware ................................................................................................... 105 4.5.1 Stimulation State machine ..................................................................... 106 4.5.2 Bluetooth Interface ................................................................................ 109 4.6 Device assembly ....................................................................................... 111 4.6.1 MSS3 Stimulator Module Assembly..................................................... 111 4.6.2 Device Sterilization ............................................................................... 113 4.6.3 Intra-Operative Bandage Assembly ...................................................... 114 4.7 Wireless Base Station ............................................................................... 115 4.7.1 Base station construction ....................................................................... 116 vii
4.7.2 Base station operation ........................................................................... 118 4.8 Benchtop Testing ...................................................................................... 119 4.8.1 ASIC validation ..................................................................................... 120 4.8.2 Wireless Communication ...................................................................... 123 4.8.3 Operating current test ............................................................................ 124 4.9 Clinical Validation .................................................................................... 124 4.9.1 Method .................................................................................................. 124 4.9.2 Bandage conductivity test ..................................................................... 125 4.9.3 Acute stimulation test ............................................................................ 126 4.10 5 Summary ................................................................................................... 128 Conclusions and Future Work .......................................................................... 129 5.1 Achievements ............................................................................................ 129 5.2 Future work ............................................................................................... 131 Appendix A: Supplement Information for Chapter 2 (MSS2)................................. 133 A.1: MSS2 Schematic ........................................................................................... 133 Appendix B: Supplemental Information for Chapter 3 (ASIC) ................................ 134 B.1 MSS ASIC Test Fixture Schematic ................................................................ 134 Appendix C: Supplemental Information for Chapter 4 (MSS3) ............................... 135 C.1: MSS3 Module Schematic .............................................................................. 135 C.2 Stimulation Bandage GATT profile ............................................................... 136 viii
C.3 MSS3 Parameter programming procedure using TI BLE Device Monitor .... 137 C.4 Base station Arduino programming procedure ............................................... 139 C.5 Base station BLE112 programming procedure .............................................. 140 C.6 MSS3 CC2541 programming procedure ........................................................ 141 References ................................................................................................................. 142 ix
List of Tables
Table 1-1: Specifications of a portable electrical stimulation unit. ............................ 29 Table 2-1: Design requirements for small-animal device ........................................... 35 Table 2-2: Review of battery chemistries in coin/button form factor ......................... 36 Table 3-1. Functional requirements for ASIC. ........................................................... 55 Table 3-2. Transistor sizes for current sense comparator circuit. ............................... 66 Table 3-3. Comparison of on-chip and off-chip boost converter switch components. 69 Table 3-4. Key characteristics of the BAT46W schottky diode [35]. ........................ 70 Table 3-5. Device sizes for bias circuit. ...................................................................... 81 Table 3-6. List of MSS ASIC bond pads and their function....................................... 84 Table 3-7: Comparison of simulated and measured bias circuit voltages. ................. 86 Table 3-8: Measured performance of the boost converter current sense circuits. ...... 88 Table 3-9: Summary of MSS ASIC circuit functionality ........................................... 96 Table 4-1: Clinical Requirements for Infected Wound Study Device ........................ 98 Table 4-2: Survey of high-capacity batteries for long-duration devices................... 103 Table 4-3: A review of microcontrollers with built-in 2.4 GHz radios. ................... 105 Table 4-4: Stimulator control parameters available through Bluetooth wireless link.
......................................................................................................................................... 111 Table 4-5: Stimulation Parameters used in MSS3 stimulator demonstration. .......... 122 Table 4-6: Measured operating current and projected device lifetime for several
compliance voltage levels. .............................................................................................. 124 Table 5-1: Electrotherapy Device Summary............................................................. 131 x
List of Figures
Figure 1-1. Example of a chronic wound on a foot [7]. .............................................. 23 Figure 1-2: The Vomaris Procellera(R) microcurrent wound dressing. ....................... 27 Figure 1-3. Example of a TENS portable stimulator (from [15]). .............................. 28 Figure 1-4: DigiTimer Biphasic stimulator for benchtop testing................................ 29 Figure 1-5: Surface Stimulation Device (SSD), from [16]. ........................................ 30 Figure 1-6: SSD attached to rabbit ear ........................................................................ 31 Figure 2-1: Block diagram of the proposed MSS consisting of a flexible printed
bandage and a reusable stimulator module. ...................................................................... 33 Figure 2-2: Rat wound model used with MSS ............................................................ 34 Figure 2-3. Graphical representation of the programmable stimulation waveform
parameters that cover a range of potentially therapeutic levels. ....................................... 35 Figure 2-4: Comparison of battery energy density for rechargeable and nonrechargeable chemistries (from [19]). ............................................................................... 37 Figure 2-5: Schematic diagram of the boost converter circuit using discrete
components for HV circuitry and microcontroller for the control loop............................ 40 Figure 2-6: Plot of the charging time vs. output voltage setpoints for several battery
voltages. ............................................................................................................................ 44 Figure 2-7. Schematic diagram showing the high-voltage stimulator circuit. ............ 45 Fig. 2-8. Schematic diagram of the high-side driver that controls PMOS switches with
a constant VSG across a 90 V supply range. ...................................................................... 46 Figure 2-9: Modular Surface Stimulator (MSS1) Version 1 Device .......................... 47 xi
Fig. 2-10. Plot of output pulses for several load resistors. .......................................... 49 Figure 2-11: Measured maximum pulse frequency vs. voltage setpoint with various
supply voltages.................................................................................................................. 50 Figure 2-12. Measured MSS1 Boost Converter Charging Efficiency for Selected
Supply Voltages. ............................................................................................................... 51 Figure 2-13: Measured MSS stimulator efficiency for selected boost capacitor
voltages. ............................................................................................................................ 52 Fig. 2-14. Measured battery lifetime versus stimulation voltage for selected output
resistances. Pulse rate is 25 Hz, pulse width is 200 µs, and stimulation interval is 10
min/hr. ............................................................................................................................... 53 Figure 3-1: NDMOS floating transistor in I2T100 process (modified from [26]). .... 56 Figure 3-2: Dickson Charge Pump Voltage Conversion Technique (from [27]) ....... 57 Figure 3-3. Representative schematic of the flyback converter architecture. ............. 58 Figure 3-4. Boost converter topology. ........................................................................ 59 Figure 3-5. Idealized inner-loop current control for the boost converter. .................. 60 Figure 3-6. MSS ASIC-based boost converter circuit. ............................................... 61 Figure 3-7. Inductor current sense method across shorting NMOS............................ 62 Figure 3-8. Inductor current sense method using battery-side PMOS. ....................... 63 Figure 3-9. Inductor current sense using off-chip resistor. ......................................... 63 Figure 3-10. Differential Amplifier used as inductor current sense amplifier. ........... 64 Figure 3-11. Schematic diagram of current sense comparator circuit. ....................... 66 Figure 3-12. Schematic diagram of boost converter output voltage sense circuit. ..... 72 Figure 3-13. High-level architecture of the stimulator circuit. ................................... 73 xii
Figure 3-14. Stimulator High-side circuits. ................................................................ 74 Figure 3-15. Schematic diagram of floating CMOS reference circuit. ....................... 76 Figure 3-16. High-voltage level shift circuit............................................................... 77 Figure 3-17. Stimulator low-side DAC. ...................................................................... 78 Figure 3-18. Schematic diagram of current DAC. ...................................................... 79 Figure 3-19. Schematic diagram of folded-cascode amplifier used in stimulator
circuits. .............................................................................................................................. 80 Figure 3-20. Schematic diagram of bias voltage generation circuit. .......................... 81 Figure 3-21. MSS SPI Serial Port Configuration. ...................................................... 82 Figure 3-22: Die microphotograph of MSS ASIC. ..................................................... 83 Figure 3-23. The MSS ASIC wirebonded in QFN40 surface mount package............ 84 Figure 3-24. ASIC test PCB with QFN-packaged DUT . ........................................... 85 Figure 3-25. Example of circuit edit using focused ion beam (FIB). ......................... 86 Figure 3-26: Test setup for differential amplifier and comparator circuits. ............... 87 Figure 3-27: MSS ASIC Boost Converter DMOS switching behavior with 50-ohm
load.................................................................................................................................... 88 Figure 3-28: Boost converter charging performance using on-chip DMOS switch. .. 89 Figure 3-29: Test setup used to measure negative voltage regulator circuit............... 90 Figure 3-30: Negative voltage regulator output for HVDD between 5 V and 25 V. .. 91 Figure 3-31: Stimulator High-side PMOS test circuit ................................................ 91 Figure 3-32: Switching behavior of stimulator high-side PMOSFET with HVDD = 20
V, f = 1 kHz, and 500 Ω resistor to GND. ........................................................................ 92 Figure 3-33: ASIC low-side stimulator test circuit. .................................................... 93 xiii
Figure 3-34: Stimulator low-side IDAC current transfer characteristic with
HVDD= 5 V. ..................................................................................................................... 93 Figure 3-35: Measured IDAC_SENSE voltage driving gate of NMOS cascode across
IDAC code range with HVDD =5 V................................................................................. 94 Figure 3-36: Stimulator compliance voltage sensitivity for several HVDD values and
RL = 100 Ω. ....................................................................................................................... 95 Figure 4-1: Approximate wound locations on porcine model for infected wound study
(2 wounds per side) [45]. .................................................................................................. 98 Figure 4-2: Stimulation bandage concept for large-wound studies. ......................... 100 Figure 4-3. Top (left) and bottom (right) views of bandage substrate ...................... 101 Figure 4-4: Test fixture and setup for substrate verification..................................... 101 Figure 4-5: Measured bandage substrate electrode impedance. ............................... 102 Figure 4-6: Block diagram of MSS3 Stimulator PCB .............................................. 104 Figure 4-7: Block diagram of boost converter controller. ........................................ 107 Figure 4-8: Generation of stimulation pulse timing signals using hardware timers . 108 Figure 4-9: Bluetooth Protocol Stack (from [57]) .................................................... 109 Figure 4-10: Communication scheme for transferring parameters and measurements
between base station and bandages. ................................................................................ 110 Figure 4-11: MSS3 microcontroller programmer connection using PCB probe. ..... 111 Figure 4-12: Snap connectors mounted on bottom side of MSS3 PCB module....... 112 Figure 4-13: MSS3 Stimulator Module PCB Assembled with Battery. ................... 113 Figure 4-14: Bandage substrates packaged for sterilization. .................................... 114 xiv
Figure 4-15: Communication network between stimulator modules, base station, and
internet data server. ......................................................................................................... 116 Figure 4-16: Wireless base station hardware PCB stack. ......................................... 117 Figure 4-17: Procedure for programming bandage stimulation parameters. ............ 118 Figure 4-18. MSS3 printed circuit board .................................................................. 120 Figure 4-19: Battery-powered MSS3 boost converter output charging waveform
(VBATT = 4.15 V) .............................................................................................................. 121 Figure 4-20: Demonstration of the MSS3 stimulator delivering a biphasic pulse. ... 122 Figure 4-21: Self-reported received signal strength (RSSI) from CC2541 radio as a
function of distance from USB dongle. .......................................................................... 123 Figure 4-22: Photograph of bandage with adhesive hydrocolloid material. ............. 125 Figure 4-23: Measured MSS3 substrate electrode impedance measured in an acute
test on living pig skin. ..................................................................................................... 126 Figure 4-24: Acute test schematic for measuring current flowing through tissue. ... 127 Figure 4-25: Electrode voltage and current waveforms measured in acute stimulation
test. .................................................................................................................................. 128 xv
Acknowledgements
I would like to thank my research advisor Dr. Garverick for his guidance and
instruction during both my undergraduate and graduate education at CWRU. I have
enjoyed the opportunity to design circuits at every level from the transistor to the
embedded system for both industrial and medical applications, and I likely would not
have had such a diverse experience elsewhere. I appreciate the respect he shows each
student and the encouragement he provides through the toughest challenges.
I could not have completed this program without Dr. Bogie, my biomedical research
advisor. She not only obtained the funding to carry out this research, but she granted me
the opportunity to extend my biomedical training far beyond that of a traditional
electrical engineer. I gained invaluable hands-on research experience by participating in
the animal experiments that will help me design better medical devices in the future. She
has been advisor both personally and professionally, and I truly appreciate all she has
done for me.
I also owe a debt of gratitude to committee members Dr. Zorman and Dr. Mohseni for
their valuable guidance in the design and testing of the devices presented in this work.
Dr. Zorman also provided me access to his lab and equipment to manufacture numerous
bandages for the animal studies. Dr. Mohseni has funded and maintained the Cadence
EDA server used to design integrated circuits for all the design groups in the department,
and without his support my work could not have continued. I also thank Dr. Saab and
xvi
James Cox for providing me technical support and access to the Mentor Graphics EDA
tools used to complete the design verification steps of the ASIC design.
Jeremy Dunning has been instrumental to the success of the research by developing
fabrication methods for the substrates, and he has spent countless hours manufacturing
them with me. I also thank Jen Graebert for the practical training in animal care and her
friendship during many long surgery days and experiments. I have also learned a lot
from other members of the PES study team, including Dr. Kristi Henzel, Bruce Kinley,
Danli Lin, and the staffs of the VA ARF and CWRU ARC facilities.
I thank the staff of the APT Center for maintaining an organization that supports
collaborations between clinicians and researchers from a wide range of technical
backgrounds and diverse organizations. In particular, I thank Ron Triolo and Suzana
Iveljic for their efforts establishing the APT Center and providing the initial seed funding
for our research. I also thank Brad Boggs for his training in industry best practices and
for supporting this project.
I thank Reza Moshtaghin, Nan Avishai, and Amir Avishai of the Swaglok Center for
the Surface Analysis of Materials for training me to use the Scanning Electron
Microscope and helping me obtain the results needed to complete my work.
I have enjoyed working part-time on SBIR-funded projects with Dr. Walt Merrill for
the past five years, and I learned enormously from this experience. I am also grateful for
the time I had working with the late David Hiscock during his time with Haric and
Scientific Monitoring.
I would like to thank my colleagues and friends in both the MSIC group and the
EECS and BME departments. I thank my labmate Steve Majerus for the all the technical,
xvii
professional, and personal discussions we have shared throughout our time together in the
APT Center and Scientific Monitoring, Inc. Thank you to Bobby Lu, Dan Goff, Chris
Roberts, Chia-Wei Soong, Kanokwan Limnuson, Srihari Rajgopal, Xinyu Yu, Paras
Samsukha, Amita Patel, Masoud Roham, David Tian, Erik Peterson, Brian Murphy, and
Christa Moss for your friendship and personal, physical, and professional encouragement
during my time in graduate school.
Mom and Dad, thank you for all the sacrifices you made throughout my childhood to
give me a strong education and for teaching me to be a honest and caring person. Betsy,
thank you for being a supportive sister and for always setting a high standard for me to
follow. Dad Walter, thank you for raising a wonderful daughter, accepting me wholeheartedly as your son-in-law, and supporting us through our years as “professional
students.”
No words can express my gratitude to my wife Kristen Walter for her love, patience,
support, encouragement, friendship, and guidance along the rocky road of graduate
school. I am so fortunate to have met my perfect complement in a partner early in life,
and you have helped me grow as a person in so many ways. We have made it through an
extended long-distance relationship, and I am excited to begin the next chapter of our
lives in a place we can pursue our dreams together.
This work has received funding and support from the Cleveland VA APT Center and
research grants from the Department of Veterans Affairs RR&D (# F7129R) and STERIS
Corporation.
xviii
List of Abbreviations and Common Signal Names
ADC: Analog to Digital Converter
BLE: Bluetooth Low-Energy wireless protocol
CC2540: “System on a Chip” microcontroller with built-in Bluetooth radio
CHSEL: Channel Select signal (MSS ASIC).
CMOS: Complementary Metal Oxide Semiconductor
DAC: Digital to Analog Converter
DC: Direct Current
DUT: Device Under Test
DVDD: Digital Voltage supply (MSS ASIC)
ES: Electrical Stimulation
FIB: Focused Ion Beam
GND: Ground level for power supplies.
HV: High voltage
HVDD: High voltage supply for stimulator circuits
HVL: Logic low level for circuits operating from HVDD supply (MSS ASIC).
IDAC_SENSE: stimulator compliance voltage feedback signal (MSS ASIC)
ISSD: Surface Stimulator Device (first-generation stimulator)
LCP: Liquid Crystal Polymer
LSB: Least Significant Bit
MCU: Micro-Controller Unit
MOSFET: Metal Oxide Semiconductor Field Effect Transistor
xix
MSS1: Modular Surface Stimulator (second-generation stimulator)
MSS3: Modular Surface Stimulator (third-generation stimulator)
NBIAS: NMOS current mirror bias signal (MSS ASIC)
NCAS: NMOS cascode transistor bias signal (MSS ASIC)
NMOS: N-type Metal Oxide Semiconductor transistor
PFM: Pulse Frequency Mode
PIC: Peripheral Interface Controller (Microcontroller)
PMOS: P-type Metal Oxide Semiconductor transistor
PTAT: Proportional To Absolute Temperature
PWM: Pulse Width Modulation
RTOS: Real Time Operating System
SAR ADC: Successive Approximation Register analog to digital converter
SPI: Serial Peripheral Interface
TENS: Transcutaneous Electrical Nerve Stimulation
VDD: Power supply voltage for digital circuits
xx
Abstract
A Wireless Electrical Stimulation System
For Wound Healing Therapy
With Biphasic High-Voltage Pulsed Current Output
Abstract
by
DANIEL STEVEN HOWE
In this research, two wearable surface stimulation systems have been developed for
use in wound electrotherapy. These self-contained, battery-powered bandages have been
demonstrated in-vivo using both a rat chronic wound model and a pig infected wound
model and provide a new means to investigate the physiological mechanisms of wound
healing.
The first stimulation bandage was designed for use with a rat wound model and
consists of a stimulator PCB module and plastic electrode bandage. The PCB is
constructed from discrete COTS components and is powered by a small button cell
battery providing at least seven days of continuous use. This voltage-mode device
generates stimulation pulses that are 10 - 90 V in amplitude, 10 - 200 µs in width, and 12
– 25 Hz in frequency. Stimulation was typically applied to wounds for 10 minutes every
hour for one week, and then the device and wound dressings were replaced.
xxi
An ASIC has been developed using the OnSemi 0.7-µm I2T100 process and is
capable of operation up to 100 V. A high-gain, current-mode boost converter addresses
challenges associated with efficient generation of the large amplitude compliance voltage
(up to 90 V) from a small battery with limited output current capability. A biphasic
current mode stimulator was demonstrated with ±21-mA output range, 0.33-mA
resolution, and a voltage headroom requirement of 3.5 V at full scale output.
The second stimulation bandage uses this ASIC and was designed for use on larger
wounds, up to 6 cm in diameter. A rechargeable lithium polymer battery allows a single
stimulator module to be used for an entire 28-day study. The disposable electrode
bandage portion of the device is easily replaced in-situ by the clinician. Continuous
monitoring of the delivered stimulation current while the device is in place on an animal
is achieved using a microcontroller with a built-in Bluetooth Low Energy radio.
Performance information from up to six devices is recorded to a wireless base station
located outside the animal pen and may also be remotely accessed by research personnel.
xxii
1
Introduction
1.1 Clinical Motivation
Chronic wounds such as shown in Figure 1-1 are a major clinical challenge for a
large clinical population (estimated to be 1 to 3 million people in the US alone [1]) with
conditions such as diabetes, paraplegia, and other physical impairments. A wound is
considered chronic if it does not heal in a reasonable period of time. A practical
definition that is often cited is longer than 1-3 months [2, 3]. Such wounds are both
difficult and costly to treat. The treatment costs average around $22,000 per patient [4]
but varies widely with severity of the wound. The total cost of treatment within Medicare
alone is over $1 billion per year [5]. In addition, these wounds cause significant physical
and emotional stress for the patient [6].
Figure 1-1. Example of a chronic wound on a foot [7].
23
In addition to the treatment of chronic wounds, there is an emerging need for new
treatments for wounds infected with multidrug-resistant bacteria [8]. Electrotherapy may
be beneficial in the treatment of infected wounds both to reduce the use of antibiotics
(which leads to drug resistance) and to treat these drug-resistant infections. This concept
will be tested in a study using a porcine infected wound model and a wearable
stimulation device that is tailored for the animal.
1.2 Electrotherapy and Wound Healing
Electrical stimulation has been proposed as an adjunctive therapy for chronic
wounds, but little scientific evidence is available to support its practice. There are
several theories supporting the use of electrical stimulation as a chronic wound treatment.
The endogenous electrical currents found in normally-healing wounds are disrupted in
chronic wounds [9], and it is believed that restoring these currents may encourage healing
[10]. This “current of injury” is believed to facilitate the movement of various cells
within wounds via galvanotaxis during each phase of wound healing. Another possibility
is that this current upregulates fibroblasts, neutrophils, and other cells involved in the
healing process. This injury current has been measured on the order of 35 µA/cm2 and
flows out of the wound and within 2-3 mm of the wound margin. In chronic wounds, the
injury current is disrupted possibly due to tissue dehydration and chronic inflammation.
In addition to maintaining a moist wound environment, it is believed that restoring this
injury current using low-level DC current from an externally applied source may promote
wound healing.
Another theory is that electrical stimulation has an antiseptic or antibacterial effect
that kills micro-organisms infecting a wound [11]. It is suggested that this wound
24
disinfection reduces inflammation and subsequent healing phases may commence. In this
approach, high-voltage pulsed current (HVPC) is applied to the tissue to induce changes
in pH or temperature that kills micro-organisms. However, if the stimulation amplitude
is too high, gas formation may result due to water electrolysis that damages the tissue.
1.3 Previous Wound Healing Electrotherapy Studies
Kloth presents an extensive review of electrical stimulation wound healing studies
[12] that cover in-vitro, animal, and human subjects. Two studies using HVPC
stimulation are presented here to describe the range of waveform parameters.
Jercinovic et al [13] performed a human study using two round electrodes either 50 or
75 mm in diameter positioned on healthy skin 3 cm from the wound. Biphasic chargebalanced pulses 250 µs in width were delivered at 40 Hz for 4 s followed by a 4 s pause
for 2 hrs/day, 5 days/week, for 4 weeks. The current amplitude was increased to just
below the muscle contraction threshold of 35 mA. In their study with spinal-cord injured
(SCI) patients, they found healing improved with the application of ES. As a follow up
to the study, the initial control group was offered the stimulation treatment and this new
“crossover” group showed significant improvement as well. This is particularly
compelling because the same wounds failed to heal using standard care but healed after
ES treatment.
Weber et al [14] used their ImpediStim device to deliver HVPC stimulation to healthy
human subjects using a multi-electrode array. A twin-peaked waveform consisting of two
100-µs pulses at 100 Hz and up to 200 V in amplitude were used. The patients adjusted
the amplitude to just below the threshold of pain. The acute experiment applied
25
stimulation for 30 minutes, then measured the skin temperature, TcPO2, and tissue
impedance at 1-h and 6-h timepoints. The brief experiment did not show any change in
the measured parameters that would indicate a physiological effect. However, erythema
(redness of the skin) was observed under the electrodes. The authors suggest increasing
the treatment duration or the stimulation amplitude may result in detectable differences
between stimulated and non-stimulated skin.
1.3.1
Current research limitations
In the reported research studies, ES therapy is limited to relatively brief sessions
lasting no more than two hours per day and five days per week. This is generally due to
researcher/caregiver time limitations and participant acceptance that allows treatment
only during dressing changes as part of standard care. A wearable device will allow
continuous ES therapy between dressing changes that may accelerate wound healing by
maintaining a consistent dosage.
Animal studies are particularly valuable for understanding the effects of electrical
stimulation on the physiological mechanisms of wound healing. In particular, chronic
wounds with consistent etiology, size, and shape can be studied in a study population
with similar age, health, medication, and nutrition. Animal experiments also facilitate
the study of control wounds in the same animal that rarely occur in a patient population.
In a patient population, it is ethically required that ES be used only as an adjunctive
therapy to the standard care although it may confound study results. For infection
studies, it is preferable to inoculate wounds with a selected monoculture that is obviously
restricted to animal studies.
26
1.4 Existing electrical stimulation therapies for wound care
Currently there are two major types of electrical stimulation therapies in clinical use.
Microcurrent devices attempt to restore the natural “current of injury” to the wound using
low-voltage stimulation producing a DC current on the order of 10-100 uA. Other
stimulators such as re-purposed TENS machines deliver stimulation current pulses up to
20 mA.
1.4.1
Microcurrent devices
The Vomaris Procellera® (Figure 1-2) is a commercial wound dressing that
generates micro-currents on the order of 10 µA and a voltage potential from 300 to 900
mV using printed “micro batteries” consisting of elemental silver and zinc. This product
is indicated for use in the treatment of stage III and IV wounds but produces a much
lower level of stimulation consistent with the galvanotaxic approach of wound healing.
The silver ions may also contribute a bacteriostatic effect on the wound environment.
Each dressing may remain in place for up to 7 days, and the product is indicated for no
more than 28 days of continuous use.
Figure 1-2: The Vomaris Procellera(R) microcurrent wound dressing.
27
1.4.2
TENS/ pain management devices
The LGMedSupply TEC Elite portable TENS unit (Figure 1-3) is indicated for
pain management and muscle therapy but the stimulator has similar output characteristics
as the proposed device (Table 1-1). Many of the HVPC studies utilize bedside or portable
TENS units such as this device.
Figure 1-3. Example of a TENS portable stimulator (from [15]).
1.4.3
Research Device: DS4
The Digitimer DS4 shown in Figure 1-1 is a research tool used in short-term
neurodynamics studies and has similar stimulation specifications as the proposed device,
especially the compliance voltage. Note that it is not a stand-alone stimulator as it
requires an external waveform generator to create stimulation pulses.
28
Figure 1-4: DigiTimer Biphasic stimulator for benchtop testing
Table 1-1 shows a comparison between the specifications of the TENS and DigiTimer
stimulators with the proposed wearable device.
Table 1-1: Specifications of a portable electrical stimulation unit.
Specification
TEC-Elite
Digitimer DS4
Proposed
device
Indication
Approved for Human Use
Non-human use only
Animal use
Power Supply
9V rechargeable battery
(24 hr lifetime)
AC Adapter
Handheld
Battery powered:
10 x GP123A (12V each)
40 mAh per cell
190 x 110 x 80 mm
500 g
Battery powered
0-105 mA
square biphasic pulse
4 TENS modes: Burst, constant,
modulation, modulation 1A
3 Muscle Stimulation modes:
Synchronous, Asynchronous,
Delay
1-60 minutes
±10 µA to ±10 mA
analog biphasic pulse
Analog input
(up to ±10V FS)
±20 mA
N/A
1-60 minutes/hr
for 1 week
up to 5 kHz
12.5-25 Hz
5-40 µs rise time
10-200 µs
±48V output
(battery-powered)
±15 V output
(line-powered)
0-90 V
Stimulator size
Current
Amplitude
Treatment
Period
Pulse
Frequency
Pulse Width
TENS mode: 0.5-150 Hz
MS mode: 2-5 Hz, 90-130Hz
50-300 µs
Compliance
Voltage
0-50 V (1000 Ω load)
Must fit on
species used in
research study
Biphasic
29
1.5 Previous work
1.5.1
Surface Stimulation Device (SSD)
The Surface Stimulation Device shown in Figure 1-5 was the first system
developed in the current program to study wound healing. The device specifications,
design, and performance are reported in [16]. The device consists of a PCB measuring
44.5 x 31.75 mm and a polyimide substrate with two platinum electrodes. The PCB
includes an 8-bit microcontroller, an IrDA serial port, and a discrete boost converter
stimulator powered by a 3-V Lithium coin battery. The device was hand-soldered using
lead-free solder, then encapsulated in heat shrink and silicone.
Figure 1-5: Surface Stimulation Device (SSD), from [16].
The device was tested on a rabbit ear wound model (Figure 1-6), and several key
design issues were revealed. The rabbit ear was too flexible to support the weight of the
device, and it was not well-tolerated by the animal. The electrode gels adhered poorly to
the skin, and it was necessary to use adhesive Tegaderm® film to wrap the device in
place. The infrared IrDA® serial port was difficult to use in a clinical setting, and there
was otherwise no visual indication of the device operation. The folded-tab contacts used
on the single-layer substrate were also prone to cracking.
30
Figure 1-6: SSD attached to rabbit ear
1.6 Research Objective
The objective of this work is to develop a wearable stimulation system suitable for
research studies that may be readily adapted for use in a range of applications and
species. This device will enable research studies not previously feasible due to the need
for tethers or device physical size and weight limitations. The following intermediate
goals will lead to the realization of the main research objective:
1) Design a discrete-component device suitable for use with a pre-clinical small
animal research studies (Chapter 2).
2) Develop an ASIC boost converter controller and current-mode stimulator circuit
for stimulation up to 90 V or 21 mA (Chapter 3).
3) Develop a wearable, wireless system with an integrated circuit stimulator and
provisions for simultaneous use of multiple devices, then validate device
operation in an ex-vivo animal study (Chapter 4).
31
This research has been conducted as part of a research project sponsored by the US
Department of Veterans Affairs focusing on the Physiological Mechanisms of Electrical
Stimulation in ischemic wounds.
32
2
Modular Surface Stimulator for Rodent Studies (MSS)
A stimulation bandage design is needed for use on a rat wound model. As compared
to the SSD [16, 17], the new device needs to be smaller and lighter weight and must also
accommodate a tissue interface that has a smaller radius of curvature that is subject to
twisting and bending. The entirely re-designed device, named the Modular Surface
Stimulator (MSS), has a smaller stimulator PCB module, lower power consumption, and
longer run time. A high-level block diagram of the proposed device is shown in Figure
2-1. A reusable, battery-powered stimulator mounts to a disposable, flexible plastic
“bandage.” The bandage adheres to the skin surrounding the wound, and two hydrogel
electrodes conduct the stimulation current into the tissue.
Figure 2-1: Block diagram of the proposed MSS consisting of a flexible printed
bandage and a reusable stimulator module.
2.1 Research Study Overview
The research study which uses this device seeks to identify the physiological
mechanisms of electrotherapy in wound healing and identify the most effective
stimulation waveform parameters. A previously-validated ischemic wound model in rats
33
[18] was adapted for electrotherapy studies using the proposed device. As shown in
Figure 2-2, an ischemic tissue flap is created along the back of the rat according to the
wound model procedure. Two ischemic wounds are created using a biopsy punch in this
slowly-healing region. Two more wounds are created in the healthy tissue adjacent to the
ischemic wounds. These control wounds do not receive stimulation but are used to
normalize the rate of healing between subjects.
Figure 2-2: Rat wound model used with MSS
The MSS device is placed over the wounds on the back of the rat, then covered with
additional protective dressings and a fabric jacket. Electrical stimulation is delivered to
the wounds for 7 days, and then the device is replaced during weekly dressing changes.
The stimulator PCB is removed from the disposable substrate, cleaned, and then
refurbished for later use.
The stimulator performance requirements listed in Table 2-1 include the expected
range of effective stimulation waveform parameters and the required battery lifetime for
use in in-vivo studies. The exceptionally low duty factor of the shortest pulse has a large
34
impact on the design of the high-voltage circuits, and severely limits their power
efficiency.
Table 2-1: Design requirements for small-animal device
Design Requirement
Specification
Battery lifetime
Stimulator weight
Stimulator size
Overall cost
Compliance voltage
Current Amplitude
Pulse Width, Frequency
Stimulation Period
Up to 168 hrs
Less than 50 g
No more than 3 x 5 x 0.7 cm
Less than $100
5-90 V
0-20 mA, biphasic
10-200 µs, 12-25 Hz
1-60 minutes/ hr
The programmable timing parameters shown in Table 2-1 are represented graphically
in Figure 2-3. The size and shape of ischemic wounds varies widely, so the device must
provide stimulation with a wide range of current pulse amplitude and width. Tissue
impedance also varies greatly, and typical skin preparation for electrode application
cannot be performed on the compromised tissues surrounding a wound, so a compliance
voltage of 90 V is required.
Figure 2-3. Graphical representation of the programmable stimulation waveform
parameters that cover a range of potentially therapeutic levels.
35
A key technical challenge of such a device is the generation of stimulation pulses
with the required high voltage compliance in a controlled and power-efficient manner,
and with a compact form factor. It is proposed that stimulation pulse shape is less critical
to wound healing than the listed parameters, so this stimulator design maximizes output
range and power efficiency at the expense of control of the pulse shape.
2.1.1
Battery review
The Lithium battery chemistry used in the ISSD does not provide the required
energy capacity within the size constraints of the MSS device, so other primary (nonrechargeable) battery chemistries were surveyed and the available types are presented in
Table 2-2. Rechargeable cells were not included in the survey because their energy
density is generally lower than non-rechargeable cells, as shown in Figure 2-4. Zinc
AirLithium batteries use the only chemistry that produces an output voltage sufficient to
directly power the microcontroller (>2.5V); other chemistries would require an
intermediate voltage step-up converter. Among the button-cell chemistries, the Zinc-Air
type provides the most appreciable increase in energy density compared to Lithium.
Thus, Zinc-Air batteries were chosen for this device.
Table 2-2: Review of battery chemistries in coin/button form factor
Chemistry
Type
Nominal
Voltage
Maximum
Cell Capacity
Available
Lithium
Alkaline
Silver
Zinc-Air
Coin
Button
Button
Button
3.0V
1.50V
1.55V
1.40V
130 mAh
150 mAh
195 mAh
635 mAh
Size of Max.
Capacity Cell
(Diameter x
thickness)
16 x 32 mm
11.6 x 5.4 mm
11.6 x 5.4 mm
11.6 x 5.4 mm
Approx.
Energy
Density
mWh/cm3
61
394
530
1558
Discharge
Curve
Linear
Linear
Flat
Flat
36
Figure 2-4: Comparison of battery energy density for rechargeable and nonrechargeable chemistries (from [19]).
Zinc-Air batteries are safe for use with humans and are commonly used in hearing
aid devices. The battery has perforations on the top of the case to allow air to enter the
cell, and this poses two potential drawbacks. First, the open cell may be exposed to
wound exudate or moisture. However, the battery is located on the top of the bandage
away from the wounds, so the likelihood of this event is low. Second, Zinc-Air cells will
self-discharge within one month [20] so they must be replaced frequently. This is also
not an issue because the MSS devices will be replaced weekly during dressing changes.
2.1.2
VDD Step-up converter
Though this appears inconvenient, a step-up converter would help avoid the
battery voltage brownout condition experienced with the ISSD. In addition, a step-up
converter allows any battery type to be discharged more deeply than would otherwise be
limited by the microcontroller operating voltage. The discharge depth is limited by the
the minimum input voltage of the low-voltage boost converter. An integrated boost
37
converter designed for battery operation as low as 0.8V [21] was selected for the MSS.
The system voltage was selected to be 2.0 V, the minimum allowed for the
microcontroller, to minimize energy consumption.
2.1.3
Microcontroller
New microcontroller product lines released since the ISSD have moved to
foundry processes with a smaller feature size, and both memory capacity and device
functionality have increased. These processors are designed to run programs compiled a
high-level language such as C, and the code development time is reduced. The 16-bit
CPU and memory bus decreases the number of instructions required per computation,
and the program execution time per pulse is reduced. Advancements in microcontroller
power management now allow the clock frequency for the CPU and the peripherals to be
individually controlled. The CPU is often the largest consumer of energy on the
microcontroller, so throttling its clock rate while the communication and timer
peripherals run at the oscillator frequency significantly reduces power consumption. For
example, the IDD current drops more than two orders of magnitude (from 3 mA to 15 µA)
when the clock is throttled from 16 MHz to 32 kHz [22]. The CPU may be switched
back to normal operation when faster control is required.
2.2 Boost Converter Circuit
A discrete boost converter with a novel controller is presented in this section. The
following discussion assumes an understanding of the principles of boost converter
operation [21] and will focus on the key differences with the presented circuit.
38
2.2.1
Battery Limitations
The MSS uses a replaceable Zinc-Air button cell battery because it satisfies the device
size and weight requirements and has higher energy density than rechargeable battery
chemistries. A microcontroller and other logic circuits are directly powered from this
battery. Lithium coin cell batteries have relatively limited output current capability, and
this is a key restriction in the design and operation of the boost converter circuit.
2.2.2
Operational Considerations
A voltage step-up circuit is required to generate the stimulator compliance voltage from
the 1.4-V battery. The boost converter topology was selected over flyback and switched
capacitor technologies because it doesn’t require a physically large transformer, it
achieves a high step-up ratio in a single stage, and the controller is simple and robust.
However, there is no monolithic boost converter IC presently available with
programmable high-voltage capability and the required power efficiency for this pulsed
stimulator.
In a conventional voltage-mode boost converter circuit (Figure 2-5), the transistor
switch is controlled by a pulse-width modulated (PWM) signal. The duty factor is
modulated to maintain a specified output voltage in response to changes in the load
current. When the load current is low, the controller goes into a pulse-skipping or PFM
mode of operation. When the load current is too high, the duty factor is internally limited
to protect the switch and source power supply [2].
39
Figure 2-5: Schematic diagram of the boost converter circuit using discrete
components for HV circuitry and microcontroller for the control loop.
In this pulsed-output application, however, the boost converter only operates at the
two load extremes. When the compliance voltage capacitor is not fully charged, the
boost converter operates in a PWM mode at a maximum output current limited by the
battery. The boost converter is constrained to operate in discontinuous mode to prevent
the inductor current from increasing each cycle. After the capacitor is charged, the boost
converter operates in pulse-skipping mode to maintain the desired level.
2.2.3
Critical Component Selection
The inductor selection is the most critical to the boost converter operation, and it is
constrained by the limited output current of the battery and the microcontroller clock
frequency. The frequency determines the minimum possible PWM pulse width
∙
, and the average inductor current when the MOSFET switch is on is given
by
40
∙
∙
,
2
∙
(1)
2
where T is the PWM period, D is the duty factor, and
∙
is the on-time.
Since the boost circuit operates in discontinuous mode, i.e. the inductor current ramps
from zero to a maximum during each cycle. The time-average current drawn from the
battery is
,
∙
∙ ∙
,
(2)
The inductor value must be chosen to limit the average inductor current to within the
capability of the battery.
Within these limitations, the smallest value inductor was
selected to reduce package size and to minimize series resistance.
The remaining boost converter components must also be selected with consideration
for high-voltage yet efficient operation. The n-channel MOSFET switch and Schottky
diode rectifier are rated for low reverse-bias leakage. The output storage capacitor is a
ceramic X7R dielectric with a 100-V rating that achieves high capacitance in a small
package.
2.2.4
Output capability
Based on energy conservation, the steady-state output voltage of the converter, prior to
activation of the stimulator was calculated as given in equation (2), where RFB is the
resistance of the voltage divider used in the feedback loop, and  is added to reflect the
voltage step-up efficiency.
41
∙
∙
∙
∙
2
(3)
The formula has been found to be accurate in predicting the output voltage. For
example, to achieve an output of 80 Volts from a 3-V battery, with L = 330 µH, RFB =
4 M, and measured  = 20%, this boost converter uses T = 6 µsec and D = 0.71. This
represents the maximum output voltage that can be achieved with the indicated
parameters. The low efficiency value is expected since the step-up ratio is large and
average output current is small [21, 23].
2.2.5
Controller firmware
The control loop for the compliance voltage is implemented in firmware. The output
voltage is sensed with a resistor divider, and is then digitized using the microcontroller
ADC. The sensed output voltage is compared to a threshold, and the controller operates
at a fixed duty cycle (determined by the battery voltage to limit peak inductor current)
when below the threshold and is completely turned off when above threshold. The
feedback loop does not require compensation for stability. The high voltage step-up ratio
limits the feed-forward gain, and the ADC conversion cycle intrinsically limits the
control loop bandwidth.
The boost converter generally operates in discontinuous conduction mode since the
step-up ratio from the battery voltage to the compliance voltage is large. However, when
first powered, the converter will operate in continuous conduction mode unless the peak
inductor current is temporarily limited. If the average inductor current rises due to
42
continuous conduction, the high internal resistance of the battery will result in a
significant voltage drop and potentially cause a system malfunction. Some monolithic
converters can sense the inductor current directly, but the high voltage operation of this
circuit makes such an approach impractical. Instead, a “soft start” feature is implemented
in the controller firmware by reducing the PWM duty factor until the output voltage
reaches a level where discontinuous operation can be assured.
The voltage feedback divider continuously loads the converter output, so it is critical
that the output be charged just prior to each stimulation pulse to minimize losses. For a
given PWM duty factor, the charging time would tend to increase as the battery voltage
drops. Some monolithic converters compensate for the drop in battery voltage using
voltage feed-forward, by adjusting the slope of an analog sawtooth waveform used to set
the PWM duty factor. However, this approach requires additional components and the
circuit may be difficult to stabilize when the battery voltage fluctuates due to its high
internal resistance.
In this converter, the microcontroller periodically senses the battery voltage using the
internal ADC and adjusts the PWM duty factor in firmware to achieve a consistent
maximum inductor current. Since time to reach maximum current is inversely
proportional to battery voltage, the energy drawn from the battery during one PWM pulse
is independent of battery voltage. Thus the average power drawn from the battery is
independent of battery voltage. In other words,
2
∙
∙
∙
and
∙
2
(4)
43
As shown in Figure 2-6, the measured charging time remains relatively constant for a
given output voltage regardless of the battery voltage, although non-idealities have
increasingly large effect at the higher set points. The consistent charging time saves
energy since the boost converter is not activated until the last possible instant before each
pulse.
Figure 2-6: Plot of the charging time vs. output voltage setpoints for several battery
voltages.
The fitted curves in Figure 2-6 use a power-law relationship with an exponent of
approximately 3/2. If components were ideal, the exponent would be 2 since energy
stored on the output capacitor is proportional to voltage squared, and the time-average
power supplied by the battery is approximately constant for all battery voltages and set
points, as described above. The faster than expected charging can be explained, in part,
by the behavior of the high-voltage capacitor, namely that incremental capacitance
decreases as voltage increases [24].
44
2.3 Stimulator Circuits
A high-voltage stimulation circuit is needed to deliver pulses of current from the boost
converter circuit to the tissue. The pulse timing is generated using hardware timers inside
the microcontroller, so the stimulation circuit must accept an input at a logic level of 3 V
or less.
The stimulator circuit is shown in Figure 2-7. High-voltage MOSFETs M1 and M2
form a push-pull driver that applies the stimulation voltage stored on CBOOST to the
electrodes through DC-blocking capacitor CBLOCK, which insures a biphasic output. M3
is used to DC-couple the stimulator to the electrodes during impedance measurements.
Figure 2-7. Schematic diagram showing the high-voltage stimulator circuit.
The high-side driver for PMOS switches is shown in Fig. 2-8. When the low-side
NMOS is switched on, the Zener diode operates in reverse breakdown and the Zener
voltage is applied to the VSG of the high-side PMOS pass switch. Since, the PMOS VSG
is independent of VBOOST, the on-resistance will remain relatively constant as VBOOST
decays during a stimulation pulse. When the LV control is turned off, the voltage across
45
the Zener diode collapses and the PMOS transistor is turned off. The turn-off time will
be relatively slow, but this is not important since precise pulse shapes are not required,
and because the duty factor of stimulation is very low.
Fig. 2-8. Schematic diagram of the high-side driver that controls PMOS switches
with a constant VSG across a 90 V supply range.
The output current can be controlled indirectly by estimating the tissue resistance and
charging the capacitor to the required voltage. Resistance is measured by charging the
compliance voltage capacitor to a set voltage, then examining the discharge waveform for
a fixed period of time. Resistance is then calculated in the microcontroller based on the
slope of the discharge waveform, under the assumption of 1st-order RC discharge. Based
on the voltage resolution (10 mV) and sample rate of the ADC (100 ksps), the
computation has a resolution of 500 Ω.
2.4 Experimental Results
The boost converter and stimulator circuits were fabricated as part of the MSS1
surface stimulation device prototype shown in Figure 2-9. The stimulator module
measures 3.0 x 5.0 x 0.7 cm and weighs 7.2 grams.
46
2.4.1
Modular Surface Stimulator Version 1(MSS)
As compared to the SSD stimulator reported in [16], the MSS1 was designed to fit on
the back of a rat and for improved battery lifetime. The infrared serial port was replaced
with a pushbutton and LED to simplify use during surgery. An EEPROM memory was
added to increase the amount of data logger storage capacity. The device is programmed
through a basic UART and pin header connection.
Figure 2-9: Modular Surface Stimulator (MSS1) Version 1 Device
The MSS1 PCB was soldered using solder paste stencils and a reflow oven, then
cleaned using an ultrasonic bath. The encapsulation was changed from heat shrink and
silicone to a deposited parylene coating to improve function and reliability.
A new substrate was designed by another member of the project team. The
material was changed from polyimide to liquid-crystal polymer (LCP) and the folded-tab
PCB contacts were replaced with conductive adhesive tape. The platinum electrodes
were replaced with hydrogel material designed for electrical stimulation applications.
47
The initial animal experiments using the MSS1 yielded several valuable design
insights. The mechanical switch used to activate the stimulator would stop functioning
after repeated use, possibly due to corrosion or other fouling. While useful during
benchtop testing, the switch did not provide benefit to the clinicians since the stimulator
is activated immediately after inserting the battery into the device. In later builds, the
device begins stimulating automatically after powering up.
The lithium battery used did have sufficient capacity for week-long usage, and a
larger size cell could not be accommodated within the size constraints of the device. In
addition, it is believed that the microcontroller would reset itself when the partiallydrained battery browned out during boost converter operation.
The lithium cell was
replaced with a Zinc Air battery and low-voltage boost converter circuit occupying the
same PCB area.
The microcontroller firmware for basic operation of the stimulation module
consumed all available program and data memories, so it was not possible to expand
functionality or otherwise improve performance of the device through code. In addition,
the CPU architecture was not designed for high-level programming in the C language,
and modifications to the assembly code would often result in misdirected data pointers.
2.4.2
Stimulator Performance
A series of representative stimulation pulse waveforms are shown in Fig. 2-10 with
pulse width parameter set to 200 μs. The pulse current drops as the compliance voltage
capacitor decays, as expected for a relatively low value of resistance. A larger output
48
capacitor would improve output voltage compliance at the cost of charging time and
overall power efficiency.
The stimulator pulse rise time is approximately 10 μs and the fall time is negligible. The
rise time is limited by the turn-on current within the high-side driver, which has been kept
low to minimize current draw from the high-voltage boost capacitor.
The turn-off
transition is faster because the NMOS in the push-pull driver is driven directly by the
low-impedance driver of the microcontroller.
Fig. 2-10. Plot of output pulses for several load resistors.
The maximum pulse rate for a specified voltage is shown in Figure 2-11 at several
supply voltages and is limited by the boost capacitor charging time. The effect of the
battery internal resistance is more apparent as the output voltage increases. The pulse
49
frequency could be increased by decreasing the fixed 32-ms interval devoted to system
tasks during each period of stimulus.
Maximum Pulse Frequency (Hz)
27
25
23
21
3V source
19
2.8V source
2.6V source
17
3V battery
15
0
20
40
60
Stimulation Voltage Setpoint (V)
80
Figure 2-11: Measured maximum pulse frequency vs. voltage setpoint with various
supply voltages.
2.4.3
Boost converter efficiency
Boost converter efficiency is the key factor in determining the battery lifetime, and
the efficiency will vary with the stimulation settings. The overall efficiency is the
product of two components: first, the boost converter efficiency in charging the output
storage capacitor, and second, the efficiency in discharging the capacitor into the tissue.
The boost converter draws high-frequency spikes of current that are challenging to
accurately capture, so a low-pass filter circuit was used in the test circuit to average the
50
current drawn from the power source. Efficiency is defined as the energy stored on the
capacitor after charging divided by the measured current integrated over the charging
period.
As shown in Figure 2-12, the efficiency is much lower than values typically found in
datasheets for boost converters. This is not an unexpected result since the input current
limit due to the battery and the high step-up ratio lead to large losses in the switching
components. The efficiency is slightly lower when a battery is used instead of a supply
because of the internal impedance of the cell.
Figure 2-12. Measured MSS1 Boost Converter Charging Efficiency for Selected
Supply Voltages.
The second efficiency component is simply the ratio of energy discharged from the
capacitor per stimulation pulse divided by the total energy stored on the capacitor. Any
residual charge on the capacitor is lost between cycles through the voltage sense resistor
51
network. As shown in Figure 2-16, the efficiency decreases with increasing load
resistance because the capacitor is discharged less per cycle. The theoretical calculation
was made using the first-order equation for a capacitor discharging through a resistor:
∙
(5)
Figure 2-13: Measured MSS stimulator efficiency for selected boost capacitor
voltages.
The energy capacity of coin cell batteries is heavily influenced by the discharge rate
and other operating characteristics. The labeled capacity of Lithium batteries is typically
measured under low, continuous current until the battery voltage drops to 2 V, but the
actual capacity is heavily influenced by the detailed output current characteristics. The
capacity of battery chemistries such as Zinc-air is generally reported for a set of
conditions using a pulsed output current [4].
52
Device run time was measured for a selection of stimulation parameters and load
resistances. As shown in Fig. 2-14, the battery lifetime decreases with stimulation
voltage but exceeds the 168-hour requirement for all tissue resistance values. The
lifetime remains constant between 50 V and 80 V because the stimulator pulse rate is
reduced for the highest output voltages.
Device Lifetime (Hr)
300
250
200
150
5k
100
10k
20k
50
50k
0
0
10 20 30 40 50 60 70 80
Stimulation Voltage (V)
Fig. 2-14. Measured battery lifetime versus stimulation voltage for selected output
resistances. Pulse rate is 25 Hz, pulse width is 200 µs, and stimulation interval is 10
min/hr.
The device power efficiency is defined as the ratio of the energy delivered to the
stimulator output to that drawn from the battery, per cycle, including the energy
consumed by the microcontroller, boost converter, and stimulator circuits. The measured
device power efficiency ranges from 5% for the lowest output voltage to 20% for the
highest. This relatively low efficiency is not surprising given the large voltage step-up
ratio and intermittent operation. The main sources of energy loss in the boost converter
are capacitive switching losses in the NMOS switch and diode and DC losses in the
inductor as well as the feedback resistor divider.
53
2.5 Summary
A 90-V boost converter and stimulator circuits have been developed for a wearable
electrical stimulation system. The boost converter control algorithm has been designed to
support the generation of high-voltage, low duty-cycle current pulses in a power-efficient
manner. Voltage feed-forward and soft-start techniques are implemented in software to
support operation from coin cell batteries with high output impedance. The battery
lifetime of the fabricated device meets the 168-hour requirement for the experimental
application.
Future designs will incorporate the boost converter and stimulation circuitry in an
integrated circuit to improve power efficiency and reduce overall device size. A currentmode stimulator will be implemented to achieve rectangular pulses.
54
3
Application-Specific Integrated Circuit
An application-specific integrated circuit (ASIC) is necessary to achieve high system
performance and move towards a single-chip electrical stimulation device. The goal of
this first version of the stimulation ASIC is to improve the stimulator performance by
implementing a new high-voltage boost converter controller and a biphasic current DAC.
The logic for the stimulation waveform timing, memory, and the wireless communication
circuits will remain on a microcontroller to maintain design flexibility to meet the needs
of wound healing research studies.
3.1 Functional Requirements
The functional requirements are presented in Table 3-1 and are derived from the
clinical application specification.
Table 3-1. Functional requirements for ASIC.
Requirement
Compliance voltage range
Specification
10-90 V
Stimulator
Pulse current
Current resolution
Pulse width
Pulse frequency
(period)
±20 mA
0.5 mA
10-200 µs
12.5-25 Hz
(40-80 ms)
Load
Tissue resistance
5- 50 kΩ [25]
Boost converter
These requirements present design challenges for the high-voltage stimulator that
include efficiently generating a large compliance voltage from a high-impedance source,
interfacing control signals between low-voltage analog/logic circuits and the high-voltage
stimulator, and modulating the boost converter output based on the load impedance and
stimulation variables.
55
3.2 Foundry Process Features
This ASIC was fabricated using the ON Semi I2T100 mixed signal process. The
process was specifically designed for 100-V operation and includes a “floating pocket”
feature that electrically isolates a region of the substrate using a buried oxide layer
between the epi silicon layer and bulk. As shown in Figure 3-1, a lightly-doped NTUB
is also used to reduce the electric field between high and low voltage sections of the chip.
A library of high-voltage diffusion MOSFETs having both analog and digital
characteristics includes models that enable accurate design and simulation.
Figure 3-1: NDMOS floating transistor in I2T100 process (modified from [26]).
In the low-voltage sections of the chip, the transistors are designed for CMOS logic
operating at 5 V. However, a low-threshold PMOS device is available for use in analog
circuits. The circuits in this ASIC are designed to operate at 3.3 V using an off-chip
regulator.
3.3 High-Voltage Boost Converter
The variable compliance voltage (10-90 V) is generated from a small battery and
requires a large step-up ratio capability and efficient operation at low output current.
56
3.3.1
Step-up Topology
Several step-up circuit topologies were considered for this application as discussed
here.
The charge pump technique [27] can be generally described as storing, then
transferring, charge between a series of capacitors in such a manner that the output
voltage is increased at each stage. The Dickson topology [27] shown in Figure 3-2
requires a cascade of high-voltage capacitors and diode-connected MOSFETs that occupy
a large silicon area on an integrated circuit. The theoretical voltage of the Dickson
converter is
(3-1)
Typically the voltage is only increased 2 to 3 times per diode-capacitor stage, so
numerous stages would be required to achieve a total voltage gain of 60. The step-up
ratio is not easily modified. Depending on the specific implementation, high-voltage
interface circuits may be required at each stage.
Figure 3-2: Dickson Charge Pump Voltage Conversion Technique (from [27])
57
The flyback boost converter shown in Figure 3-3 has been used to achieve large stepup ratios but requires a transformer. The turns ratio of the transformer is fixed, but the
output voltage can be adjusted by the duty factor of switching. When the desired voltage
gain is lower than the turns ratio of the transformer, large voltage spikes may appear at
the output.
Figure 3-3. Representative schematic of the flyback converter architecture.
Ultimately, the boost converter topology shown in Figure 3-4 was selected since it
provides the desired variable step-up ratio, has a low component count, and a wide range
of control algorithms may be used to tailor the performance for the specific application
requirements. This is the same boost converter topology used in the previous discrete
versions of the device, but traditional PWM control is not used. Instead, new control
methods are used to directly control the peak inductor current each cycle.
58
Figure 3-4. Boost converter topology.
3.3.2
Control loop operation
Current mode control in boost converters refers to an inner control loop that sets the
peak inductor current on a cycle-by-cycle basis [28]. This method accounts for large
changes in the input voltage (such as in battery-operated systems) and achieves a similar
benefit as input voltage feed-forward. Directly sensing the inductor current is preferred
when the input voltage source has large internal resistance and the voltage may droop as
the output current increases. Limiting the inductor current reduces the possibility of
battery brown-out and increases the control system robustness.
Two control loops govern the operation of the boost converter. The first (inner)
current control loop limits the cycle-by-cycle peak inductor current. When the output is
near the desired high-voltage level, the forward (open-loop) voltage gain is small so
modulating the peak inductor current has little use. Therefore, the peak inductor current
is a constant, independent of the output level. In this application, the boost converter
switch should turn off when the inductor current reaches a maximum level and then
59
switch on again when the current approaches zero. The desired behavior is diagrammed
in Figure 3-5.
Figure 3-5. Idealized inner-loop current control for the boost converter.
The second (outer) control loop regulates the output voltage by enabling and
disabling the inner loop current-mode control. This loop operates at a rate determined by
the specific stimulator implementation but will always be slower than the inner inductor
current control loop. As shown in Figure 3-6, an attenuated and buffered version of the
output voltage is driven off-chip to an ADC input on a microcontroller (MCU). The
boost converter control loop state machine algorithm determines if the PWM operation
should be enabled to increase the output voltage. Since the boost converter may require
several milliseconds to initially charge the output capacitor, the microcontroller may
sleep in a low-power mode until the output approaches the desired level.
60
Figure 3-6. MSS ASIC-based boost converter circuit.
3.3.3
Inductor current sense
Several methods for sensing the inductor current were considered for this application.
A common method is to measure the current flowing out of the inductor by sensing the
drain-source voltage across the NMOS switch, as shown in Figure 3-7. The NMOS
switch operates in the triode region during the inductor charging phase, so VDS increases
approximately linearly with the ramping IDS (since (VGS -VT) >> VDS) and may be used
as the feedback signal. However, the voltage at the drain node rises to a high voltage
level during the second phase of the boost converter operation and would require
additional circuitry to interface with logic-level feedback circuits. Additionally, these
interface circuits would likely require a high-voltage supply and would consume energy
from the supply output.
61
Figure 3-7. Inductor current sense method across shorting NMOS.
A second method, shown in Figure 3-8, is to sense the current flowing into the
inductor from the battery side of the inductor. In this case, a second MOSFET switch is
inserted in series with the inductor and operated in the triode region. The VDS of the 2nd
MOSFET may be sensed to approximate the inductor current, and the MOSFET provides
the ability to disconnect the circuit from the power supply to further reduce energy
consumption when the stimulator is not operating. High voltages do not appear on the
battery-side of the inductor, so the feedback circuitry may be operated at logic levels.
However, the VGS of the MOSFET must be carefully controlled to maintain a consistent
IDS-VDS transfer characteristic. As previously described, the battery voltage is subject to
large fluctuations when the internal resistance is high and the gate voltage would need to
accurately track this level.
62
Figure 3-8. Inductor current sense method using battery-side PMOS.
The third method, shown in Figure 3-9, was selected for this circuit and is simply a
low-value resistor inserted in series with the inductor. The voltage across the resistor is
proportional to the current flowing through the inductor, and again the low voltage
feedback signal may be easily processed with circuitry operated at logic voltage levels.
This component is kept off-chip so the peak inductor current may be easily modified to
match the characteristics of the power supply available for the device in a particular
study.
Figure 3-9. Inductor current sense using off-chip resistor.
63
3.3.4
Current sense amplifier
The peak voltage across the current sense resistor is kept small to minimize energy
loss. The differential voltage VR across the resistor is amplified using the single-stage
differential amplifier shown in Figure 3-10.
Figure 3-10. Differential Amplifier used as inductor current sense amplifier.
Since M4,5 are diode connected, voltage gain is dominated by transconductance and is
approximately independent of bias current, as given by
,
,
≅
⁄
⁄
,
,
∙
(3-2)
Bias current was set to 30 µA to obtain adequately low noise floor and output
impedance. A simple hand calculation using square-law models suggests a voltage gain
of about 10, but the PMOS input pair is operating in weak inversion. SPICE simulation
predicts a voltage gain of ~5.
64
The common-mode input range is
0
,
to
(3-3)
2.45 ,
,
(3-4)
where
This easily accomodates the expected 0.8 V to 1.5 V range of many single-cell
batteries.
The common-mode output range of a single-ended output is
0.73
,
∙
to
,
(3-5)
1.72
(3-6)
The maximum differential output is the difference of the single-ended extremes
approximately ±1 V.
3.3.5
Current sense comparator
The differential output of the inductor current sense amplifier drives both inputs of a
comparator used to control the inductor switch MOSFET. The comparator uses a
regenerative feedback circuit [29] that allows the hysteresis thresholds to be determined
using device size ratios. The schematic diagram for the current sense comparator circuit
is shown in Figure 3-11. The transistor sizes are provided separately in Table 3-2.
65
Figure 3-11. Schematic diagram of current sense comparator circuit.
Table 3-2. Transistor sizes for current sense comparator circuit.
Component Type
W/L (µm / µm)
M1
PMOSL1 224 / 2.8
M2a, M2b
PMOSL 56 / 5.6
M3a, M3b
NMOS
5.6 / 1.4
M4a, M4b,
NMOS
4.2 / 1.4
M5a, M5b,
M6a, M6b
M7a,M7b
PMOSL 11.2 / 1.4
M8a, M8b
M9a, M9b
PMOSL 56 / 2.8
1. PMOSL is a low-VTP device
Normalized Ratio
80
10
4
3
8
20
The maximum and minimum inductor current levels are determined by the upper and
lower comparator thresholds. First, ignoring PMOS devices M9a and M9b, the circuit is
symmetric and the switching thresholds are equal and opposite [29] according to the
equations:
,
where
|
|
1
1
(3-7)
,
,
66
The component sizes and bias levels were selected such that
|
|
245mV, so
35mV.
Since the thresholds are equal and opposite in a symmetric circuit, device M9a is
added to inject an offset current IOFFSET into the left side of the circuit. M9b does not
conduct current but is included to improve ac symmetry.
To determine the shifted switching thresholds for an offset current IOFFSET supplied by
M9a, the following equations are solved. At the positive switching threshold, the current
sourced by
is equal to that sunk by
, and
is beginning to conduct:
,
∙
and
where
.
2
Thus,
1
1
|
1
|
When
|
1
1
2
(3-8)
2
1
2
|
1
1
1
4/3 and
|
given that
|
|
|
2
1/4,
| ≅ 105 mV,
| has a nominal value of 245 mV.
67
For the negative switching threshold, the current sourced by
that sunk by
, and
and
is equal to
is beginning to conduct:
2
2
2
2
2
1
1
|
1
|
|
When
In other words, the
|
1
1
|
1
4/3 and
1/4,
3
14
1
7
|
|
|
| ≅ 17mV,
|
245 mV .
term accounts for a ± threshold of ±
term accounts for a positive shift of
and
(3-9)
1
1
1
|
2
2
2
|
when
2
|
| . The
in the two cases.
The previous calculations do not account for channel length modulation in the
saturated MOSFETs. VSD9 will be greater than VSD1 , so the offset current will be slightly
higher than
. This increases both
and
but both thresholds are positive and
proper switching behavior will be maintained.
68
3.3.6
NMOS Switch
The boost converter switch is implemented using an integrated DMOS transistor rated
for operation with VDS <= 100 V. The transistor was implemented from a library P-Cell
(parameterized cell) and sized as wide as possible, given the die area constraints, to
minimize on-resistance. The simulated characteristics of the on-chip DMOS are similar to
the off-chip COTS component it replaces on previous discrete versions of the device [30].
Table 3-3. Comparison of on-chip and off-chip boost converter switch components.
Size
RDS,ON
Vt
3.3.7
Off-chip switch:
BSS123W-F
Unknown (SC-70 package)
10 Ω
1.4 V nom.
On-chip DMOS
W/L = 15000/6 µm
7.2 Ω (simulated)
1.02 V
Test condition
(datasheet)
VGS = 4.5 V, 0.17 A
VDS=VGS, ID=1 mA
Output Rectifier
A key component of the boost converter is the output rectifier that prevents current
from flowing backwards to the battery after the inductor current drops to zero.
Synchronous rectification using a PMOS switch is growing in popularity for boost
converter circuits [31, 32] because the lower voltage drop across the device increases
efficiency. The PMOS also provides a means of disconnecting the output from the power
supply when the backgate diode is disconnected. However, the PMOS must be switched
at the precise time to prevent current flowing backwards after the inductor discharges.
The PMOS gate must be driven to the high-voltage output level to turn it off and a
low level to turn it on. In this application, the output voltage often exceeds the VSG rating
of the PMOS, so the gate cannot be simply pulled to ground level. Thus, additional
circuits would be necessary to control the device. These additional components would
consume additional energy that would likely erase any performance benefit for the power
69
levels in this application. Therefore, an off-chip diode is used to implement the output
rectification function.
Traditionally a diode with fast recovery, low reverse-bias leakage, and low forwardbias voltage such as a Schottky is used to achieve this function. However, the Schottky
diodes that may be fabricated in a standard CMOS process often suffer from high
substrate leakage and high series resistance [33]. Higher-quality Schottky diodes can be
fabricated with CMOS processes using additional mask steps but are not widely available
[33]. Further, the aluminum interconnect and p-doped silicon used to form CMOS
Schottky diodes with low forward voltage have similar work functions that result in a
very small depletion region [34]. This application requires a rectifier that can withstand
100 V in reverse bias, further reducing the suitability of integrating an on-chip Schottky
diode. The BAT46W [35] Schottky diode was chosen.
Table 3-4. Key characteristics of the BAT46W schottky diode [35].
Mode
Forward bias
Reverse bias
Parameter
Forward Voltage
Repetitive Peak
Current
Blocking Voltage
Reverse Current
Reverse Capacitance
3.3.8
Value
0.25 V
0.45 V
350 mA
100V
0.02 – 0.2 µA
20 pF
Conditions
IF = 0.1 mA
IF = 10 mA
Duty Cycle
< 50%
25˚C,
VR=10-100 V
VR = 0V
f = 1 MHz
Output capacitor
The selection criteria for the boost converter output capacitor were carefully
investigated in the previous work [16]. In this application, a larger value capacitor may
be used since the loading from the resistive feedback network in previous designs has
70
been mitigated. Increasing the capacitance reduces the compliance voltage droop during
pulses and allows more energy to be delivered per pulse.
3.3.9
Output Voltage Sense
The output (boost) voltage is regulated using the ADC provided by the
microcontroller. It is necessary to attenuate the high-voltage output to a level within the
conversion range (0-2.048 V) of the microcontroller. A fixed resistor voltage divider
would continuously draw current from the high-voltage output between pulses and when
no measurement is taking place.
The boost voltage sense circuit shown in Figure 3-12
employs a high-side PMOS switch to disconnect the 1:50 resistor divider when no
measurements are being taken. A high-voltage level translator is required to control the
gate voltage and is presented in section 3.4.2.3.
The resistance values used in the divider network are kept large to minimize loading
of the high-voltage output, but this large output resistance is not suitable for driving the
capacitive load of an off-chip SAR ADC input. A differential amplifier configured in
unity-gain configuration and powered from the logic-level supply provides the output
buffering required to charge the off-chip ADC input within 10 µs. The linear output range
of the op-amp is 0.25 V to 2.5 V, so the boost voltage is accurately sensed between
12.5 V and 100 V. In practical application, the boost converter output will generally be
charged to at least 15 V so this will not limit system performance. The amplifier is
disabled when the feedback network is disconnected from the output to reduce energy
consumption. The amplifier is discussed in Section 3.4.3.2.
71
Figure 3-12. Schematic diagram of boost converter output voltage sense circuit.
3.4 Stimulator
The clinical application requires the stimulator to provide biphasic pulses up to 20
mA in amplitude with a resolution of 1 mA. The stimulator is designed for a compliance
voltage up to 90 V.
3.4.1
Architecture
The biphasic stimulator is implemented in an H-bridge architecture with the high-side
section containing HV PMOS switches and the low-side section containing a
programmable IDAC and current-steering circuitry (Figure 3-13). To facilitate testing,
the two sections are joined together off-chip and then connected to the stimulation
electrodes.
72
Figure 3-13. High-level architecture of the stimulator circuit.
While the specified 20-mA range and 1-mA resolution could be covered in 21
discrete steps (including 0 mA) using a 5-bit code (32 steps), a 6-bit code is used to
improve the resolution and to allow for calibration if the output current is found to be
inaccurate due to circuit non-linearity or offset. When a 6-bit code is used, 64 steps are
available. Each step will be set nominally to 0.33 mA so integer values of current can be
produced.
3.4.2
High-side switches
The stimulator high-side circuits are shown in Figure 3-14. A pair of high-side
PMOS switches are used in conjuction with the low-side DAC in an H-bridge
configuration to steer the stimulation current in either direction through the two
electrodes. These PMOS transistors are the 100-V DMOS type and require gate drive
signals that range from HVDD (Vsg = 0) to HVL (Vsg = HVDD-HVL). The transistor
gate oxide thickness limits VSG to 15 V, so the gate voltage cannot be simply pulled to
73
ground level to turn on the device. A negative voltage regulator produces the HVL level,
and voltage level converters translate the logic-level control signals to the PMOS gate
drive level.
Figure 3-14. Stimulator High-side circuits.
3.4.2.1 Negative voltage regulator
The negative voltage regulator operates in a shunt configuration and uses a CMOS
regulator circuit to set the HVL level. The low-voltage circuits are constructed in an
isolated P-well with a floating substrate that operates at the HVL level. The Zener-like
regulator circuit was designed to generate approximately 3.2 V using minimal bias
current (100 µA) that is sunk from the circuit using a 100-V n-channel DMOS cascode
switch connected to the logic-level circuitry. When the stimulator is not active, the
regulator is shut down and the PMOS devices will turn off when the HVL supply
collapses.
74
3.4.2.2 Floating CMOS Reference
The floating CMOS reference circuit [36] shown in Figure 3-15 mimics the I-V
characteristics of a discrete Zener diode. The circuit is fabricated in a HV pocket so lowvoltage devices may be used. However, care must be taken to prevent HVL from being
shorted to a low-voltage level during operation. In steady state, the current through M2
and M3 is made equal and ∆
,
,
across R3 determines the bias current
. This value is calculated as follows:
,
,
′
′
(3-10)
′
Since
.
Thus, and ∆
With R3 = 9.5 kΩ, ′
15μ /
is 18.5 µA and ∆
,
(3-11)
and
=40,
0.175 .
,
is given by Equation (3-12):
The reference voltage
2
,
|
|
∆
,
2
2
(3-12)
With |
|= 1.05 V, and R1 = 4R3, VZ ≈ 3.0 V.
SPICE simulations predict VZ = 3.15 V.
75
Figure 3-15. Schematic diagram of floating CMOS reference circuit.
3.4.2.3 Digital voltage level shift
Efficient level shifting is critical to the low-power operation of the high-voltage
circuitry. The capacitive level shift circuit [37] shown in Figure 3-16 is used to interface
the logic-level control signals from the microcontroller to the HV level for driving the
gates of HV-PMOS switches. Inverters I1-I3 are standard CMOS inverters constructed in
the substrate biased at the system ground level. The CMOS inverters I4-I6 operating at
the HV level are constructed using low-voltage devices in a floating pocket biased at the
HVL voltage level.
Capacitors C1 and C2 are charged to HVDD-DVDD and HVL-GND, so they must be
designed to tolerate this high voltage. The capacitors must be sized to couple adequate
charge to I4-I5. When a logic transition occurs at the In input, the bottom plates of C1
76
and C2 connected to I2 and I3 will simultaneously shift up/down by DVDD. If the
capacitors hold their charge, then the top plates will also shift by DVDD. The crosscoupled inverter pair (I4 and I5) will toggle and the logic state will be latched at the highvoltage level. Inverter I6 buffers the output of I4 to drive the large gate of the HV PMOS
output switch. The initial state of the output is indeterminate but cycling the input during
system initialization will reset the voltage across each capacitor and set the output to a
known state.
Figure 3-16. High-voltage level shift circuit.
3.4.3
Low-side DAC pair
The low side of the stimulator shown in Figure 3-17 consists of a SPI-controlled
current DAC, a pair of HV NMOS current steering switches, and support circuitry. The
IDAC uses a LSB reference current of 13 µA and converts a 6-bit digital code received
from the SPI port into a reference current between 0-875 µA. A current mirror using
NMOS devices operating in the triode region [38] scales the output current to the full 21–
77
mA range. Based on the state of CHSEL, one of two HV NMOS cascode transistors is
driven to steer the output current to either the A- or B- pins.
By operating the mirroring transistors in the triode region, power dissipation is
reduced in the output stage and the minimum accurate output voltage is reduced. Unlike
[38], the VDS of the triode mirroring devices are kept the same using a second op-amp
control loop.
Figure 3-17. Stimulator low-side DAC.
3.4.3.1 Current DAC (IDAC)
The low-side current DAC (IDAC) is shown in Figure 3-18. The NCAS and NBIAS
levels are locally regenerated from a 4.1-µA reference current from the global bias circuit
to minimize inaccuracy due to device threshold mismatch across the die. A series of
binary-weighted cascade current mirrors produce a scaled version (up to 110 µA) of the
78
desired output current. A cascaded PMOS current mirror reverses the current direction
and multiplies the value by 8.
Figure 3-18. Schematic diagram of current DAC.
3.4.3.2 Stimulator MOSFET Control Amplifiers
The folded-cascode operational amplifier shown in Figure 3-19 is used both in the
stimulator to control the VDS of the triode MOSFETs, and in the boost converter to buffer
the output feedback voltage. The key requirements for this amplifier are an output swing
of below VTn and as close to the upper rail as possible. The inputs must operate at a
common-mode input as low as 0.1V. Additionally, the output must be able to charge the
gate capacitance of the stimulator output stage NMOS and DMOS devices.
79
Figure 3-19. Schematic diagram of folded-cascode amplifier used in stimulator
circuits.
3.4.3.3 Stimulator Control
The stimulator control block uses the CHSEL signal input to direct the gate control
voltage to one of the two cascode DMOS and forces off the other device. The STIM_EN
signal also controls the DMOS gate operation and is used to control the stimulation pulse
timing.
3.5 Support circuits
3.5.1
Bias generation and distribution
The bias voltages and currents are generated using a traditional bandgap voltage
reference circuit [39] shown in Figure 3-20. The MOSFET sizes are presented separately
in Table 3-5 for clarity. VLOW is the reference voltage used with the op-amp control
loops to set the desired VDS level of triode MOSFETs in the low-side stimulator circuit.
The voltage is proportional to absolute temperature (PTAT) but the stimulator circuit
does not require a precise value.
80
Figure 3-20. Schematic diagram of bias voltage generation circuit.
Table 3-5. Device sizes for bias circuit.
Subcircuit
Startup
Component
M1
M2, M3, M4
Type
PMOSL1
NMOS
W/L (µm / µm)
2.8 / 4.8
2.8 / 2.8
Enable
M5
PMOSL
28 / 2.8
Bandgap
Reference
M6 a, b
M7 a, b
M10 a, b
M8
M9
PMOSL
224/2.8
NMOS
56 / 1.4
Vlow level
M11 a, b
PMOSL
112/2.8
Nbias, Ncas levels
M12 a, b
M13
M14
M15
PMOSL
NMOS
NMOS
NMOS
64 / 2.8
16 / 2.8
32 / 2.8
IREF_SRC
IREF_SRC4
IREF_SINK
3.5.2
M16 a, b
PMOSL
M17 a, b
PMOSL
M18 a, b
NMOS
1. PMOSL is a low-VTP device
112 / 2.8
448 / 2.8
32 / 2.8
SPI Serial Port
The SPI serial port accepts a 12-bit command in this version of the ASIC to facilitate
future control register expansion. Bit assignment is summarized in Figure 3-21. Bits
81
D[11:8] are used to address the register (location 0h). Bits [7:6] are general-purpose
outputs that are presently connected to control inputs STIM_EN and BIAS_EN to reduce
I/O connections to the microcontroller. Bits [5:0] set the IDAC output current (1 LSB =
0.33 mA).
D[11:8]
Register address = 0h
D[7]
GP2
D[6]
GP1
D[5:0]
IDAC Current
Figure 3-21. MSS SPI Serial Port Configuration.
3.5.3
Pad Frame, ESD Protection, and Guard rings
A foundry-designed pad cell library was not available from MOSIS, so a custom
library was developed based on existing designs [40]. Analog and digital input pins use a
800-Ω resistor to limit input current, and diode-connected PMOS and NMOS devices
[41] clamp ESD transients and signals outside the voltage supply range. The digital
output pins are also protected by both the diode-connected MOS protection devices to
limit conduction in the body diodes of the output stage MOSFETs.
The low-voltage circuit supplies use separate pins and traces for the analog and
digital supplies. Special care was taken when routing high-voltage signals because the
field threshold voltage is only 13 V for poly traces [26]. Therefore, the high-voltage
circuits are placed in a separate region and the HVDD and HVL power traces are routed
in top-level metal (Metal 3). All high-voltage cells are surrounded by p-type guard rings
connected to ground, and substrate contacts are placed at most every 50 µm to avoid field
inversion. Double guard rings are placed around the boost converter switch and the
low-side stimulator output MOSFETs to avoid unwanted substrate current injection.
82
3.6 ASIC Test Results
3.6.1
Introduction
The MSS ASIC was fabricated using the OnSemi I2T100 0.7-µm BCD process using
the MOSIS service. The IC includes the boost converter, bias generation, serial SPI port,
stimulator low-side IDAC, stimulator high-side switches, and HVL generation circuits.
A die microphotograph is shown in Figure 3-22. The fabricated die measures 2.96 mm
by 2.96 mm and uses 40 I/O pads, including 11 for test purposes.
Figure 3-22: Die microphotograph of MSS ASIC.
A list of the ASIC bond pads with I/O type and function is provided in Table 3-6.
The die was packaged in a 40-pin QFN Open Cavity Package by the MOSIS service.
The package is shown in Figure 3-23 and measures 6 mm x 6 mm x 0.8 mm. A leadless,
small form-factor package was selected to minimize the stimulator size and facilitate
83
packaging. The packaged parts have the same pad orientation and as the bare die, so the
same signal list in Table 3-6 is used.
Table 3-6. List of MSS ASIC bond pads and their function.
#
Name
Type
Function
#
Name
Type
Function
1
AVDD
Power
Analog supply
21
HVSS
Power
Stimulator ground
2
AVSS
Power
Analog ground
22
ElecB-
Analog Out
Low stim. output B
3
VLOW
Analog Out
VDS level
23
ElecA-
Analog Out
Low stim. output A
4
NBIAS
Analog Out
Amplifier bias
24
ElecB+
Analog Out
High stim. output B
5
NCAS
Analog Out
Amplifier bias
25
ElecA+
Analog Out
High stim. output A
6
VBG
Analog Out
Reference
26
HVL
Power
High stim. gate drive
7
PBIAS
Analog Out
Amplifier bias
27
HVDD
Power
Stimulator supply
8
BIAS_EN
Digital In
Bias enable
28
VOUT
Analog In
Feedback divider input
9
D0
Digital Out
GP output
29
LX
Power
Boost converter switch
10
NC
-
Unused
30
BVSS
Power
Boost converter ground
11
DVDD
Power
Digital supply
31
SHUNT
Analog In
Shunt amplifier neg. input
12
GND
Power
Digital ground
32
BATT
Analog In
Shunt amplifier pos. input
13
SCLK
Digital In
Serial clock
33
SENSE
Analog Out
Feedback divider output
14
SDI
Digital In
Serial data in
34
SHUNT_AMP
Analog Out
Shunt amplifier output
15
SS
Digital In
Serial select
35
BOOST_MODE
Digital In
Boost mode select
16
SDO
Digital Out
Serial data out
36
IMODE_PWM
Digital Out
Comparator output
17
STIM_EN
Digital In
Stim. Enable
37
EXT_PWM
Digital In
18
CHSEL
Digital In
Channel select
38
BOOST_EN
Digital In
Boost converter
PWM input
Boost converter enable
19
IDAC_SENSE
Analog
39
SENSE_EN
Digital In
20
IDAC_MODE
Digital In
Low-side stim
gate drive
Current/voltage
mode select
40
D1
Digital Out
Feedback divider
enable
GP output
Figure 3-23. The MSS ASIC wirebonded in QFN40 surface mount package.
84
3.6.2
MSS ASIC Test Setup
The ASIC was tested using the evaluation board shown in Figure 3-24. Devices
under test are installed using an adapter board with either a QFN solder pad footprint or a
QFN socket for production device qualification. Parts packaged in a DIP40 may also be
tested using this PCB.
Figure 3-24. ASIC test PCB with QFN-packaged DUT .
3.6.3
Integrated Circuit Edits Using Focused Ion Beam
Three issues were found during initial testing that could be corrected using a Focused
Ion Beam (FIB) instrument. Repairs were simple and an updated CAD layout has been
created and verified. A SEM micrograph of such an edit is shown in Figure 3-25. In
step 1, ion beam milling is used to cut through one wire and expose the side of a second
wire. In step 2, platinum is sputtered over the exposed wires to form a new connection.
The test results presented in this section were collected from a repaired die.
85
Step 1: Cut vertical trace using ion etch in a region
that exposes desired signal connection below.
Step 2: Deposit platinum to electrically connect top
vertical trace to bottom horizontal trace.
Figure 3-25. Example of circuit edit using focused ion beam (FIB).
3.6.4
Bias circuit
The bias circuit on the fabricated IC does not operate as predicted by simulation. A
design issue was found in the startup circuit, and the bandgap reference does not settle to
the desired operating point. When the startup circuit is removed, partial function is
restored. The bias circuit voltages shown in Table 3-7 were measured on a repaired die
using a multimeter. The measured values for NCAS and VLOW are outside the range of
operation for the ASIC circuitry, so the PBIAS signal must be driven externally. The
stimulator output current is proportional to a reference current generated in the bias
circuit, so the PBIAS voltage level must be adjusted for each chip.
Table 3-7: Comparison of simulated and measured bias circuit voltages.
Level
AVDD
PBIAS
VBG
NCAS
NBIAS
VLOW
Simulation
2.36 V
1.12 V
1.05 V
0.95 V
0.18 V
Measurement
(repaired)
Measurement
(PBIAS driven
to 2.45 V)
3.30 V
2.10 V
(2.45 V)
0.13V
0.60 V
0.29 V
1.02 V
N/A (no external pin)
0.01 V
0. 05V
86
3.6.5
Boost converter operation
3.6.5.1 Current sense amplifier and comparator
The current sense differential amplifier (Figure 3-10) and comparator (Figure 3-11)
circuits were tested simultaneously using the test setup shown in Figure 3-26. The BATT
pin is normally connected to the positive terminal of a battery, and so it is connected to a
fixed DC voltage level. The SHUNT pin voltage normally decreases in amplitude
relative to the BATT pin, and is driven by a function generator in this test.
Figure 3-26: Test setup for differential amplifier and comparator circuits.
The selected VBATT amplitudes represent the voltage range of a Zinc-air battery.
The PBIAS level is set externally to 2.56 V using a resistor voltage divider. As shown in
Table 3-8, the maximum switching frequency of the circuit is limited to 400 Hz and is too
low for use in the boost converter circuit. A layout error was found in the power control
switch that limits the differential amplifier bias current. This error is in a location that
cannot be reliably corrected using FIB circuit editing techniques.
87
Table 3-8: Measured performance of the boost converter current sense circuits.
BATT Level:
0.70 V
0.80 V
0.90V
1.10 V
1.30 V
SHUNT
Offset
0.60 V
0.75 V
0.85 V
1.05V
1.25V
SHUNT
amplitude
200 mVpp
100mVpp
100 mVpp
100mVpp
100 mVpp
Max Switching
Frequency
50 Hz
50 Hz
50 Hz
50 Hz
400 Hz
3.6.5.2 DMOS switch
The boost converter DMOS switching speed and current capability was tested using a
50 Ω resistor connected between a 3 V supply and the LX pin. As shown in Figure 3-27,
fast rise and fall times are achieved at the maximum expected switching frequency of 500
kHz. The switch is shown sinking 50 mA in this test, and the peak current during
operation will be limited to below this value by the inductor size and the boost converter
PWM on-time. The observed ringing is a result of inductance in the power supply test
leads.
Figure 3-27: MSS ASIC Boost Converter DMOS switching behavior with 50-ohm
load.
88
3.6.5.3 Open-loop operation
The open-loop boost converter operation operating using an external 1 kHz, 50% duty
cycle PWM control signal to drive the on-chip DMOS switch is demonstrated in Figure
3-28. A low PWM frequency is used in this open-loop test to prevent the output voltage
from exceeding the maximum VDS rating of the DMOS switch. The boost converter
charges to the maximum rated value of 90 V within 500 ms and meets the compliance
Boost Converter Output Voltage (V)
voltage requirements for the stimulator circuits.
90
80
70
60
50
40
30
20
10
0
0
100
200
300
400
500
Time (ms)
Figure 3-28: Boost converter charging performance using on-chip DMOS switch.
3.6.6
Stimulator
3.6.6.1 Negative Voltage Regulator
The negative voltage regulator was tested using the circuit shown in Figure 3-29.
HVDD was swept from 5 – 25 V using a benchtop power supply, and VZ = HVDD –
HVL was measured.
89
Figure 3-29: Test setup used to measure negative voltage regulator circuit.
As shown in Figure 3-30, the measured value of 1.3 V is lower than predicted in
simulation (3.15 V). This result is likely caused by internal bias circuit drawing less
current (20 µA) than required for proper operation (~40 µA). After an external 200 kΩ
resistor was connected from HVL to HVSS, VZ increases linearly to 4.3 V when HVDD
is 18 V. At this operating point, the external resistor draws an additional ~70 µA through
the circuit, about twice the 40 µA bias current needed for proper operation. For HVDD
values greater than 18 V, VZ remains constant at 4.3 V. The additional bias current
supplied by the external resistor increases with HVDD to a maximum of 430 µA when
HVDD = 90 V. This external resistor was used for the remainder of the ASIC benchtop
testing. In the MSS3, the external resistor is connected to HVSS through a HV transistor
so the external bias current is turned off between pulses and power consumption from the
HVDD supply is reduced.
90
Figure 3-30: Negative voltage regulator output for HVDD between 5 V and 25 V.
3.6.6.2 High-side switches
The performance of the fabricated high-side PMOS was tested using the circuit
shown in Figure 3-31. This test also verifies the operation of the voltage level shifter
and the floating HV logic circuits that drive the PMOS gates to HVDD and HVL. The
saturation current of the high-side extended-drain PMOS switches is expected to be 100
mA when VSG = HVDD – HVL = 3.3 V, based on SPICE simulations. This is roughly 5
times the maximum stimulation current drawn by the stimulator IDAC.
Figure 3-31: Stimulator High-side PMOS test circuit
91
HVDD and RL were selected to limit the current through the PMOS switch to 40 mA,
about twice the maximum stimulation current. As shown in Figure 3-32, the PMOS turnon time is short and VDS across the PMOS switch is about 1 V. The voltage level shifter,
floating HV logic, and HV PMOS circuits operate as expected.
Figure 3-32: Switching behavior of stimulator high-side PMOSFET with
HVDD = 20 V, f = 1 kHz, and 500 Ω resistor to GND.
3.6.6.3 Low-side DAC
An open-circuit layout error was found in the IDAC_MODE signal path that was
corrected using the FIB prior to circuit characterization.
The DC current transfer function of the low-side DAC was measured using the circuit
shown in Figure 3-33. The voltage applied to PBIAS was adjusted so the full-scale
IDAC code produces a 21 mA current. In the first test, HVDD is fixed at 5 V and the
digital IDAC code is swept across the range. HVDD is kept low to reduce on-chip power
dissipation in the cascode NMOS. As shown in Figure 3-34, the output current range
from 0 to 21 mA is achieved but linearity is poor.
92
Figure 3-33: ASIC low-side stimulator test circuit.
Figure 3-34: Stimulator low-side IDAC current transfer characteristic
with HVDD= 5 V.
This behavior is partly explained by the plot of the NMOS cascode transistor gate
drive voltage, IDAC_SENSE, shown in Figure 3-35. IDAC_SENSE should increase
93
linearly with the IDAC code. At both extremes of the operating range, inadequate
performance of the replica control loop results in IDAC nonlinearity. As suggested by
[38], IDAC linearity is less critical in biomedical applications and may be compensated
using a lookup table in software.
Figure 3-35: Measured IDAC_SENSE voltage driving gate of NMOS cascode across
IDAC code range with HVDD =5 V.
In the second test, the compliance voltage sensitivity is observed by repeatedly
measuring the current transfer function for several HVDD values. Figure 3-36 shows the
IDAC current is relatively insensitive to HVDD within a range that provides adequate
voltage headroom but does not result in excessive on-chip power dissipation. The undervoltage condition is demonstrated in the HVDD = 4 V trace when the output current is
reduced for large IDAC codes. In this case, the IDAC_SENSE voltage is near DVDD
and indicates to the microcontroller that HVDD should be increased. For HVDD values
≥ 15 V, the IDAC code was swept from zero to the value when the calculated on-chip
94
power dissipation reached 50 mW. A low IDAC_SENSE voltage indicates that HVDD
is too high and should be reduced.
Figure 3-36: Stimulator compliance voltage sensitivity for several HVDD values and
RL = 100 Ω.
3.6.7
Summary
The functionality of the current version of the MSS ASIC is summarized in Table
3-9. The chip requires two FIB circuit edits and three external resistors to obtain the
desired operation for the MSS3 device. The boost converter circuit must be operated in
the external mode using a PWM signal generated by the microcontroller.
95
Table 3-9: Summary of MSS ASIC circuit functionality
Circuit
Functional status
Required modifications
Bias circuit:
Partially functional
Remove startup circuit signal connection
using FIB.
Use external resistor divider to adjust
PBIAS level.
Boost Converter:
Internal current mode
External PWM mode
Non-functional
Functional
Use external PWM mode
None
Stimulator High Side:
PMOS switches
Level translator
Negative Voltage Regulator
Functional
Functional
Partially functional
None
None
Connect 200 kΩ external resistor
between HVL and HVSS.
Partially functional
Repair open circuit in IDAC_MODE
signal using FIB.
Stimulator Low Side:
96
4
Wireless Stimulation Bandage and System for a Large Animal
Model
The ASIC and supporting electronics have been developed for use in a wide range of
animal and clinical studies. In this chapter, a stimulator module and bandage for studies
involving large wounds similar to those found clinically in humans is presented.
4.1 Large Infected Wound Study
Infection is a common issue in both chronic wounds [42] and in acute wounds. In
particular, wounds suffered on the battlefield are particularly prone to infection [8, 43].
In addition, the rise in drug-resistant infections [44] has increased the need for alternative
interventions. Electrotherapy has been shown to impede the growth of biofilms in
wounds [11] and may be an effective non-pharmaceutical therapy for treating infection.
A pilot study using electrical stimulation to treat infected wounds has been proposed
[45] using a previously-validated infected wound model [46]. As shown in Figure 4-1,
2 wounds, 6 cm in diameter, will be created bilaterally (4 total) along the paraspinous
region of a female Yorkshire pig. All four of the wounds are inoculated with a
fluorescent strain of bacterium and allowed to colonize. Two of the wounds will be
randomly selected to receive electrical stimulation therapy while the remaining two will
serve as controls and receive non-stimulating bandages as standard care. The wound
dressings and stimulation bandages are periodically removed over a 28-day period as
required by the study protocol, then tissue samples are harvested from the wounds.
97
Figure 4-1: Approximate wound locations on porcine model for infected wound
study (2 wounds per side) [45].
From a medical device engineering standpoint, the research experiment is
summarized in Table 4-1. The stimulation requirements (6-8) are similar to previous
studies, but a physically larger device is necessary to accommodate the increased wound
size.
Table 4-1: Clinical Requirements for Infected Wound Study Device
#
Design Requirement
1
2
3
4
5
6
7
8
Experiment duration
Dressing change number, interval
Bandage weight
Bandage size
Unit cost
(not including ASIC)
Pulse Amplitude
Pulse Width
Pulse Frequency
10
Number of bandages in simultaneous use
11
Secondary dressing requirements
12
Bandage feedback
Specification
28 days
7 changes: D0, D1, D3, D5, D7, D14, D21
< 500 g
No more than 10 cm x 10 cm
Less than $100
0-21 mA, biphasic
0-200 µs
12-25 Hz
(Max duty factor 0.5%)
3 stimulating bandages
3 non-stimulating bandages
Each bandage must be self-contained and have no
external connections.
Bandage must indicate operation in non-visual
manner
The requirements of this animal study have implications that require a new
stimulation bandage design. In particular, requirements (1, 2, and 10) result in a large
number of bandages needed (4 wound locations * 7 changes = 28 bandages) during the
98
experiment. The stimulator modules may be refurbished, re-sterilized, and reused, but
there is insufficient turn-around time during the first 7 days of the experiment to do so.
Therefore, it would be preferable to re-use the stimulator portion of the device in-situ
between changes and replace only the substrate portion of the bandage that is in contact
with the skin. The bandage substrate presented in section 4.6.2 has been re-designed to
fit the wound model and to facilitate in-situ replacement on the stimulator module.
In order for the stimulator module to last the entire 28-day experiment, a battery with
greater capacity is needed. Fortunately, the size of both the substrate and the animal is
larger in this experiment, so different battery chemistries and packaging were evaluated,
as presented in section 4.2.
A method of communicating the device status to the clinical users is very important
so clinicians can confirm the stimulator is delivering current into the skin. The devices
will be obscured by secondary dressings and a protective jacket while in use, so a nonvisual indicator is needed. Low-power Bluetooth radios targeted for battery-powered
applications became commercially available in 2011 and provide the desired capability.
Based on these experimental requirements, the device concept shown in Figure 4-2
was created. Similar to previous design, 1 cm x 8 cm hydrogel electrodes are attached to
the bottom side of a flexible plastic substrate. If desired, an observation window may be
cut out from the substrate over the wound area. On the top side of the bandage, flexible
conductive traces are painted to connect the stimulation electrodes to a snap connector
that mates with the stimulator PCB module. The painted traces are connected to the
hydrogel electrodes through 3 paint-filled vias made through the substrate and spaced
99
~2.5 cm. A stimulator PCB with a battery is attached to the substrate via the snap
connector and double-sided adhesive film.
Figure 4-2: Stimulation bandage concept for large-wound studies.
4.2 Bandage Substrate
The flexible bandage substrate shown in Figure 4-3 provides a convenient and
repeatable method of positioning the stimulation electrodes over wounds. The substrate
also functions as an occlusive wound dressing to maintain tissue hydration. Springloaded snaps [47] are used to electrically connect the stimulator module to the electrodes
via painted conductive traces. The substrate is fabricated from Rogers Liquid Crystal
Polymer (1 mil thickness). This material exhibits a lower rate of moisture absorption
than polyamide substrates typically used in flex circuit applications.
The complete substrate fabrication process was co-developed with Jeremy Dunning.
100
Figure 4-3. Top (left) and bottom (right) views of bandage substrate
4.2.1
Bandage Electrode characterization
The surface resistivity of the painted traces is 55 Ω/sq (for a thickness of 50 µm) [48]
and the width is 0.8 cm, so the resistance is expected to be ~70 Ω/cm length. The
impedance to points along the length of the electrodes will not be uniform due to the
impedance of the painted trace, hydrogel electrode, and the limited number of vias
between the two. Electrode impedance was tested using the test fixture shown and
diagrammed in Figure 4-4. The electrode is contacted by eight equally-spaced copper
pads measuring 0.8 cm in width. The painted trace is contacted via the snap connector
permanently mounted on the bandage.
Figure 4-4: Test fixture and setup for substrate verification.
101
The electrode was hydrated using approximately 5 mL sterile saline, and then adhered
to the test structure. Using an LCR meter, the magnitude of the complex impedance was
measured between the snap connector and each of copper contact pads at frequencies of
100 Hz, 1 kHz, 10 kHz, and 100 kHz. As shown in Figure 4-5, the magnitude of the
electrode impedance is less than 4 kΩ for frequencies above 1 kHz and is relatively
insensitive to position. Given this value, up to 4 V of stimulator compliance voltage
headroom will be lost within the electrode trace per 1 mA of stimulation current. The
impedance decreases with increasing frequency to a value of about 3 kΩ at 10 kHz. The
observed behavior is consistent with a series resistor-capacitor model and reflects the
physical construction of a resistive carbon traces and capacitive hydrogel electrode, such
as modeled in [49].
Figure 4-5: Measured bandage substrate electrode impedance.
102
4.3 Battery Selection
The larger device required for this particular study can accommodate a larger battery
that will permit larger stimulation current amplitudes and will allow the device to be
immediately re-used between dressing changes without recharging the battery. The
relaxed battery size requirement allows battery chemistries and form factors other than
coin cells to be considered for this application. Lithium-chemistry cells offer high energy
density and those shown in Table 4-2 were evaluated. Non-rechargeable, cylindrical
lithium cells provide the largest capacity but the diameter would increase the overall
device thickness. Prismatic lithium polymer cells provide the desired capacity in a thin
form factor.
Table 4-2: Survey of high-capacity batteries for long-duration devices.
Type
Chemistry
Lithium
Thionyl
Chloride
(Li-SOCl2)
* Note: liquid
cathode is toxic
Lithium
(Li/FeS2)
Lithium
(CxF-Li)
Lithium Ion
(MnO2 Li)
Rechargeable
Lithium
Polymer
Rechargeable
Size
Dimensions
Weight
AA
Ø 14 x 50 mm
18g
AA
Ø 14 x 50 mm
14.5g
AAA
Ø 10 x 44 mm
8.55g
123A
Ø 17 x 34 mm
16.5 g
2/3A
Ø 17 x 34 mm
13.5g
RCR-123A
Ø 5 x 4 cm
Prismatic
35 x 25 x 5 mm
9g
51 x 34 x 6 mm
22g
Capacity
mAh
Connector
Voltage
3.6 V
Max.
Continuous
Current
100 mA
2100 mAh
Solder Tab
3000 mAh
Brand
Tadiran
Button
1.5V
3A
Energizer
1200 mAh
Button
1.5V
1.5 A
Energizer
1500 mAh
Button
3V
1.5 A
Energizer
1450 mAh
Button
3V
2.5 mA
Panasonic
600 mAh
Button
3.7V
400 mAh
2-pin plug
3.7V
2C
= 800 mA
Tenergy
1000 mAh
2-pin plug
3.7V
2C
= 2000 mA
Tenergy
Pearstone
103
4.4 Stimulator Module PCB
A block diagram of the MSS3 stimulator module PCB is shown in Figure 4-6. A
single-cell Lithium Polymer battery supplies power to the PCB and to the high-voltage
boost converter portion of the ASIC. A buck-mode switching regulator steps down the
battery voltage to 3.3 V and supplies power to the Bluetooth SoC microcontroller and
ASIC. A battery fuel gauge IC monitors the battery voltage and calculates the state of
charge using a proprietary algorithm. The microcontroller coordinates all functions on
the PCB and includes a radio for wireless communication. The ASIC boost converter
generates the high-voltage supply and the biphasic current-output stimulation. The PCB
connects to the bandage gel electrodes via mechanical snaps (not shown).
Figure 4-6: Block diagram of MSS3 Stimulator PCB
4.4.1
Wireless communication protocol and processor selection
A radio operating in a band near 2.4 GHz is desirable because a ¼-wavelength
antenna is only 31.25 mm long. Inverted-F and fractal (chip) topologies were considered
for this application because they consume the least PCB area and allow the antenna to be
closely integrated on the PCB or as a surface-mount component. A printed meandered-F
104
antenna was ultimately selected because it occupies the least board area [50], does not
require additional components, and has a generally omnidirectional radiation pattern [51]
suitable for use on an ambulatory animal.
The key selection criterion for the wireless protocol is the power consumption. The
power consumption is determined by several factors including active power consumption,
wake-up time, and maximum channel period. For this reason, Bluetooth Low-Energy
was chosen. The ANT protocol has similar characteristics to Bluetooth Low-Energy but
its proprietary status increases system development cost and limits its usage with COTS
devices such as smart phones. Furthermore, the CC2541 System on a Chip (SoC) serves
as a general-purpose microcontroller with integrated Bluetooth LE radio.
Table 4-3: A review of microcontrollers with built-in 2.4 GHz radios.
Protocol
Reference Part
Frequency
Band
Year
Intro.
TX
Power
Data
rate
Wakeup Time
Notes:
>100 ms
Max
Connection
Interval
65 sec
Bluetooth 2.0
CC256x [52]
2.4 GHz
1994
1-3
Mbps
Zigbee
(802.15.4)
CC2530Fxx
[53] [54]
ANT
CC2570 [55]
Bluetooth
Low-Energy
(BLE)
CC254x
nRF51822 [56]
915 MHz
2.4 GHz
2003
39 mA
@
4 dBm
29 mA
@
1 dBm
250
kbps
15 ms
250-786 sec
Proprietary
specification
Integrated
8051 MCU
Proprietary
Interface
Integrated
8051 MCU
(8 bit)
2.4 GHz
2005
2.4 GHz
2011
18.2 mA
@
0dBm
1 Mbps
3 ms
240 sec
2.4 GHz
2011
10.5 mA
@0dBm
1 Mbps
3 ms
240 sec
1Mbps
500 ms
Not available
as SoC
ARM M0
(32 bit)
4.5 Firmware
In order to maintain the required timing for the Bluetooth protocol, the firmware is
written within a Real-Time Operating System (RTOS) provided with the microcontroller
105
development kit [57]. In an RTOS, code is executed as a series of tasks in order of
descending priority. This helps ensure that the most time-critical functions are always
executed within the required time. In this case, the BLE protocol stack has strict timing
requirements in order to maintain the connection with the host. After all tasks have been
executed, the microcontroller enters a low-power mode for the remainder of the current
cycle.
4.5.1
Stimulation State machine
In the MSS2 design, a state machine activated the boost converter and stimulator
sequentially to generate each pulse. In the MSS3 architecture, the boost converter runs
independent of the stimulator and the output voltage is continuously adjusted to maintain
the compliance voltage required to deliver the programmed stimulation current.
4.5.1.1 Boost converter controller
A block diagram depicting the operation of the boost converter controller is shown in
Figure 4-7.
Recall from 3.4.3.3 that, in steady state, the voltage applied to the stimulator cascode
devices should be approximately mid-scale, indicating there is sufficient compliance
voltage headroom to force the commanded current. The analog voltages
V(IDAC_SENSE) and V(HVDD_SENSE) produced by the MSS ASIC are converted to
digital values by the microcontroller ADC. The V(IDAC_SENSE) signal is sampled at
the end of each stimulation pulse, and V(HVDD_SENSE) is continuously sampled. Both
digital values are averaged using 16-point FIR filters implemented in firmware.
IDAC_SENSE is compared to a mid-scale threshold to produce an error value that is
106
ultimately used to control the HVDD level. The HVDD setpoint value is compared to the
sensed HVDD value at a 100-Hz rate and the result determines the state of the
BOOST_EN control signal sent to the ASIC.
Figure 4-7: Block diagram of boost converter controller.
4.5.1.2 Stimulation controller
Stimulation pulses are controlled by setting the DAC output of the ASIC to the
appropriate value, then setting the STIM_EN and CHSEL pins to drive current in the
selected direction. Although there are software timer functions available within the
RTOS, the execution time cannot be precisely controlled. Given that a Bluetooth
timeslot is only 625 µs, it is undesirable to stall the program execution for up to 400 µs
total during each stimulation pulse pair. For accurate timing of the stimulation pulse, the
pulse generation should not rely on or be interrupted by other RTOS tasks.
The stimulation pulse timing signals, STIM_EN and CHSEL, are generated entirely
within the microcontroller hardware peripherals using two hardware timers operating in
tandem. The timers also trigger two CPU interrupts that call functions to set the
stimulator current for positive and negative pulses. This technique uses the on-chip logic
that is commonly used in infrared transceivers to modulate a carrier wave with
communication symbols. The stimulation pulse control signal generation process is
107
shown in Figure 4-8. Timer 3 operates as an 8-bit modulo counter with a period of 250
µs. Timer 3 output compare channel 0 produces the clock input to Timer 1 and has a
period of 250 µs. Timer 3 channel 1 sets the pulse width for the positive and negative
stimulation pulses. Timer 1 operates as a 16-bit modulo counter with a period from 1 ms
to 32,000 ms that sets the stimulation pulse interval. (The maximum pulse interval is
limited in software to 1000 ms.) The pulse width of Timer 1 channel 1 is fixed to 500 µs,
the maximum combined width of two stimulation pulses. Timer 1 channel 2 produces the
CHSEL signal used to select either a positive or negative current pulse. Timer 1 channel
2 also produces an interrupt on the falling edge of the first STIM_EN pulse that sends the
negative pulse amplitude value to the SPI serial port. Timer 1 channel 3 produces a
complementary interrupt on the falling edge of the second STIM_EN pulse that sends the
positive pulse amplitude value in preparation for the next pulse pair. If mono-phasic
current pulses are desired, the negative pulse width may be set to zero in the Timer 3
channel 1 register and a zero amplitude value loaded in the SPI port register.
Figure 4-8: Generation of stimulation pulse timing signals using hardware timers
108
The large-timescale stimulator timing (minutes per hour and total hours) is
straightforward and is controlled within a 1-Hz RTOS task that implements a Real-Time
Clock function.
4.5.2
Bluetooth Interface
The Bluetooth wireless interface is used to program the stimulation parameters and to
monitor the stimulator operation while the device is in use on an awake animal. The
protocol stack shown in Figure 4-9 is provided by the manufacturer as a pre-compiled
firmware library. The custom firmware written for this device is contained within the
application layer, GAP (Generic Access Profile), and GATT (Generic ATTribute)
profiles. The application layer contains the RTOS tasks that implement the boost
converter and stimulator controller state machines. The GAP role profile determines how
the device will function within a Bluetooth network and sets the communication timing
constraints. GATT profiles are collections of device data (called characteristics) such as
the stimulation parameters, battery status information, and device measurements.
Figure 4-9: Bluetooth Protocol Stack (from [57])
109
Bluetooth devices can operate in one of four GAP roles: broadcaster, observer,
peripheral, or central. This device operates as a peripheral, meaning it will periodically
advertise itself to the network and will allow other devices to pair with it. This mode is
more power-efficient than the broadcaster role because data is broadcast only when
another device is listening. To communicate with the stimulator module, a host device
(such as a PC a BLE dongle or the base station presented in 4.7) operates in the central
role, meaning it listens for advertising devices and then creates a connection (called
pairing) to communicate with them. In traditional networking terms, a peripheral device
operates as a slave and a central device as the master.
Regardless of the GAP role used, each device within the network may operate as
either a data client, data server, or both. As shown in Figure 4-10, the base station
(operating in the central GAP role) is a client that reads from and writes data to a bandage
server (operating in the peripheral GAP role).
Figure 4-10: Communication scheme for transferring parameters and
measurements between base station and bandages.
For common types of data (such as the current time or a heart rate measurement), data
structures called Services are pre-defined by the Bluetooth standard [58]. There is
110
currently no existing standard Service for a stimulator device, so a custom service was
created to organize the stimulation parameters. The parameters and measurements
available through the Bluetooth protocol are listed in Table 4-4. The GATT profile
implementation of this service is included in Appendix C.3.
Table 4-4: Stimulator control parameters available through Bluetooth wireless link.
Index
1
2
3
4
5
6
7
8
9
10
Parameter
Battery Voltage
Battery SOC
Positive Pulse Amp.
Negative Pulse Amp.
Positive Pulse Width
Negative Pulse Width
Pulse Period
Stim Minutes per Hour
Stimulation Hours
Stimulation Enable
Data Type
Byte
Byte
Byte
Byte
Byte
Byte
Byte
Byte
Byte
Byte
LSB Unit
20 mV
0.5 %
0.33 mA
0.33 mA
1 µs
1 µs
500 µs
1 min
6 hr
1 = run
0 = stop
Data Range
0–5V
0-99 %
21 mA
21 mA
200 µs
200 µs
1-128 ms
1-60 min
1-1000 hr
0-1
Client Access
Read
Read
Read/ Write
Read/ Write
Read/ Write
Read/ Write
Read/ Write
Read/ Write
Read/ Write
Read/ Write
4.6 Device assembly
4.6.1
MSS3 Stimulator Module Assembly
The MSS3 stimulator module PCB is assembled using standard surface mount reflow
techniques and lead-free solder paste. The microcontroller is programmed using the
CC Debugger programming tool [59], SmartRF Flash Programmer [60] program, and a
custom probe cable. The custom cable used to connect the programmer to the PCB
programming pins is shown in Figure 4-11.
Figure 4-11: MSS3 microcontroller programmer connection using PCB probe.
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The packaged MSS ASIC is hand-soldered to the PCB after the other module
components have been electrically tested to avoid potentially damaging this high-cost
part. The plastic battery connector is installed, and female component of the electrode
snaps is soldered to the bottom side of the PCB, as shown in Figure 4-12. The
completely-assembled PCB is tested using the procedures described in Section 4.8.
Figure 4-12: Snap connectors mounted on bottom side of MSS3 PCB module.
The assembled and tested PCB is covered with a conformal coating of parylene to
protect the components from moisture and other potential contamination in the wound
environment. This step was performed by Jeremy Dunning of the APT Center. A thin
film chemical vapor deposition process [61] is used to apply the layer of p-xylylene
polymer (trade name Parylene®) with a target thickness of 25 µm. The PCBs are
suspended in the reaction vessel to apply a uniform coating to all surfaces. The polymer
coating is manually removed from the battery connector terminals and electrode snaps to
re-expose these electrical connections. Electrical performance of the module is rechecked after the coating process.
The MSS3 stimulator module PCB is integrated with the Lithium Polymer battery
using clear heat shrink tubing [62] with a pre-shrunk diameter of 38.1 mm. The
112
arrangement of the PCB and battery is shown in Figure 4-13. This semi-permanent
attachment method does not damage either component and provides an additional layer of
protection from the wound environment.
Figure 4-13: MSS3 Stimulator Module PCB Assembled with Battery.
4.6.2
Device Sterilization
The MSS3 stimulator modules and substrates must be sterilized prior to use in animal
or human studies. Medical instruments are commonly sterilized using steam in an
autoclave at 121 ˚C -132 ˚C [63]. The LCP substrate [64] is rated for continuous
operation at 150˚C and has low water absorption (0.04 % at 23˚C over a 24 hour period)
and may be suitable for steam sterilization. However, the conductive paint used for the
traces is not indicated for use at high temperature.
Ethylene Oxide is a low-temperature, dry gas sterilization process that takes place at
temperatures ranging from 30 ˚C - 60 ˚C. However, the process is hazardous and
requires special validation after processing to ensure no carcinogenic compounds [65]
remain on the substrates or devices before use.
Ultimately, the STERRAD® process was selected for processing the MSS3 devices
and substrates. The process uses hydrogen peroxide plasma and operates at a temperature
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range from 47 ˚C -56 ˚C [66]. In addition, this process is specifically indicated for
sterilizing Lithium polymer batteries, and this will permit the stimulator and battery to be
sterilized simultaneously.
The stimulator modules and substrates are individually packaged in Tyvek® pouches
[67] specifically designed for low-temperature sterilization processes. The packed
substrates are shown in Figure 4-14, and the stimulator module PCBs are packed in the
same manner.
Figure 4-14: Bandage substrates packaged for sterilization.
4.6.3
Intra-Operative Bandage Assembly
The reusable MSS3 stimulator module is connected to a fresh bandage substrate
during the initial surgery and each dressing change. Using sterile handling, the modules
and bandages are removed from their packaging and delivered to the surgical field. The
stimulator modules are sterilized with the battery unplugged, so this connection must be
made prior to use. The adhesive strip on the top size of the bandage is exposed, then the
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module mounted and the snap connections made. The device is now ready for
application to the animal and activation using the radio interface.
4.7 Wireless Base Station
In previous research studies using stimulation bandages, devices were occasionally
damaged or removed by the animal. Future versions of the stimulation bandage may
include additional sensors for monitoring the wound healing process, and this feedback
could be provided to the clinician without creating a physical connection to the device on
the awake animal.
The stimulation bandage is controlled by manipulating the GATT profile values using
either of two PC software programs that are currently available from Texas Instruments
[68, 69]. These programs provide limited functionality and have three key limitations
that reduce their effectiveness within the laboratory. First, there is no method to display
the binary-coded data values read from the bandage in a meaningful representation to the
clinical user. Second, neither program allows more than one simultaneous connection, so
they cannot be used to monitor two operating bandages as required by the animal study.
Third, the programs lack the ability to continuously log values read from the bandage
(such as the battery voltage or the stimulator compliance voltage), and the
microcontroller on the stimulation bandage has limited data memory to store
measurement data.
To address the limitations of the existing BLE interfaces, the wireless base station
shown in Figure 4-15 is proposed. Each MSS bandage communicates using their
Bluetooth radio to a central host radio. Although only two bandages will be active on an
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individual animal, the Bluetooth network supports up to 6 simultaneous connections and
there may be multiple animals in the facility at a given time. The base station host radio
is controlled by an application processor that requests data from each of the connected
bandages at regular intervals (typically 1 set of measurements every 30 seconds). The
application processor then stores the received bandage data to a SD flash memory card
for post-experiment analysis. The processor also transmits the measurements to an online
data server through a WiFi radio connected to the wireless network within the facility.
Figure 4-15: Communication network between stimulator modules, base station,
and internet data server.
4.7.1
Base station construction
The base station is constructed using components based on the open-source Arduino
microcontroller development platform [70]. The bottom board in the stack is an Arduino
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Mega 2560 microcontroller “board” containing an Atmel ATmega2560 8-bit
microcontroller that executes the application firmware. Three “shields” stack on the
microcontroller board: a WiFi shield [71] to connect to a wireless internet access point, a
BLE shield [72] to communicate with the MSS3 stimulator modules, and a LCD shield
[73] to provide display the four MSS3 battery voltages to investigators while in the
facility.
A piezo transducer provides audible cues to clinicians during device activation,
and a fuel gauge module monitors the base station battery charge. The base station is
powered from a PC through a USB cable during testing, and a 2200 mAh rechargeable
Lithium Polymer battery pack when deployed in the animal facility. The base station
hardware is shown in Figure 4-16.
Figure 4-16: Wireless base station hardware PCB stack.
The PCB stack is mounted in a plastic enclosure to protect the electronics while
inside the animal facility. The base station is then placed outside the animal pen and
requires no clinician interaction after initial activation of the bandages.
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4.7.2
Base station operation
The two main functions of the base station are to program the stimulation parameters
into each bandage and to monitor performance during the experiment. The ability to
program the devices wirelessly avoids the use of exposed signal pins on the stimulator
module PCB that may be damaged during use on the animal. Continuous remote
monitoring improves research data quality by helping to identify when bandages have
been displaced from the wounds or the device battery charge is low.
4.7.2.1 Programming bandage stimulation parameters
The bandage programming process is diagrammed in Figure 4-17. Device
stimulation parameters are loaded from the PARAMS.CFG text file stored on a micro SD
card installed on the WiFi shield. Users may modify stimulation parameters using a text
editor on a PC, then transfer them to devices using simple prompts on the base station
LCD display. The stimulation parameters are then written into each bandage according
to the BLE bandage profile given in Appendix C.3.
Figure 4-17: Procedure for programming bandage stimulation parameters.
4.7.2.2 Wireless bandage data transfer
Each bandage transmits its performance information within the characteristic ID as
part of the periodic BLE connection maintenance. The data samples for each device are
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stored in separate files on the SD card. The samples are also uploaded to a secure
internet server via the WiFi shield for continuous experiment monitoring while outside
the animal facility. The base station reads measurements from the Bluetooth profile of
each bandage every 30 seconds and stores the data in CSV text files on the micro SD
card. These log files may be transferred to a PC using a SD card reader at the conclusion
of the experiment. The base station uploads a measurement from each bandage to an
internet data server every 10 minutes.
4.7.2.3 Base Station Clock Synchronization
The Mega 2560 processor board is constructed using a 16-MHz resonator [71] with
initial frequency tolerance of 0.5%. Over a period of 28 days, the accumulated time error
could potentially exceed 200 minutes. The frequency error is compensated in two ways:
first, a calibration value is loaded into a clock control register from EEPROM to reduce
the timing error within the microcontroller. Second, the base station connects to the US
Naval Observatory Master Clock time server [74] using the WiFi shield and downloads
the correct time (within 1 second) once per hour.
4.8 Benchtop Testing
The assembled stimulator module PCB shown in Figure 4-18 measures 31 mm tall by
38 mm wide and weighs 4.9 grams. The two-layer PCB is constructed using 0.062-in
thick FR-4 laminate and 1-oz copper traces (equivalent to 1.4 mil thick [75]). The
schematic drawings are provided in Appendix C.1 and the assembled PCB is shown in
Figure 4-18.
119
Figure 4-18. MSS3 printed circuit board
(actual size 38 mm x 31 mm)
There are three main objectives in the stimulator module PCB benchtop testing. First,
the function of the MSS ASIC when battery-powered and operated by the microcontroller
was confirmed. Second, the wireless performance of the stimulator module PCB was
measured. Third, the operating current of the device was measured under several
operating modes and the battery lifetime was estimated.
4.8.1
ASIC validation
4.8.1.1 Boost converter
The Lithium Polymer battery used in this version of the device has a low internal
resistance, and the current-limiting boost converter circuits designed for coin batteries on
the MSS ASIC are not necessary. More importantly, the nominal cell voltage of a
Lithium Polymer battery is outside the designed input voltage range of the current
sensing amplifier. Therefore, the boost converter on the MSS3 is operated using a PWM
signal generated by the microcontroller. As shown in Figure 4-19, the boost converter
output charges to about 35 V within a few ms when the MSS3 is powered by a Lithium
Polymer battery using a PWM frequency of 125 kHz and duty factor of 50%. Large
voltage ripple occurs when the output current is small because the boost converter outer
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control loop code is executed once per 1 ms. In section 4.8.1.2, it will be shown that the
variations in the boost converter output (HVDD) do not affect the current delivered to the
electrodes.
Figure 4-19: Battery-powered MSS3 boost converter output charging waveform
(VBATT = 4.15 V) .
4.8.1.2 Stimulator
Biphasic operation of the MSS3 stimulator is demonstrated in Figure 4-20 using a
1 kΩ resistor load and the stimulation parameters in Table 4-5. The STIM_EN signal
controls the pulse width of the positive (cathodic) and negative (anodic) pulses. The
CHSEL signal determines if the current flowing from ELECA to ELECB to positive
(CHSEL = L) or negative (CHSEL = H). The stimulator outputs ELECA and ELECB
remain biased at HVDD between pulses. The common mode level drops during the
stimulation pulses due to trace impedance. The current delivered to the load is constant
during each pulse despite voltage fluctuations on the HVDD supply.
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Figure 4-20: Demonstration of the MSS3 stimulator delivering a biphasic pulse.
Table 4-5: Stimulation Parameters used in MSS3 stimulator demonstration.
Parameter
Positive Pulse Current
Positive Pulse Width
Positive Pulse Current
Positive Pulse Width
Pulse Period
HVDD
Value
5 mA
100 µs
2.5 mA
200 µs
80 ms
35 V
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4.8.2
Wireless Communication
4.8.2.1 Range test
The Bluetooth radio is specified to operate at distances up to 10 m, but actual
performance is extremely dependent on the PCB layout and device construction. In this
application, the typical distance is only 3 m, and the radio transmitter and receiver power
settings will be reduced automatically by the radio to extend battery life. In this
experiment, the BLE USB dongle is attached via a USB cable to a PC and the MSS3
stimulator module is positioned at a variety of distances in an office environment. The
stimulator module is powered from a battery attached to PCB with heat shrink in the
same manner as the animal-ready device. The RSSI value reported by the SoC varied
within a few codes, so repeated measurements (N=5) were taken at each distance to
determine the average received signal strength (RSSI). The internal RSSI measurement
range for the part is -79 dBm to -15 dBm [76] with an accuracy of ± 6 dBm. The plot
shown in Figure 4-21 indicates that a signal of -65 dBm is received at a distance of 3.05
m.
Figure 4-21: Self-reported received signal strength (RSSI) from CC2541 radio as a
function of distance from USB dongle.
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4.8.3
Operating current test
The battery current is measured for each of the three operating modes of the device:
stimulation, sleep, and communication. The stimulator current consumption was
measured using a 4.1 V power supply in place of the LiPoly battery and the stimulation
parameters previously listed in Table 4-5. When a 450 mAh LiPoly cell is used, the
MSS3 will operate for 161 hours. To achieve a 1-week battery lifetime, the stimulation
time should be reduced from 10 to 9 minutes per hour.
Table 4-6: Measured operating current and projected device lifetime for several
compliance voltage levels.
Mode
Standby
Communication
Stimulation
Measured
Current
300 µA
8.60 mA
average
15 mA
Total calculated
average current
consumption
Estimated lifetime
(450 mAh battery):
% of operation time
50 min/ hr
= 83%
15 ms/ 30 sec
= 0.05%
10 min/ hr
= 17%
Contribution to
Average Current
250 µA
4.3 µA
2.55 mA
2.8 mA
161 hr
4.9 Clinical Validation
4.9.1
Method
An acute test of the surface stimulation bandage was performed to validate the
electrical and mechanical function of the device. First, the infected wound model is
created on an anesthetized pig according to the IRB-approved study protocol. The
bandage is affixed to the animal skin using pre-cut pieces of sterile hydrocolloid dressing
[77] as shown in Figure 4-22 . Light pressure is applied to the dressing for approximately
2 minutes (per the product instructions) to increase the bandage adhesion. Bonding is
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confirmed by manually applying a small sheer stress load to the ventral edge of the
bandage.
Figure 4-22: Photograph of bandage with adhesive hydrocolloid material.
4.9.2
Bandage conductivity test
The first test is to measure the skin impedance through bandage electrodes using a
benchtop LCR meter connected to the electrode snaps. The stimulus voltage is a 1-V sine
wave with frequency varying from 100 Hz – 100 kHz and the resulting current is below
the threshold of sensation. Impedance was measured across the electrodes five minutes
after the substrate was applied to the skin. Two substrates were tested using this method,
and the magnitude of the measured impedance is lower than 6 kΩ above 10 kHz for both
samples, as shown in Figure 4-23. Approximately 3 kΩ of the measured impedance is
attributed to the substrate traces and electrodes. A trace on substrate 1 partially cracked
during the test, and a large impedance magnitude of 64 kΩ was measured at 100 Hz. This
test confirms the bandage is making reliable electrical contact with the skin and the
impedance is within the range predicted by benchtop tests.
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Figure 4-23: Measured MSS3 substrate electrode impedance measured in an acute
test on living pig skin.
4.9.3
Acute stimulation test
The second electrical test is to activate the bandage stimulator using the on-board
wireless Bluetooth radio link and a laptop PC to verify the electrical operation of the
device in-situ. The stimulator was configured to generate 50 µs pulses of 5 mA and
-2.5 mA to the skin, and the circuit shown in Figure 4-24 was used to measure current
flowing from each electrode. An oscilloscope was used to record the stimulation current
pulse flowing into the skin (RSKIN) through two 10 Ω sense resistors (RTEST) mounted on
the stimulator module PCB. These current sense amplifiers are unipolar and only
measure positive current values. For this test, commercial electrodes [78] were used in
place of the MSS3 substrate to connect the stimulator to the skin.
126
Figure 4-24: Acute test schematic for measuring current flowing through tissue.
The measured electrode voltage and current waveforms are shown in Figure 4-25.
Compared to the test using a resistor load (Figure 4-20), the acute test shows the effect of
the complex impedance associated with the hydrogel electrodes and skin on the electrode
voltage waveform. The positive and negative current pulses are delivered with the
expected 50 µs width, but a third pulse is created between the two when the high-side
PMOS devices switch state. This behavior is consistent with a series capacitive element
in the electrodes or skin discharging back into the stimulator. A smaller cathodic current
pulse is generated when t≈1500 µs and the PMOS switch to their original state.
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Figure 4-25: Electrode voltage and current waveforms measured in acute
stimulation test.
4.10 Summary
Based on benchtop and acute in-vivo test results, the MSS3 device is suitable for use
in long-term animal studies that require a programmable stimulator with wireless
communication capability. Battery lifetime is dependent on the capacity of the LiPoly
cell used and the selected stimulation parameters. A MSS3 that uses a 450 mAh battery
to stimulate for 9 minutes per hour is expected to operate for 7 days before recharging is
necessary.
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5
Conclusions and Future Work
Wound electrotherapy is currently applied in short sessions in a clinical setting,
typically no more than 1 hour in duration, on a daily basis. Treatment duration is limited
by caregiver resources and patient acceptance (e.g., restricted movement due to tethers).
The objective of this research was to develop a wearable stimulation system suitable for
research studies that could be readily adapted for use in a range of applications and
species. The new medical devices presented in this work provide a pathway to new
alternatives that will deliver continuous electrotherapy in a less-invasive manner.
5.1 Achievements
The first bandage presented in this work was designed for use with a rat wound model
and consists of a stimulator PCB module constructed from off-the-shelf components and
is powered by a small button cell battery providing at least seven days of continuous use.
This voltage-mode device generates stimulation pulses that are 10 – 90 V in amplitude,
10 – 200 µs in width, and 12 – 25 Hz in frequency. A disposable plastic electrode
substrate was co-developed that transmits the stimulation current from the PCB to the
skin. Stimulation was typically applied to wounds for 10 minutes every hour for one
week, and then the device and wound dressings were replaced.
An ASIC has been developed using the OnSemi 0.7-µm I2T100 process and is
capable of operation up to 100 V. A high-gain, current-mode boost converter addresses
challenges associated with efficient generation of the large amplitude compliance voltage
(up to 90 V) from a small battery with limited output current capability. In particular, the
control circuits limit the peak inductor current on a cycle-by-cycle basis to avoid battery
brown-out. A biphasic current mode stimulator was also demonstrated with ±21-mA
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output range, 0.33-mA resolution, and a voltage headroom requirement of 3.5 V at full
scale output. The digitally-controlled current is programmed using a SPI serial interface.
The second stimulation bandage uses this ASIC and is designed for use on larger
wounds, up to 6 cm in diameter. A rechargeable lithium polymer battery allows a single
stimulator module to be used for an entire 28-day study. The battery may be quickly
recharged as needed during weekly dressing changes. The disposable electrode bandage
portion of the device is easily replaced in-situ by the clinician. Continuous monitoring of
the delivered stimulation current while the device is in place on an animal is achieved
using a microcontroller with a built-in Bluetooth Low Energy radio. Performance
information from up to six devices is recorded to a wireless base station located outside
the animal cage and may also be remotely accessed by research personnel.
A summary of the key features of the two new devices are presented in Table 5-1.
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Table 5-1: Electrotherapy Device Summary.
Target species
Bandage1
Substrate
Traces
Electrodes
Connector
Stimulator
PCB Dimensions
Mass
(w/o battery)
Communication
Battery Type
LG TEC-ELITE [15]
TENS device
ISSD [16]
Previous work
MSS1
Discrete device
MSS3
Integrated device
Human
Rabbit
Rat
Pig
Individual
Wire leads
3.8 cm x 3.8 cm
plug
Polyimide
Platinum
ECG gel, 1 cm x 2.5 cm
Folded tab
LCP
Nickel under acrylic
Hydrogel, 1 cm x 2.5 cm
Conductive tape
LCP
Carbon under acrylic
Hydrogel, 1 cm x 8 cm
Snap
(handheld)
N/A
44 mm x 32 mm
6.8 g
25 mm x 25 mm
3.4 g
38 mm x 31 mm
4.9 g
LCD
Rechargeable 9V
IrDA (infrared)
Lithium (CR1632)
Wired Serial
Zinc-Air (675 size)
130 mAh @ 3.0 V
< 7 days
Yes
Voltage
RC
620 mAh @ 1.4 V
7 days
Yes
Voltage
RC
Bluetooth 4.0 (RF)
Lithium Polymer
(model 063048)
500 mAh @ 3.7 V
28 days (rechargeable)
Yes
Current
(biphasic rectangular)
Battery Capacity
Lifetime
Reusable
Pulse Type:
N/A
24 hrs
Yes
Current
Biphasic square
1. Developed with Jeremy Dunning
5.2 Future work
Further technological development is needed to adapt the existing device for
additional pre-clinical studies using animal wound models and to prepare a human wound
electrotherapy device suitable for widespread clinical use.
In future studies using a small mammal (e.g. rodent) wound models, a stimulator
module using the MSS ASIC could be designed to take full advantage of the currentmode boost converter circuitry using small batteries with limited output current
capability. Given the small size of the animal, a reusable stimulator module with a
flexible PCB would reduce unwanted skin irritation and increase animal comfort.
131
Future versions of the ASIC could include additional circuits to adaptively control the
compliance voltage supplied by the boost converter based on a feedback signal either
generated within the stimulator (such as IDAC_SENSE) or one based on a direct current
measurement. If required by the electrotherapy study, the stimulator current resolution or
range may be easily expanded.
The stimulation pulse waveform timing signal
generation could also be integrated on-chip with the addition of extra SPI serial port
registers and logic circuits. This would reduce the microcontroller interaction and may
reduce overall power consumption.
The ASIC-based stimulator module with wireless communication capability can
accommodate sensors to provide continuous information to researchers about the state of
wound healing in a particular subject. For example, a temperature sensor would indicate
the degree of inflammation, and an pH sensor might detect changes in tissue oxygenation.
Although it was not the focus of this work, the design and manufacturing methods
used to fabricate the electrode substrate portion of the devices need further refinement to
enable large-scale production. A clinic-ready device will likely have the stimulator
electronics assembled directly on a flexible bandage.
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Appendix A: Supplement Information for Chapter 2 (MSS2)
A.1: MSS2 Schematic
133
Appendix B: Supplemental Information for Chapter 3 (ASIC)
B.1 MSS ASIC Test Fixture Schematic
134
Appendix C: Supplemental Information for Chapter 4 (MSS3)
C.1: MSS3 Module Schematic
135
C.2 Stimulation Bandage GATT profile
136
C.3 MSS3 Parameter programming procedure using TI BLE Device Monitor
Software: TI BLE Device Monitor V 1.1 or later.
Equipment: TI CC2540 USB Dongle (included with CC2540 Mini Development Kit)
loaded with the CC2540_USBdongle_HostTestRelease_All.hex firmware image.
Step 1: Install USB dongle into PC and start the Device Monitor Application. A red
LED on the dongle should be illuminated and the dongle address should appear in the
BLE Network pane.
Step 2: Make sure the MSS3 bandage battery is plugged in, then click scan in the
BLE Network pane. The MSS3 device should appear under the Host device.
137
Step 3: Double-click on the MSS3 device, and the program will discover the
characteristic parameters used to program the device.
Step 4: To read a parameter value, double-click on the desired characteristic name (in the
mnemonic column). To write a parameter value, double-click on the value column and
enter the desired value. (If an entry box does not appear, then the parameter is read-only.)
It is good practice to read back a parameter after writing a new value to verify the change
has taken place.
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C.4 Base station Arduino programming procedure
Software: Arduino IDE (available from www.arduino.cc)
Equipment: USB cable, Base Station hardware
Step 1: Connect Arduino Mega 2560 processor board to computer using the USB cable.
A serial port driver may automatically be installed.
Step 2: Start the Arduino IDE software and open the base station project:
File SketchbookSketchesbase_station.
Step 3: Verify the correct serial port and board are selected:
Tools Serial Port <COM port>
Tools Boards Arduino Mega2560 or Mega ADK
Step 4: Click the upload button (right arrow) to compile and upload the firmware to the
Arduino Mega2560 board. It may take a few moments to compile the project before
uploading begins.
Step 5: If desired, open the serial monitor to verify the base station boots successfully.
Otherwise, the base station may be disconnected from the computer.
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C.5 Base station BLE112 programming procedure
Software: BlueGiga BLE Update, bglib_master firmware image (project.bgproj).
Equipment: CC Debugger with mini 10 pin ribbon cable (included with CC2540
Mini Development Kit), BLE Shield, Arduino processor board, USB cable.
Note: The BLE112 module on the BLE Shield is controlled using commands issued
from the BGLIB library on a host processor. This procedure should only be used to
program a BLE112 on a newly-fabricated BLE shield.
Step 1: Install the BLE shield on an Arduino microprocessor board to safely supply
power to the BLE112. Connect the CC Debugger to the BLE shield using the ribbon
cable (observe polarity markings for pin 1),
Step 2: Connect the Arduino and CC Debugger to the PC using one standard USB cable
and one “mini B” USB cable. Press the button the CC Debugger and observe the LED
change to green, indicating a successful connection to the BLE112.
Step 3: Start the BlueGiga BLE Update program and select the correct port and firmware
image. Leave the license key field blank.
Step 4: Click “Update” and wait for the CC Debugger to program the BLE112. The
dialog window will turn green when the part is successfully programmed.
Step 5: Unplug the ribbon cable from the BLE shield, then unplug the CC Debugger
from the PC. Leave the Arduino board stack connected to the PC.
Step 6: Use the serial monitor within the Arduino IDE or another terminal program to
observe the base_station program serial output stream. Verify the BLE112 responds
during the initialization routine.
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C.6 MSS3 CC2541 programming procedure
Software: TI Flash Programmer, IAR Embedded Workbench IDE.
Equipment: CC Debugger with ribbon cable and custom 5 pin probe cable
Step 1: Connect the CC Debugger to the PC using a USB cable. The LED should
illuminate red, indicating power is present but no target device has been found yet.
Step 2: Connect the MSS3 PCB to the CC Debugger using the ribbon cable and 5 pin
probe. Apply power to the board using a battery or lab supply. Press the button the CC
Debugger and observe the LED change to green, indicating a successful connection to the
CC254x microcontroller.
Step 3: Start the IAR Embedded Workbench IDE application. Load the project
workspace, modify the code as necessary, then re-compile by pressing F7.
Step 4: The on-chip debugger may be invoked by pressing Ctrl-D or selecting “Debug
Download” from the menu. Start program execution by pressing F5, and terminate the
debug session by pressing Ctrl-Shift-D.
Step 5: To program a firmware image into the Flash memory, start the TI Flash
Programmer application. Note that the debugger must not be active in the IDE in order to
use the programmer. The CC254x device on the MSS3 should appear in the System-onChip window.
Step 6: Select “Erase, Program, and Verify” option from the Actions menu, then click the
“Perform Actions” button to program the part. A message will appear in the text box
below the button indicating if the part was successfully programmed.
141
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