Journal of Electrical Engineering
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ENVELPPE CYCLOCONVERTER BASED ON INTEGRAL HALF
CYCLE SELECTION AND HALF CYCLE OMISSION TECHNIQUE
(1)
Mohammed. S. Al-KHESBAK(1), Mohammed. T. LAZIM(2)
Department of Electronic and Communications Engineering, Nahrain University, Iraq.
(2)
Department of Electrical Engineering, Philadelphia University, Jordan.
mohsaheb@yahoo.com , drmohamadtofik@yahoo.com
Abstract This paper proposes a new three-phase
envelope cycloconverter based on integral half
cycle selection with selected half cycle omission
(HCO) technique. The advantage of using this
technique is to eliminate completely the dominant
supply frequency harmonic component and other
undesirable harmonic components that may
present in the conventional frequency division
cycloconverter output voltage waveforms. Total
harmonic distortion is drastically reduced and the
possible risk of supply short-circuit is effectively
minimized. The output voltage and current of this
converter are smooth compared with corresponding
waveforms obtained by conventional converters.
Keyword:powerelectronics,a.c.converters,cycloconverters,frequency changers
1. INTRODUCTION
In conventional cycloconverters, the alternating
voltage at supply frequency is converted directly to a
lower frequency voltage without any intermediate
DC stage. Two types of cycloconverters are
commonly used in industry, namely, the phase
angle controlled and the envelope cycloconverters
[1]. The operating principles of these types of
frequency changers are well known and their circuits
are well developed in the last five decays due to the
invention of the thyristor and other power
semiconductor devices. The microprocessor and
microcomputer -based circuitry have led to a revival
of interest in the cycloconverter principle.
Sophisticated control circuits permit the conversion
of a fixed input frequency to a variable output
frequency at variable voltage, and such schemes are
attractive for AC motor drives [1,2].
Some new conversion methods from ac to ac have
also been reported such as matrix converter and
integrated PWM converter/inverter systems [3,4]
These methods have gained considerable
acceptance as schemes in the field of power
frequency changers and speed control of induction
motors..
The envelope cycloconverter proposed in this
paper, is based on half cycle selection technique
which is considered as a form of integral cycle
control [2]. The output voltage at the desired level
and frequency is produced by sequentially
triggering
chosen
bidirectional
MOSFET's
connected directly across the input a.c. voltages
without using input transformer.
The main circuit diagram of the proposed
single-phase to single-phase cycloconverter is
shown in Fig. 1. The two single-phase P and N
converters are operated as bridge rectifiers; their
delay angles are such that the output voltage of one
converter is equal and opposite to that of the other
converter. If converter P is operating alone, the
average output voltage is positive and if converter
iS
L
+
-
g1
S1(P)
iL
S3(N)
+
g3
+
g4
S4(N)
L
O
A
D
N
+
g2
S2(P)
Fig.1 Single-phase to single-phase circuit diagram
of the proposed cycloconverter.
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N is operating, the output voltage is negative. The
output waveform will result in (T) number of
positive supply half cycles driven by the P –
converter and the same number of the negative
supply half cycles driven by the N-converter to form
a periodic output sinusoidal load voltage with
frequency fo equal to fsupply / T as shown in Fig.2.
important advantages. First, is the effective total
harmonic distortion reduction, and the second is the
phase balancing improvement in a three-phase
cycloconverter. These two techniques will be
discussed in the following subsections.
2.1 Harmonic Reduction Using HCO Technique
Consider the T=3 cycloconverter output
waveform shown in Fig. 3, where the last half cycle
of
both the
vLj (ωt)
Load voltage
vL (ωt)
i =0
Negative group
Positive group
2V
i =1
2V
T=3
T=3
Angular frequency (ωt)
0
π
2π
3π
π
5π
6π
 j  2
j
T
N=1
N=2
T
 j  3
 j  4
 j  5
 j  6
T
T
T
T
ωt
 j 
N=T+1
T
N=T+2
Fig.2. Typical load voltage waveform for the cycloconverter
(T=3).
According to this principle of operation, the
cycloconverter can be viewed as a frequency divider
(down converter), that is, it is possible to divide the
supply frequency by any positive integer number
(T). Suitable switching timing of the two groups of
bidirectional MOSFET's, from the assigned control
circuits with pre-specified T number of cycles
(division factor) will result in a desired
cycloconverter output waveform with frequency fo.
2. HALF CYCLE OMISSION TECHNIQUE
When the output voltage of the envelope
cycloconverter with a waveform shown in Fig. 2 is
controlled by omitting appropriate half cycles
completely or partially from the positive as well as
the negative half cycles, the harmonic content of the
resulting waveform will be greatly affected. This
technique may be called half cycle omission (HCO).
As one interesting feature of HCO is to omit the
last half cycle of the positive half cycle bank as well
as the last one from the negative bank of half cycles
to avoid any possible short circuit between the Pand N-converters in conventional envelope
cycloconverter. This will lead to avoid using the
expensive intergroup reactors as well as solve the
circulating current problem.
The application of HCO technique to the
cycloconverter output waveforms will offer two very
Fig. 3 Typical load voltage waveform for single-phase
integral-cycle based cycloconverter with half cycle
omission (T=3).
positive and negative bank of half cycles are
omitted completely. The Fourier series coefficients
for this waveform will be derived to investigate the
effect of applying the HCO technique on the
harmonic content of the waveform. However, this
waveform can be expressed in terms of summation
of half cycles of the original sine waveform as
follows;
For the jth phase of a multi-phase system, the
output voltage waveform can be expressed as:i 1 N ((1i )T ) 1
v L j ( t )  
i 0

 j  N
(1)
N 1i
N 1iT
2V sin(T t   j )
T
 j  ( N 1)
(1)
T
and the Fourier coefficients are
1  j  2
a0 j  
v L (t ) dt  0

(2)
j
For n  T
(3)
1  j  2
an j  
v L (t ) cos n t dt
 j

2V T i 1 N ((1i )T )1 2 N  2i  n
n

(1)
 cos ( j  N )  cos ( j  ( N  1) ) 

2
2 
T
 (T  n ) i 0 N 1iT
 T

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bn j 
1  j  2
v L (t ) sin n t dt
  j
Cn
Cn
0.9
0.8
0.8

2V T i 1 N ((1i )T ) 1 2 N  2 i  n
n

  (1)  sin T ( j  N )  sin T ( j  ( N  1) ) 
 (T 2  n 2 ) i 0 N 1iT
0.7
0.7
0.6
0.6
0.5
THD%=69.8%
T=4
0.5
THD%=55.3%
T=4
0.4
0.4
0.3
(4)
and the Fourier coefficients of the supply frequency
component, where n  T are
aT j 
2V
4 T
i 1 N ((1i )T )1
  (1)
i 0
N 1i
 2 sin  
(5)
j
N 1iT
0.3
0.2
0.2
0.1
0.1
0
012.5
50
100
150
200
250
0
350012.5
300
Frequency (Hz)
(a)
50
100
150
200
Cn
250
300
350
Frequency (Hz)
(b)
Cn
0.9
0.8
THD%=70.5%
T=5
0.8
THD%=56.9%
T=5
0.7
0.7
0.6
50 Hz component to be
eliminated by HCO.
0.6
2V
bT j 
4 T
i 1 N ((1i )T ) 1
  (1)  2 cos  
i 0
N 1iT
N 1i
j
0.5
(6)
0.5
0.4
0.4
0.3
0.3
0.2
0.2
The amplitude Cn of the nth harmonic component
and the angle ψn for the jth phase can be found as;
cn j  an2 j  bn2 j
(7)
and,
 n j  tan 1
an j
0.1
0.1
0
0
0 10
50
100
150
200
250
Frequency (Hz)
(c)
0 10
50
100
(d)
150
200
250
Frequency (Hz)
Fig. 4 Frequency spectrum of cycloconverter output
waveforms (a) (50 to 12.5)Hz (b) (50 to12.5)Hz with
HCO, (c) (50 to10) Hz, and (d) (50 to10) Hz with HCO.
(8)
bn j
for n=1,2,3, ………
Using eqs. (2)-(6) for the HCO waveforms and eq.
(7) describing Cn , the effect of half cycle omission
on the frequency spectrum can be obtained and
compared to the spectrum of the cycloconverter
output waveforms without HCO. Fig.4 shows two
converted frequencies with and without HCO
application for j=1.
It is clear from Fig. 4 that the major effect of half
cycle omission is to reduce the amplitudes of the
dominant harmonic components. In addition, it
results in cancellation of some undesirable harmonic
frequency components that leads to reduce THD.
Moreover, the application of HCO will result in
reduced r.m.s. value of the output voltage as it is
clear in each case.
Without using HCO, the supply frequency
component is generated with significant amplitude
for odd values of (T). One of the most important
advantage resulted from using symmetrical half
cycle omission is the complete elimination of the
supply frequency component.
1
2T
VLrms 
2 (T 1)
 

(9)
2
2V sin( t ) d t
0
with further simplification, the resultant load r.m.s.
voltage is found to be
VLrms  1 
1
V
T
(10)
From eq. (10), it is clear that VLrms is a function of
T. The reduction in harmonic content due to HCO
can be seen clearly by comparing the THD% before
and after application of HCO curves given in Fig. 5.
72
THD%
70
68
conventional
cycloconverter
66
64
62
60
cycloconverter with
HCO
58
56
54
2
3
4
5
6
7
8
9
10
11
T
12
Fig. 5 Effect of varying cycloconverter number of cycles
T on the THD% of outputs with and without the
application of HCO.
The r.m.s. voltage value of the half cycle omitted
cycloconverter output waveform of T cycles can be
simply derived using eq. (1). From which the load
r.m.s. output voltage is
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2.2 Three-Phase Balancing Improvement Using
HCO
In the three-phase systems, it is not adequately
sufficient to generate a desired output waveform
with desired output frequency, but also to ensure that
the three generated phases have balanced phase
displacement relations of the operating frequency
component.Unfortunately, in all
types
of
conventional envelope three-phase cycloconverters
the fundamental desired frequency as well as the
harmonic components is generated with random
unbalanced phase relationships with phase angle
displacement depends on the values of T. This fact is
also appears in conventional 3-phase integral cycle –
based frequency changers, although little attempts
had been made to improve the phase relationships.
[ 5,6,7 ].However, using the proposed HCO
technique, phase balancing improvement for the
fundamental and other certain harmonic components
could be achieved.
If we consider the T=3 (16.6 Hz) cycloconverter
output waveform shown in Fig. 6, where the
application of HCO is experienced through the
testing of the three possible waveforms resulted from
omitting the first, second, or the third half cycle. The
original output waveform as well as its phase
displacement angle is shown in Fig. 6(a) with the
resulted THD%. Now, if a comparison is made
between waveforms depicted in Figs. 6 (b), (c), and
(d) with that for the original waveform in Fig. 6 (a),
it is clear that HCO can be used to change the phase
angle displacement of the fundamental harmonic
component by a phase angle Φo with reduced THD.
For any output waveform defined by the value of
(T), the modification phase angle is found to be
o 

(radian)
If HCO is applied such that vA is shifted by +30o
(No=T=3) and vC by -30o (No=1), the resultant phase
relation for the fundamental component will still
90
400
1
120
60
0.8
THD%=67.9%
300
0.6
150
200
30
0.4
100
0.2
0
180
0
-100
-200
210
330
-300
-400
240
0
0.01
0.02
0.03
0.04
0.05
0.06
300
270
(a)
90
0.8
120
400
60
0.6
THD%=54.7%
300
0.4
150
30
200
0.2
100
180
0
0
-100
210
-200
330
-300
240
-400
0
0.01
0.02
0.03
0.04
0.05
300
270
0.06
(b)
90
400
0.5
120
60
0.4
THD%=170.2%
300
0.3
150
200
30
0.2
0.1
100
180
0
0
-100
-200
210
330
-300
-400
240
0
0.01
0.02
0.03
0.04
0.05
0.06
300
270
(c)
90
400
0.8
120
THD%=54.7%
300
60
0.6
0.4
150
200
30
0.2
100
0
180
0
-100
-200
210
330
-300
240
-400
0
0.01
0.02
0.03
0.04
0.05
0.06
300
270
(d)
Fig. 6 Single phase waveform and fundamental component
(16.6Hz) phase angle displacement for T=3 cycloconverter (a)
without HCO (b) 1st cycle omission No=1 (c) 2nd cycle
omission No=2 (d) last (3rd) cycle omission No=T=3.
(11)
2T
The location of the omitted half cycle will be
defined by No, where No will range from 1 (first half
cycle) to T (last half cycle) as shown in Fig. 6. In
other words, No must take two convenient values (1
and T) only for obtaining both reduced THD% and
phase displacement modification.
In order to clarify the application of HCO for
three-phase cycloconverter, Fig. 7 (a) depicts the
phase relations for the first four harmonic
components for T=3 (16.6 Hz) output waveform,
which is to be improved for the 16.6 Hz fundamental
frequency component.
have significant unbalanced displacement angles
between the three phases. A more acceptable phase
relation improvement can be obtained by shifting vA
by +30o and vB with -30o, while inverting and then
shifting vC by +30o as shown in Fig. 7(b). It is also
possible to reverse the direction of rotation for the
fundamental component by simply reversing vB and
shifting vC by -30o. The phase improvement offered
by half cycle omission shown for the T=3
cycloconverter output, can also be applied for all
values of T.
Another advantage of using HCO for phase
balancing is that it eliminates completely the
harmful tripplen harmonics, which are usually
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associated with waveforms generated by the
conventional envelope cycloconverters for odd
values of T. This can be clearly shown from Fig.
7(b).
120
90 1
60
0.5
150
150
30
120
90 0.4
60
vC
0.2
30
vA
180
vA
0
210
vB
vC
240
330
210
240
270
150
90 0.5
180
vC
210
240
150
120
vC
150
0
180
vC
210
330
330
vB
240
270
0
240
1st (16.6Hz)
120
vC
150
180
vA
210
240
270
120
30
180
330
vC
vB
240
(b)
60
0.1
210
300
5th (83.3Hz)
90 0.2
150
0
vB
270
300
3rd (50Hz)
60
0.2
330
210
300
90 0.4
30
180
0
Now the load voltage expression of the bold line
waveform is
i 1
v L j ( t )   (1) N 1i 2V sin(T t   j )
 j  ( N 1) 
T
 j  ( N 1)
(12)
T
where N=(1+i)T, and the Fourier coefficients are
90 4e-016
60
120
Fig. 8 shows the output waveform of partial half
cycle omission (PHCO) with ψ as phase control
angle, that ranges from 0 (complete HCO) to 180o
(no HCO). The bold line shown in this waveform
represents only the partially introduced part of the
previously omitted half cycle. The Fourier
coefficients of the dashed line for HCO waveform
shown in Fig. 3 are given in eqs. (2) to (6), and the
overall Fourier series coefficients will be the
summation of that of the bold line in Fig. 7 with
that previously obtained for the dashed line (Fig. 3).
i 0
150
30
vA
210
270
300
7th (116.6Hz)
60
180
0
330
vB
240
(a)
0.5
30
vA
5th (83.3Hz)
90 1
60
0.1
300
270
90 0.2
120
30
vA
270
300
3rd (50Hz)
60
0.25
vB
330
vB
300
1st (16.6Hz)
120
0
180
between 0 and 90o/T by means of phase angle
control of the half cycle to be partially omitted.
30
0
a0 j 
 j  2
1
 
v Lj (t ) dt  0
(13)
j
For n  T
an j 
1 
 
j
 2
v Lj (t ) cos n t dt
j
n
n


  cos( j   ) cos((  1) j  (( N  1)   )) 
T
T


 n

n
n
 sin(  j   ) sin((  1) j  (( N  1)   )) 
i 1

2V T
T
T
T


(1) 2 N i 
2
2 
 (T  n ) i 0
  cos( ) cos(( n  1)  n (( N  1) ))

j
j


T
T


  n sin( ) sin(( n  1)  n (( N  1) ))

j
j
T
T
 T

(14)
330
vA
270
300
bn j 
7th (116.6Hz)
Fig.7 Harmonic phase relationships for T=3
cycloconverter output showing the 1st, 3rd, 5th, and
7th harmonic component for
(a) ordinary
cycloconverter (b) Improved phase displacement by
HCO applied to vA with No=T and vB with No=1, while
vC is inverted and then applied with HCO of No=T.
1
 j  2
 
vLj (t ) sin n t dt
j
n


  cos( ) sin ( j  ( N  1)   ) 
T



2V T i 1
n
n
2 N i 

(

1
)

sin(

)
cos
(


(
N

1
)



)



j
 (T 2  n 2 ) i  0
T
T


n



sin
(


(
N

1
)

)
j


T


(15)
2.3 Partial HCO Technique
Half cycle omission may not only be introduced
by omitting complete half cycles from both the
positive and negative bank of half cycles, but partial
half cycle omission is also possible. Partial half cycle
omission (PHCO) will offer a fractional modification
of the phase displacement relation of a certain
frequency component by offering phase deviation
and the supply frequency component, where
n  T is
i 1
aT j 
2V
4 T
 (1)
bT j 
2V
4 T
 (1)
N 1i
i 0
i 1
i 0
N 1 i
cos 
j
 sin 
j
 cos( j  2 )  2 sin  j 
(16)
 sin( j  2 )  2 cos  j 
(17)
The frequency spectrum and the phase angle
displacement relations, which are investigated by
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Journal of Electrical Engineering
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means of Fourier series coefficients for PHCO, show
that, in addition to the fractional change in phase
displacement from 0 to 90o/T, obtained by varying ψ,
a further reduction in THD% is also achieved. The
effect of varying ψ on both, the THD% and the
phase displacement obtained by PHCO, can be
shown in Fig. 9.
vLj (ωt)
i =0
i =1
2V
T=3
N=1
N=T
N=2

 j 
j
 j  ( N  1)
T
T
T
 j  ( N  1)
T
 j  4
 j  3
T
T
characteristics.
3.1 Resistive load: Harmonic reduction and
phase balancing effect of using HCO
technique is tested with three-phase resistive
load for 1:2 (25 Hz) output cycloconverter
voltage. Phase voltages and neutral current for
conventional envelope cycloconverter output
are monitored and shown in Fig. 10. The
neutral current is found to have a d.c.
component. This is due to the unbalance phase
relations of the generated voltage harmonic
components.
 j  ( N  1) 
T
 j  6
N=T+1 N=T+2
T
 j  ( N  1) 
T
ω
t
N=2T
vOA
120o
Fig. 8 Typical load voltage waveform for singlephase integral-cycle based cycloconverter with
partial half cycle omission (T=3).
T=2
T=3
T=4
T=5
T=6
++++++
70
vOB
(a)
vOA
THD%
75
Scale
V : 200V/Div.
Herz. : 10 msec/Div.
240
(25 Hz)
(16.6 Hz)
(12.5 Hz)
(10 Hz)
(8.3 Hz)
o
Scale
V : 200V/Div.
Herz. : 10 msec/Div.
vOC
(b)
65
iN
Scale
I : 1A/Div.
Herz. : 10 msec/Div.
60
55
50
ψ
0
20
40
60
80
100
120
140
160
180
(a)
Phase change (degree)
50
++++++
40
T=2
T=3
T=4
T=5
T=6
(25 Hz)
(16.6 Hz)
(12.5 Hz)
(10 Hz)
(8.3 Hz)
30
10
HCO
(Ψ=0o)
0
PHCO
(0o <Ψ<180o)
)
)
-10
0
20
40
60
80
100
120
140
160
(b)
180
ψ
Fig. 9 Effect of PHCO control phase angle ψ, for
different output frequencies, on (a) THD%, and (b)
resultant phase change.
3.
EXPERIMENTAL
RESULTS
(c)
Fig.10. Voltage and current oscillograms of a 1:2 (25 Hz)
frequency division cycloconverter 3-phase waveforms (a)
vOA and vOB (b) vOA and vOC (c) Neutral Current .
conventional
cycloconverter
(Ψ=180o)
20
Approx. DC
level
WORK
AND
The proposed cycloconverter based on
half cycle selection and HCO techniques is
built and tested in the laboratory with various
types of loads. The results confirm both the
principles and the theoretically predicted
When the HCO technique is used to make the phase
displacement balanced, the neutral current is found
to have less estimated d.c. component as shown in
Fig.11 (c) and it has more smooth waveform.
3.2 Performance with 3-Phase AC Motor Load:
The converter was tested with a 3-phase, 500W,
220 V, 50 Hz, Y-connected, 2-pole induction motor.
Each generated waveform is evaluated according to
the shaft speed that the motor will run at, and how it
is close to the corresponding frequency designed to
cause such a rotating speed. The synchronous speed
ns of the induction motor with ignored slip factor is
given by the following well known fundamental
equation : nS=120* f / p , where p = number of
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Journal of Electrical Engineering
www.jee.ro
poles and f = supply frequency. This indicates that ns
can be varied by changing the frequency f.
vOA
vOB
Scale
V : 200V/Div.
Herz. : 10
msec/Div.
theoretically in Fig. 14(e). However, this will have
no significant effect on the motor performance.
corresponds to 8.25 Hz component, which is not
even appears in the frequency spectrum as shown in
Fig. 12(d). However, unsmooth and unstable
rotation of the motor is observed in this case.
vOA
vOA
(a)
vOA
vOC
Scale
V : 200V/Div.
Herz. : 10 msec/Div.
120o
Scale
V : 200V / Div.
Horz. : 10 msec / Div.
Scale
V : 200V / Div.
Horz. : 10 msec / Div.
240o
vOC
vOB
(b
)
(a)
iN
Scale
I : 1A/Div.
Herz. : 10 msec/Div.
(b)
Scale
I : 5A/ Div.
Horz. : 10 msec / Div.
90 1
120
150
30
0.5
150
120
60
vOA
iS
(c)
180
0
180
vOB
Fig. 11 1:2 (25 Hz) frequency division cycloconverter 3phase waveforms phase balanced and quality
improvement using HCO (a) vOA and vOB (b) vOA and vOC
(c) Neutral Current .
The motor is tested first with the conventional
envelope cycloconverter generating 3-phase 25 Hz
output voltage. Oscillograms for the stator voltage
and currents are shown in Fig.12 (a) and (b)
respectively. Frequency spectrum and three-phase
displacement angles of the fundamental 25 Hz
component are also shown in Figs. 12(c) and (d)
respectively. The motor is supposed to run at half of
the synchronous speed (1468 r.p.m.) corresponding
to 25 Hz component. The measured speed is found to
be 485 r.p.m. and according to eq. (18) this speed
corresponds to 8.25 Hz component, which is not
even appears in the frequency spectrum as shown in
Fig. 12(d). However, unsmooth and unstable rotation
of the motor is observed in this case.
Now, when HCO technique is applied to obtain a
phase balance to the three-phase output voltage
waveforms by omitting the last half cycle of phase
A, first half cycle of phase C, and leaving phase B
unchanged as shown by Fig. 13. Phase balancing
improvement achieved results in motor shaft speed
of 1467 r.p.m. corresponding to 24.97 Hz with
smooth rotation. The application of HCO results in
slight voltage amplitude unbalance as shown
210
300
240
25 Hz
(d)
(c)
Cn
(normalized)
270
210
330
vOC
240
120
0.9
150
0.8
90 0.2
120
60
0.1
150
30
0.7
0.6
0
180
THD% = 62.3%
T=2
210
0.5
0.4
180
330
210
0.3
240
0.2
0
270
300
240
Frequency (Hz)
0.1
0 25 50
100
150
200
300
400
500
600
(e)
Fig.12 25 Hz unbalanced 3-phase waveform generation
causing motor to run at speed of nr=485 r.p.m. (a) vOA
and vOB (b) vOA and vOC (c) stator current (d) 25 Hz
component phase relations (e) theoretical phase voltage
frequency spectrum.
However, when HCO technique is applied to
obtain a phase balance to the three-phase output
voltage waveforms by omitting the last half cycle of
phase A, first half cycle of phase C, and leaving
phase B unchanged as shown by Fig. 13. Phase
balancing improvement achieved results in motor
shaft speed of 1467 r.p.m. corresponding to 24.97
Hz with smooth rotation. The application of HCO
results in slight voltage amplitude unbalance as
shown theoretically in Fig. 13(e). However, this
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Journal of Electrical Engineering
www.jee.ro
will have no significant effect on the motor
performance.
vOA
vOA
120o
240o
Scale
V : 200V / Div.
Horz. : 10 msec / Div.
Scale
V : 200V / Div.
Horz. : 10 msec / Div.
vOC
vOB
(a)
iSB
voltage and current waveforms are smooth and
largely free from spikes and undesirable
discontinuities. As an a.c. drive, this cycloconverter
has many advantages over PWM inverter drives
since it imposes no problems on the a.c. machine
windings due to the high frequency switching
associated with PWM techniques [8]. Moreover, it
is almost free of switching losses; therefore, its
conversion efficiency is relatively high.
REFERENCES
(b)
Scale
I : 1A/ Div.
Horz. : 10 msec / Div.
Scale
I : 2A/ Div.
Horz. : 10 msec / Div.
[1] Gyugi, L. and Pelly, B.R.: Static Power
Frequency Changers. John Wiley and Sons,1976,
New York, USA.
[2] Lazim , M.T. and Shepherd ,W.: A General
Modulation Theory for Single-Phase Thyristor
Systems. In: Journal of Franklin Inst. Vol. 312, No.
6, pp. 399-415, December 1981, USA.
iSA and iSC
(c)
90
(d)
[3]Bayoumi,E.H.E., Maamoun,A., Pyrhönen,O.,
Khalil,
M.O., and Mhfouz, A. : Enhanced
30
vOA
Method
for Controlling PWM Converter180 v
180
0
0
OC
Inverter
System,
In: Proc.of the IASTED
vOB 330
330
210
210
International Conf. of Power and Energy
300
240
300
240
270
270
Systems,
PES\'02,
California,
USA,
(e)
May2002,pp.425-430.
90 0.2
90 0.1
Fig. 13 25 Hz balanced
3-phase
waveform generation
120
60
120
60
[4] Bayoumi, E.H.E.: Analysis and Design of
using HCO technique
causing
motor
to
run
at
speed
of
0.1
0.05 30
150
150
30
Linear and Variable Structure Control
nr=1467 r.p.m. (a) vOA and vOB (b) vOA and vOC (c) phase B
Techniques
for
PWM
Rectifier,
In:
0
180(e) 25 Hz
stator Current180
(d) phase A and C 0stator Current
Electromotion
Scientific
Journal,
Vol.
11,
No.
4,
component phase relations
330
330
210
210
pp. 205-212, Oct.-Dec., 2004.
300
240
240
300
270
270
CONCLUSION
[5] AlKhasbak, M.S.M : Integral-Cycle Based
Theoretical and practical behavior characteristics of
Cycloconverter with Voltage Control Using
a proposed three-phase variable frequency
PWM Technique. Ph.D Theses, Nahrain
cycloconverter have been presented. This
University – Iraq, 2007.
120
150
1
60
0.5
120
30
90 1
150
cycloconverter is based on half cycle selection and
half cycle omission (HCO) technique. It is found that
using HCO technique solves the inherent problems
associated
with
conventional
envelope
cycloconverters of similar type. With HCO
significant reductions in harmonics, elimination of
the supply frequency component, and balancing the
output voltage waveforms in the three phases are
easily achieved. Experimental results found to agree
well
with
the
theoretical
investigations.
The suggested frequency changer proves to have
many advantages over conventional types , such as
inverters and cycloconverters , in that it is more
simple, requires no complex triggering circuits due
to the few MOSFETs used , need no input
transformers and need no filter because the output
60
0.5
[6] Yong Nong Chang, Gerald Thomas Heydt ,and
Yazhou Liu.: The impact of switching strategies on
power quality for integral cycle controllers.In:
IEEE Trans. Power Delivery, 18 (3): 10731078,2003.
[7] Mahmood, A. L. , Lazim, M. T. ,and Badran, I.:
Harmonics phase shifter for a three- phase system
with voltage control by integral-cycle triggering
mode of thyristors.In: American Journal of
Applied sciences,Vol.5, No. 11, 2008, USA.
[8] Chambers, D., et all: Eliminate the adverse
effects of PWM operation [in AC motor drives] .
In: Industry Applications Conference. Thirty
Sixth IAS Annual Meeting. Conference Record of
the IEEE Vol. 3 pp. 2033 - 2040 ,2001 .
8