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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 5, SEPTEMBER/OCTOBER 1997
Analysis and Experimental Evaluation of
Single-Switch Fast-Response Switching
Regulators with Unity Power Factor
K. W. Siu, Y. S. Lee, and C. K. Tse, Member, IEEE
Abstract— Based on a single-stage isolated power-factor-corrected power supply (SSIPP or S2 IP2 ) employing a boost powerfactor-correction (PFC) cell, the relationship between the input
line voltage and line current of such switching regulators is
studied. The conditions for unity power factor are derived. A
control scheme to reduce the switch voltage stress while maintaining unity power factor is proposed. Practical implementation
of the proposed control scheme is discussed. It is found that,
by adding a simple input voltage feedforward to a conventional
PWM controller, the switch voltage stress can be much reduced,
while keeping the power factor very close to unity. Analysis and
experimental results indicate that such a scheme is effective and
feasible.
Index Terms— Fast output-voltage regulation, power-factor
correction, SSIPP.
I. INTRODUCTION
I
NTERNATIONAL regulatory standards such as IEC 10003-2 [1] impose restrictions on the magnitudes of the harmonic components of the line current of switch-mode power
supplies. Although this problem can be solved by adding
a power-factor-correction (PFC) preregulator to the power
supply circuit, such a design approach is far from optimum
in terms of the size and cost of the power supply. Various
alternative schemes have been proposed to optimize the design
[3]–[12]. However, these schemes suffer from one or more of
disadvantages such as slow output-voltage regulation [3], [4],
complex circuitry [5]–[12], and low conversion efficiency [8].
In order to overcome these disadvantages Redl, Balogh, and
Sokal proposed a new family of single-stage isolated powerfactor-corrected power supplies (SSIPP or S IP ) [2]. In this
family of power supplies, six S IP topologies are based on the
boost PFC cell. However, these boost-based S IP still suffer
from the drawback that the line currents drawn are inherently
distorted.
Paper IPCSD 97–42, presented at the 1996 IEEE Applied Power Electronics
Conference and Exposition, San Jose, CA, March 3–7, and approved for
publication in the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS by the
Industrial Power Converter Committee of the IEEE Industry Applications
Society. This work was supported in part by the Research Grants Council
of the University Grants Committee of Hong Kong, the Research Committee
of The Hong Kong Polytechnic University, and the Industry Department of
the Hong Kong Government. Manuscript released for publication May 16,
1997.
The authors are with the Department of Electronic Engineering, The Hong
Kong Polytechnic University, Kowloon, Hong Kong.
Publisher Item Identifier S 0093-9994(97)06555-9.
In this paper, a brief review on the principle of singlestage power-factor-corrected switching regulators is first given.
Then, the factors affecting the shape of the input line current of
such regulators will be studied. Based on these findings, a new
control scheme to achieve a power factor up to unity will be
proposed. A practical version of the control scheme, using only
four additional passive components, will be described. The
circuit represents a truly simple and low-cost implementation
of near-unity power factor S IP . Steady-state analysis based
on the new control schemes will be given. Experimental results
will also be presented.
II. PRINCIPLE OF SINGLE-STAGE
POWER-FACTOR-CORRECTED SWITCHING REGULATORS
Based on an S IP using a boost PFC cell [2], the principle
of operation of single-stage power-factor-corrected switching
regulators will be discussed in this section.
Fig. 1 shows the simplified schematic of an S IP . This
S IP can be understood as a cascaded connection of a boost
converter and a forward converter, as identified by the dashedline boxes. The two converters (called cells) share the same
electronic switch
. Nodes
and
serve as the output
terminals of the boost cell and, at the same time, the input
terminals of the forward cell. The presence of diode switch
prevents the primary current of transformer
from
circulating through
.
During normal operation, the boost cell is operated in
discontinuous mode and the forward cell in either continuous
or discontinuous mode. In order to achieve a power factor of
1, the waveform of the averaged input current in Fig. 1,
(averaged over a switching period of
), should resemble
that of the rectified sinewave of the ac mains. With this
assumption, the averaged output current of the boost cell
should also have the same rectified sinusoidal waveform. This
implies that, inherently, the output voltage of the boost cell
must contain a ripple component, the amplitude of which
would depend on the value of the capacitance
and the
loading current. Usually, this ripple can be maintained within
a few percent by using a sufficiently large
. The output
voltage
is regulated via a feedback loop which dynamically
adjusts the duty cycle of the electronic switch
to keep
fixed.
It should be obvious that, since the duty cycle of the
electronic switch
is used to regulate the output voltage
, it cannot be used at the same time to shape the input
0093–9994/97$10.00  1997 IEEE
SIU et al.: SINGLE-SWITCH FAST-RESPONSE SWITCHING REGULATORS
1261
Fig. 1. An S2 IP2 using a boost PFC cell.
Fig. 2. Low-frequency behavior model of S2 IP2 assuming a continuous-mode-operated forward cell.
current
of the boost cell (so as to make it look like the
rectified sinewave). In the topologies proposed in [2], the
design makes use of the natural shape of the boost-cell input
current (under discontinuous mode operation) to achieve a
nonideal power-factor correction.
It should be noted that an S IP using a buck-boost input
cell operating in discontinuous mode will theoretically have
a unity power factor. However, it is not the preferred input
cell, because a buck-boost converter has a poorer conversion
efficiency compared with a boost converter.
III. CONTROL SCHEME TO ACHIEVE UNITY POWER FACTOR
Based on well-established low-frequency behavior models
of boost and forward converters [14]–[16], the relationship
between the line voltage and the input current of an S IP
with a boost PFC cell will be studied in this section. The
conditions for unity power factor will then be derived.
Fig. 2 shows the low-frequency behavior model of the S IP
given in Fig. 1. For the sake of analytic simplicity, it is here
assumed that the boost cell always operates in discontinuous
mode and the forward cell always in continuous mode. The
meanings of the symbols used in Fig. 2 are as follows:
switching period of
;
duty cycle of
;
angular frequency of ac line input;
(1)
(2)
, in fact, is the duty cycle of the diode switch
where
From Fig. 2, the input current is found to be
.
or
(3)
In the actual operation of the S IP , the following assumptions
are valid.
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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 5, SEPTEMBER/OCTOBER 1997
1) The ripple of
is small, because
is large.
2) The duty cycle
of the electronic switch
(of
the continuous-mode-operated forward cell) is nearly
constant within each half cycle of the ac mains.
With these assumptions, it can be found that if the term
in (3) is made small, the power factor of the S IP will
be close to 1 (because
is then proportional to ). This
requirement is, however, difficult to meet, because it implies
a very large , which will place a very large voltage stress
on the electronic switch
. The large voltage stress is often
a limiting factor that prevents the SSIPP from being used in
high-voltage circuits.
As an alternative solution to the problem, we propose here
that the parameter
(the switching period of the electronic
switch) be modulated according to
(a)
(4)
is the switching period when
. If this
where
could be done, the input current
in (3) would be directly
proportional to and the power factor would become 1. This
new control scheme makes the design of single-switch fastresponse switching regulators with unity power factor possible
without the need for a very large .
IV. PRACTICAL IMPLEMENTATION OF CONTROL SCHEME
An exact implementation of (4) in a pulse width modulation
(PWM) IC will require a complex control circuit in the
oscillator. However, it will be shown that, in practice, a much
simpler control circuit can be used to keep
low, while still
meeting the IEC 1000-3-2 requirements.
Consider the typical sawtooth oscillator circuit (for a
UC3844 PWM IC) shown in Fig. 3(a) as an example, where
the oscillating period
is given by
(5)
(b)
Fig. 3. Sawtooth generator of UC3844 PWM IC. (a) Normal fixed-frequency
circuit. (b) Proposed frequency-modulation circuit.
We then have, from (8) and (9),
In order to implement the control law of (4), ideally, we need
to have, from (4) and (5)
(10)
The implementation of even (10) is not straightforward,
because it involves the division of by . However, if
is replaced by a constant , so that (10) becomes
or
(6)
However, knowing that
(6) as a series:
is less than 1, we can express
(7)
(11)
the implementation will be much easier. A simple circuit that
can implement the function of (11) is shown in Fig. 3(b),
where
(12)
may then be approximated by (neglecting the
The required
second and higher order terms)
(8)
Assume that
is chosen to have a dc bias of
assuming
, so that
(9)
(13)
The corresponding switching period
then given by
of the converter is
(14)
SIU et al.: SINGLE-SWITCH FAST-RESPONSE SWITCHING REGULATORS
1263
For a given value of , the range of frequency variation
implied is between
and
.
Although in the derivation of (11) the parameter
is
supposed to replace
, it does not mean that must have
.
a value equal to, or even close to, the averaged value of
In order to evaluate the performance of the proposed control
scheme, a steady-state analysis will be performed in Section
V to find out the relationships among voltage stress, power
factor, and harmonic distortion for different values of .
V. ANALYSIS
The steady-state analysis will be carried out based on the
circuit model of Fig. 2. One of the important parameters of
the circuit is the voltage stress
across the storage capacitor.
This voltage can be determined by equating the average output
current of the boost cell during a line half cycle to the average
input current of the forward cell during the same half cycle.
Referring to Fig. 2, the equality can be written as follows:
or
(15)
Fig. 4. Voltage ratio V^i =Vc as a function of
.
undesirable continuous-mode operation of the boost cell. To
ensure the boost cell is operating in the discontinuous mode,
we must have
or
(19)
or
Substituting (14) and the circuit parameters, the following
equation for the voltage
is obtained:
or
(16)
Equation (19) imposes an upper limit for the voltage ratio
. To maintain a high voltage ratio, the value of should
be chosen such that the voltage ratio is close to that limit at
full-load condition.
Now, we calculate the power factor of the regulator. The
instantaneous input current
is given by
where
(20)
(17)
The rms value of the input current is
Fig. 4 is a set of curves showing the relationship between
and the voltage ratio
, for different values of . (Note
ratio indicates a lower voltage stress.) It
that a higher
can be seen that, for a given value of , the proposed control
scheme
gives a higher
ratio (and, therefore,
a lower ).
It should be understood that the graphs shown in Fig. 4
are based on the assumption of a continuous-mode-operated
forward cell. If the load resistance
is sufficiently large (or
sufficiently small), the forward cell will eventually enter into
discontinuous mode operation, and the voltage ratio
will
then stop to decrease [2]. The condition for the forward cell
to operate in continuous mode is given by
(21)
The power input is
(22)
Hence, the power factor of the system is
(18)
.
where
Also, it is obvious from Fig. 4 that a larger inductance
value of
will give a higher (and, therefore, better)
voltage ratio. However, too large a value of
may result in
(23)
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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 5, SEPTEMBER/OCTOBER 1997
Fig. 7. Measured storage capacitor voltage Vc versus output power.
the peak of the rectangular current through the primary side
of transformer
(ignoring magnetizing current):
Fig. 5. Power factor as a function of voltage ratio V^i =Vc .
(25)
The voltage stress of the electronic switch
is given by
(26)
VI. EXPERIMENTAL RESULTS
An experimental 100-W (18-V/5.6-A) converter has been
built and tested to verify the proposed control scheme. The
output voltage of the forward cell is regulated using voltagemode control with a UC3844 IC. The control network is
optimized for the forward section to operate in both continuous
mode and discontinuous mode.
The following is a list of circuit parameters used in the
experiment:
Fig. 6. THD as a function of voltage ratio V^i =Vc .
Fig. 5 shows a plot of the power factor versus the
ratio for different values of . It is interesting to see that,
while the proposed control scheme improves the power factor
for large values of
(when the voltage stress
is low),
it also results in a slightly worse power factor (compared with
the case of
) for small values of
. This, in fact, is
due to the approximated control scheme of (11). If the original
ideal control scheme of (4) was used, the power factor would
always be equal to 1. However, this worsening effect may
not appear in the actual operation of the converter, because
the ratio
can be kept large by properly selecting the
operating range of .
The total harmonic distortion (THD) of the line current can
be found as
(24)
ratio. It is
Fig. 6 shows the THD as a function of the
, the proposed control scheme keeps
found that with
the total harmonic distortion down to about 10% for a
ratio up to 0.75
.
The current stress of the electronic switch
is equal to
the sum of the peak of the sawtooth current through
and
s
H
F
H
F
With this set of circuit parameters, from (17) and (19),
and achievable voltage
the calculated maximum value of
ratio
are 1.3 and 0.81, respectively. From the set of
curves shown in Fig. 4, the optimum value for the parameter
is found to be 2. Fig. 7 shows the experimentally measured
versus power-level characteristics for different values of
. It is found that a reduction of maximum voltage stress
from 330 V (
constant) to 250 V
is obtained with the proposed control law. Fig. 8 shows the
averaged input line voltage and current waveforms when
the converter is operating at full-load condition. This is the
worst case operating condition with respect to line-current
distortion. When the proposed control law of (10) is applied
(with
), the THD of the line current is improved from
25%
to 15%
and the power factor is
improved from 0.97
to 0.99
. Fig. 9 shows
SIU et al.: SINGLE-SWITCH FAST-RESPONSE SWITCHING REGULATORS
1265
(a)
Fig. 11. Transient response of the output voltage to step changes in the
loading current from 100% to 50% and back.
(b)
Fig. 8. Line voltage and current waveforms. (a) When T is kept constant
0.97, THD
25%). (b) When proposed control law is applied with
(p.f.
2 (p.f. = 0.99, THD = 15%).
=
K=
=
100 W), the efficiency is 80.8%. This efficiency is rather low,
because an SSIPP is effectively a cascaded connection of two
converters. However, the measured harmonic components of
the input current comply well with the IEC 1000-3-2 Class D
requirement, as shown in Fig. 10.
Fig. 11 shows the transient response of the output voltage
(middle trace) of the circuit due to step changes in the
load current from 100% to 50% and back over a half line
cycle (bottom trace). The response is very fast because it
is determined solely by the forward cell with its wide-band
voltage-regulating loop. It should be noted that the output
voltage ripple at peak ac line voltage is smaller than at zero
ac line voltage because of the frequency-variation nature of
the control scheme.
VII. CONCLUSION
Fig. 9. Converter efficiency versus output power.
The factors affecting the power factor of single-switch fastresponse switching regulators using a boost input cell have
been studied. A control scheme to reduce switch voltage stress
and to improve the power factor has been proposed. A simple
and low-cost implementation of the proposed scheme, using
only four passive components, has also been discussed. Experimental results have confirmed that single-switch regulators
employing the new control scheme can be designed to have a
relatively low voltage stress, while maintaining a high power
factor.
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for Harmonic Current Emissions (Equipment Input Current 16A Per
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voltage,” in Conf. Rec. IEEE PESC’94, Taipei, Taiwan, R.O.C., 1994,
pp. 1137–1144.
[3] E. X. Yang et al., “Isolated boost circuit for power-factor correction,”
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<
Fig. 10.
Harmonic contents of input current.
the converter efficiency at different power levels with
.
Under the condition of 110-V ac input and 18-V dc output (at
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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 5, SEPTEMBER/OCTOBER 1997
[6] H. Watanabe, Y. Kobayashi, Y. Sekine, M. Morikawa, and T. Ishii, “The
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K. W. Siu received the B.Eng.(Hons.) degree in
electronic engineering in 1992 from The Hong Kong
Polytechnic University, Kowloon, Hong Kong,
where he is currently working toward the Ph.D.
degree in power electronics.
His research interests include power-factor
correction circuits for ac–dc converters and
computer-aided design of switching power supplies.
Y. S. Lee received the M.Sc. degree from the
University of Southampton, Southampton, U.K., and
the Ph.D. degree from the University of Hong Kong,
in 1974 and 1988, respectively.
He was with Cable & Wireless, Rediffusion Television, and the General Post Office, all in Hong
Kong, before joining The Hong Kong Polytechnic University, Kowloon, in December 1969 as a
Member of the Academic Staff. He is currently a
Professor in the Department of Electronic Engineering. He is the author of the book Computer-Aided
Analysis and Design of Switch-Mode Power Supplies (New York: MarcelDekker, 1993) and 60 technical papers on the design of power electronics
and analogue circuits.
Dr. Lee is a fellow of the Institution of Electrical Engineers (U.K.) and the
Hong Kong Institution of Engineers.
C. K. Tse (M’90) received the B.Eng.(Hons) and
Ph.D. degrees from the University of Melbourne,
Melbourne, Australia, in 1987 and 1991, respectively.
He is currently an Assistant Professor at The
Hong Kong Polytechnic University, Kowloon,
where his research interests include circuit theory
and power electronics.
Dr. Tse was awarded the L. R. East Prize by the
Institution of Engineers, Australia, in 1988. He is a
Chartered Professional Engineer in Australia.