PIERS Proceedings, Stockholm, Sweden, Aug. 12–15, 2013
1566
A 1-kW Wireless Power Transfer Link for Welding Rollers
R. Trevisan1 and A. Costanzo2
1
DEI, University of Bologna, IMA Industries Srl, Italy
2
DEI, University of Bologna, Italy
Abstract— This paper proposes an innovative approach to inductive wireless power transfer
links based on a contactless energy transfer device. The key parts of the device, that are a
rotary transformer and its windings, are presented. Since many configurations are possible, in
terms of number of turns, wire, and compensation, different configurations are also discussed.
Two setups able to transfer up to 1.3 kW in a contactless fashion are also prototyped and tested
as replacements for slip rings used to supply welding roller resistors of automatic machineries.
Experimental measures carried out on the test bench are used to validate the analytical model
and the finite element simulations of the wireless power transfer link. Trade-offs emerge from the
setups, in particular concerning efficiency and electro-magnetic compatibility. As a consequence,
the choice of the optimal configuration depends on the final application.
1. INTRODUCTION
Traditionally, electrical joints between static and rotatable parts of automatic machinery are realized by mean of sliding contacts, usually slip rings. Older materials employed in sliding contact
manufacturing were subject to early degradation, leading to unacceptable increase in the contact
resistance [1]. In order to overcome this phenomenon and improve contacts’ reliability, some studies
toward innovative brush plates were made [2]. In spite of that, rotatable joints represent a common
point of failure, because their lifetime is strongly related to the cumulated number of revolutions.
For this reason, contactless rotary joints would significantly increase the overall machine’s reliability, reducing costs of maintenance and the consequent downtimes. Contactless rotatable joints may
widely replace the unreliable slip rings employed in rotary organs of automatic machines. A system
based on a rotatable transformer fulfills such requirement. Indeed, an inductive contactless channel
was proposed to achieve power transfer between stationary and rotatable parts of tool machines or
robots [3]. A pot-core transformer is composed by two physically separated half cores, as shown in
Fig. 1. Separation of the halves allows the rotation of the one in respect to the other, as the static
and the rotary part of a slip ring [4].
In this paper, we propose a wireless power transfer link based on a pot-core transformer to
be used as replacement for the slip rings usually applied to supply welding roller resistors. Two
different setups are also presented and their measured performances are compared.
2. INDUCTIVE CET SYSTEMS
Inductive CET systems based on pot-core transformers are usually composed by a switching power
supply, the inductive coupler (i.e., the rotatable transformer, which comprises the half cores and the
windings), and the application load [5]. Optionally, a rectifying stage might be required between
the rotary part of the transformer and the load. The main goal of such system is its ac-ac efficiency,
C1
L lk 1
Lm
R1
N1 : N2
L lk 2
Rc
C2
R load
R2
Swit ching
Powe r Supply
Figure 1: 3D image of two pot half
cores and magnetic flux lines.
Figure 2: Equivalent representation of the inductive CET device.
Progress In Electromagnetics Research Symposium Proceedings, Stockholm, Sweden, Aug. 12-15, 2013 1567
which can be expressed as:
¯
¯2
¯ R
Pload ¯¯
n
¯
load
η =
= ¯ n2 (Z +R )
,
¯
2
load
¯
¯
Pin
Re[Z
in ]
+
1
Zm
N1
1
n =
, Zi = jωLi +
+ Ri , i = 1, 2,
N2
jωCi
jωLm Rc
n2 (Z2 + Rload )Zm
Zm =
, Zin = Z1 + 2
.
jωLm + Rc
n (Z2 + Rload ) + Zm
(1)
(2)
(3)
A circuit equivalent representation of the CET link, is shown in Fig. 2, where the quantities to
compute the system efficiency are outlined. In (1) Pload is the power delivered to Rload and Pin
is the power available at the Switching Power Supply port, in sinusoidal excitation condition. To
effectively account for the strong dependence of system efficiency on the physical parameters of the
pot core and of its windings, a systematic design procedure need to be implemented.
2.1. Rotatable Transformer
To ensure the physical separation of the rotatable transformer of Fig. 1, a small air gap (0.6 mm in
our case) between the ferrite cores is realized. The air gap does not break the magnetic flux path,
though significantly reduces the coupling coefficient of the transformer [6].
An estimation of the core’s inductances can be obtained by the approach presented in [7]. Single
reluctances are computed starting from geometrical and magnetic properties of the core such as
radii, widths, heights, and permeability. The electrical equivalent circuit is then solved to calculate
primary, secondary, and mutual inductances, as known from magnetics theory [8].
Despite the better coupling of pot-core transformers, compared to loosely coupled inductive
power transfer (LCIPT) devices, compensation of leakage inductances would further improve its
efficiency. Four different arrangements (SS, SP, PS, and PP) of the compensation capacitances are
possible, depending on the positioning of each capacitance in series (S) or in parallel (P) to the
primary or secondary winding, respectively [9]. Theoretically, SS compensation represents the best
choice because it does not depend on the load, and therefore is the only considered in the following.
Series capacitors usually compensate the self-inductance by the well-known relationship:
C1 =
1
ω 2 L1
,
C2 =
1
ω 2 L2
.
(4)
2.2. Windings
In order to allow relative rotation, hence physical separation of the halves, the windings must be
detached as well. Thus, two different winding topologies are possible: adjacent windings and coaxial
windings [10]. In the former topology each windings is placed in its own half core, whereas in the
latter windings are concentric. In this work, we choose the coaxial winding topology, because of its
greater reliability under mechanical vibrations.
A number of parasitic effects need to be included in the CET design to obtain accurate prediction
of its performance. First of all, the well-known skin effect which results in an increase in the wire
resistance when ac currents at high-frequencies circulate through them [11]. Second, wires stacked in
layers, the ones above the others, also suffer from the proximity effect, that contributes to reduce the
effective wire’s current cross-section due to the induced magnetic field from adjacent conductors [12].
Many works estimate the equivalent ac resistance of windings in magnetic components, when skineffect and proximit-effect losses are considerable, such as [13, 14]. We have obtained a strong
reduction of these effects by choosing stranded wire, also called litz wire. Sullivan in [15] proposes
a method to optimise the number of strands in a litz wire, by considering wire’s thickness as well
as core’s geometries. Given the dc resistance of the winding Rdc , its dc + ac resistance can be
estimated as follows:
µ
¶
π 2 ω 2 µ20 N 2 n2 d6c k
Rdc+ac = Rdc 1 +
,
(5)
768ρ2c b2c
in which the current is supposed sinusoidal and ω is the radian frequency, n the number of strands,
N the number of turns, dc the diameter of the copper in each strand, ρc the resistivity of the copper,
bc the breadth of the window area of the core, and k a factor usually equal to one.
PIERS Proceedings, Stockholm, Sweden, Aug. 12–15, 2013
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3. CET DEVICE EXPERIMENTAL SETUPS
Slip rings of welding rollers commonly employed in automatic machines represent reasonable components to be replaced with a wireless power transfer links. Therefore, we set up a prototypal test
bench able to transfer up to 1.3 kW through a CET device to a resistive load. Because of the wide
variety of configurations in terms of number of turns, wire types, and compensating capacitances,
we decided to test two different setups in order to seek the optimal trade off between efficiency,
feasibility, reliability, and compatibility. The following considerations emerge:
1. The equivalent input inductance should not be cancelled, otherwise the power inverter that
supplies the CET device would require an additional inductor to achieve resonant commutation
of its switches.
2. The copper losses increase with the winding wire length. Hence, the fewer turns, the better.
3. The ferrite losses increase with the magnetic flux density. Hence, the more turns, the better.
4. The voltage magnitude across C1 (or C2 ) decreases with the inverse of the capacitance, thus
decreases with the increase in the inductance. Hence, the more turns, the better.
The two setups that we will discuss below, named setup 18/17 and setup 27/24, use the same
P66/56 3C81-ferrite core and 50-kHz switching frequency, whereas they adopt different windings,
capacitances, and number of turns. The 3C81 material is a MnZn soft ferrite suitable for power
applications at frequencies up to 200 kHz, with minimal losses around 60◦ C [16]. We pursued the
following design priority constraints for the setups.
3.1. Setup 18/17
Minimal copper loss constraint is pursued. The litz strand section is ideal for 50-kHz signals, and the
number of strands is maximised. Thick wires imply a small number of turns, that results in greater
ferrite losses. 18 turns of 500 × 0.1-mm strand litz wire for the primary winding represent a good
compromise at 50 kHz to prevent saturation. The number of turns (17) of the secondary winding
is chosen to obtain a suitable voltage on the load. According to Eq. (4), the small inductances of
this setup result in big compensating capacitances, thus high voltage ratings (almost prohibitive).
For this reason, we decided not to include any compensation in this case.
3.2. Setup 27/24
Minimal EMI emission constraint is pursued. The highly-distorted current waveform of a purelyinductive CET system (e.g., of setup 18/17) may induce electro-magnetic incompatibilities. Adding
primary compensation results into two benefits: 1. LC filtering network on current’s harmonics; 2.
partially compensation of inductance. Since the load is fixed, only primary compensation can be
adopted. In this case, its value can be computed as follows:
C1 =
1
,
0
ω Im [Z2 + jωLlk1 + R1 ]
Z20 =
n2 (jωLlk2 + R2 + Rload ) Zm
,
n2 (jωLlk2 + R2 + Rload ) + Zm
(6)
With respect to the considerations above, a good compromise is represented by 27/24 turns of
84 × 0.18-mm strand litz wire and C1 = 100 nF. Despite the higher copper losses of 27/24 turns,
lower EMI and lower ferrite losses are expected.
4. RESULTS
The setups described above are used to validate the analytical model presented, including the finite
element simulations of the pot-core transformer.
First, electrical parameters of the circuit-equivalent model are compared with the results of the
FEM simulation. Then, a systematic set of measurements are carried out. The FEM software
used to simulate the transformer is Finite Element Method Magnetics (FEMM) free software [17].
Comparisons among electrical parameters of the analytical model, FEM simulation based on the
physical quantities of the CET link, and the experimental measurements are summarized in Table 1
and in Table 2 for the setup 18/17 and the setup 27/24, respectively. Measures of the efficiency at
the given conditions are also shown, and the input impedance of the CET link is plotted in Fig. 3
for the two configurations.
Table 3 provides a short comparison between the tested setups. Although either configuration
is able to transfer more than 1 kW of electrical power by loosing less than 5% in ferrite and copper
losses, the second setup also guarantees low EMI with moderate additional efforts in terms of circuit
Progress In Electromagnetics Research Symposium Proceedings, Stockholm, Sweden, Aug. 12-15, 2013 1569
Figure 3: Electrical parameters of setup 18/17.
R1
R2
Llk1
Llk2
Lm
η
Analytical
54.6
47.2
0.028
0.025
0.303
–
FEM
–
–
0.030
0.027
0.296
–
Measured
–
–
0.048
0.010
0.269
0.977
Units
mΩ
mΩ
mH
mH
mH
·
Figure 4: Electrical parameters of setup 27/24.
R1
R2
Llk1
Llk2
Lm
η
Analytical
255
174
0.063
0.050
0.682
–
FEM
–
–
0.068
0.053
0.671
–
Measured
–
–
0.065
0.049
0.670
0.956
Units
mΩ
mΩ
mH
mH
mH
·
Figure 5: Input impedance of the CET link.
Table 1: Comparison between setups.
Complexity
Ferrite losses
Copper losses
Efficiency
EMI
Setup 18/17
Low
Medium
Very Low
Very High
High
Setup 27/24
Medium
Low
Medium
High
Low
complexity. Further tests should be carried out in order to verify the effective electro-magnetic
compatibility of the proposed solutions. Finally, the plots of the measured input and output power
waveforms are shown in Fig. 4 and Fig. 5 for setup 18/17 and setup 27/24, respectively. The
bad shape of the power waveforms in setup 18/17 warns that severe EMI may occur. In case of
extensive wiring of the automatic machine, such interference might become a serious issue and must
be taken into account. Yet, the setup 27/24 represents a suitable solution for EMI, despite the
slight inefficiency with respect to setup 18/17.
Figure 6: Power waveforms of setup 18/17.
Figure 7: Power waveforms of setup 27/24.
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PIERS Proceedings, Stockholm, Sweden, Aug. 12–15, 2013
5. CONCLUSIONS
We have introduced a solution for a wireless power transportation link, to transfer up to 1.3 kW,
which represent a valid replacement for slip rings employed in welding rollers of automatic machines.
We have adopted an analytical design procedure starting from a reference circuit equivalent scheme
of the CET link. The design results are checked by a suitable set of measurements carried out on
the prototype confirming the accuracy of the analytical models adopted. Two different solutions
are compared to outline the different trade-offs that may be required in different CET application
scenarios.
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