A CMOS SUBHARMONIC MIXER WITH
INPUT AND OUTPUT ACTIVE BALUNS
Brad R. Jackson and Carlos E. Saavedra
Department of Electrical and Computer Engineering
718 Walter Light Hall
Queen’s University
Kingston, ON K7L 3N60, Canada
Received 28 April 2006
ABSTRACT: A CMOS 0.18 ␮m subharmonic mixer is experimentally
demonstrated that uses active baluns at the local oscillator (LO) and RF
inputs, as well as at the output. With this subharmonic mixer, a 2.1 GHz
RF input and a 1.0 GHz LO input produce a 100 MHz output signal.
The conversion gain is 8 dB, the LO and RF input reflection coefficients are better than ⫺10 dB, and IIP3 is ⫺8.5 dBm. © 2006 Wiley
Periodicals, Inc. Microwave Opt Technol Lett 48: 2472–2478, 2006;
Published online in Wiley InterScience (www.interscience.wiley.com).
DOI 10.1002/mop.21957
Key words: RF CMOS; subharmonic mixer; active balun; MMIC; direct conversion receiver
1. INTRODUCTION
During the past several years there has been considerable interest
in direct conversion receivers. The primary advantage in using
direct conversion is that there is no image frequency produced, and
consequently much simpler and inexpensive filters can be used.
However, direct conversion receivers have disadvantages such as
local oscillator (LO), self-mixing at the receiver that can seriously
degrade the performance of a receiver through increased noise and
intermodulation distortion [1]. To combat this problem, several
techniques have been suggested such as the use of a frequency
doubler at the output of the LO [2] and the use of a subharmonic
mixer [3–7]. With a subharmonic mixer, the LO frequency is
effectively doubled, thus producing mixing components from the
RF frequency and double the LO frequency. The ability to use an
LO with half the frequency that would have been otherwise required can simplify the LO design and improve the performance.
At very high-frequencies in particular, it may be difficult to design
an LO with the required output power and phase noise, which
could make the subharmonic mixing technique attractive for applications other than in direct-conversion receivers. Many mixers
rely on a Gilbert-cell [8, 9]. Modifications to this topology were
shown in Refs. 3 and 4 to realize a subharmonic mixer where four
LO signals are used with relative phase shifts of 0°, 90°, 180°, and
270° to effectively provide switching at twice the rate of the
traditional Gilbert-cell. A block diagram of the subharmonic mixer
using this technique is shown in Figure 1. Often, the use of
Figure 1
case)
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Subharmonic mixer concept (output shown for downconverter
single-ended, or unbalanced, signals is necessary, in particular
when interfacing with off-chip components in multichip modules.
Since the Gilbert-cell relies on differential signals, baluns must be
used to perform this conversion if single-ended signals are used.
CMOS subharmonic mixers are presented in Refs. 5–7; however,
only simulation results are shown, and conversion between balanced and unbalanced signals is not discussed. In this work, the
design and measured results of a CMOS subharmonic mixer that
uses active baluns at the input and at the output is presented.
Section 2 will discuss the subcircuits in the proposed design,
Section 3 details the measured results, and Section 4 concludes the
paper.
2. CIRCUIT IMPLEMENTATION
2.1. Subharmonic Mixer Core
The core of the subharmonic mixer is shown in Figure 2, which is
based on the Gilbert-cell topology [10]. Gilbert-cell mixers, in
general, have high isolation between ports due to their doublebalanced structure. Corresponding to the block diagram in Figure
1, the circuit requires RF inputs with relative phase shifts of 0° and
180°, and LO inputs of 0°, 90°, 180°, and 270°. With this topology,
first proposed in [3], the currents I1 and I2 are effectively switched
at twice the LO frequency, which is the mechanism that results in
subharmonic mixing. For insight into how the LO frequency is
doubled, consider the circuit in Figure 3. Since the LO input is
generally a large signal, the MOSFETs will turn on and off
corresponding to the amplitude of the voltages at their gates. As
the 0° LO signal rises well above the threshold value, transistor M5
turns fully on, causing I1 to increase and flow predominately
through M5, since the gate voltage at M6 is 180° out of phase and
therefore near the minimum (cutoff). When the amplitude of the 0°
LO signal begins to drop and the 180° LO amplitude begins to rise,
neither transistor is fully on and, as a result, the current I1 decreases. As the 180° LO signal nears its maximum and the 0° LO
nears its minimum, M6 is turned fully on and M5 is turned off,
meaning that the current I1 increases and flows through M6. By the
end of the cycle, neither the 0° nor the 180° LO signals are at a
maximum and the current I1 decreases again. Therefore, during
one period of the LO signal, I1 has two cycles of increasing and
decreasing current, thus indicating a doubling of the LO frequency.
The same operation occurs for the other LO transistor pair (90° and
270°) and the resulting current, I2, is 180° out of phase with I1.
Therefore, mixing will occur at the RF frequency and twice the
input LO frequency.
To improve the linearity of the mixer, source degeneration was
used. This allows an increase in the amplitude of the input signal,
but has the penalty of reducing the conversion gain. Resistive
degeneration was used in this work rather than inductive because
it requires significantly less silicon space and because it does not
have a frequency dependence. As a result, a more compact layout
of the circuit can be obtained compared to implementations with
inductive degeneration. However, with resistive degeneration the
noise figure of the mixer will be increased somewhat. The output
of the mixer is taken differentially between the drains of M1 and
M4.
2.2. RF and LO Input Baluns
In many cases it is necessary to convert an unbalanced (singleended) signal to a balanced (differential) signal and vice versa.
Passive on-chip baluns for RF and microwave frequencies can
consume a large area specifically at lower frequencies. For this
reason, baluns are often realized off-chip, which adds to assembly
costs and may degrade the conversion gain/loss. It is possible to
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 48, No. 12, December 2006
DOI 10.1002/mop
Figure 2 Core circuit of the subharmonic mixer
use transmission line baluns on-chip, but since their size is proportional to wavelength they are not economically feasible except
possibly at millimeter-wave frequencies (⬎30 GHz). Several techniques exist to perform the balun operation using active components. The simplest active balun is a FET with resistors in the drain
and the source as shown in Figure 4 [11]. By properly choosing the
value of these resistors, the amplitude of the two outputs can be
made equal. Specifically, there is approximately unity gain at V01
(i.e. V01/VIN ⬇ 1) since it is a follower circuit. Therefore, to have
equivalent output amplitude at V02, a design equation can be easily
obtained for the resistors R1 and R2 to a first-order approximation
with V02 /VIN ⫽ ⫺ R1 /共R2 ⫹ 1/gm兲 ⫽ ⫺ 1 and R1 ⫽ R2
⫹ 1/gm. The negative sign in V02/VIN indicates that the output at
Figure 3
LO transistor pair used to double the LO frequency
DOI 10.1002/mop
the drain has (ideally) a 180° phase-shift relative to the input,
whereas V01 has the same phase as the input. Of course, this basic
structure has limitations at high frequencies due to the parasitic
elements associated with the transistor. In particular, the gate-drain
parasitic capacitance, Cgd, seriously degrades the performance at
high frequencies since the input signal can feed through this
capacitance directly to the output. Two techniques that have improved performance are the differential pair and the cascaded
common-gate/common-source (CG-CS) [12], shown in Figures 5
and 6, respectively. The differential pair circuit ultimately has a
similar frequency limitation as the circuit in Figure 4 by considering the parasitics in the half-equivalent circuit. Several other
active balun circuits have been proposed [13, 14]; however, the
CG-CS pair of Figure 6 was chosen in this work, which has an
advantage over the differential pair in that active input matching is
possible, although its performance is more sensitive to process
variations. It is desirable to have a low input reflection coefficient
so that the input power will be absorbed by the circuit, and not
reflected. Since the input impedance to the gate of a FET is
typically very high due to its large capacitive component, the input
reflection coefficient to a CS device, or a differential pair, is
generally poor. In contrast, the input impedance to a CG device is
approximately 1/gm. Therefore, an appropriate selection of device
size and biasing can yield a 50 ⍀ input impedance, as desired.
Since the input impedance of the CG device is in parallel with the
very high input impedance of the CS transistor, the resulting input
impedance is approximately that of the CG transistor. This is a
significant advantage over other topologies that do not have an
acceptable input reflection coefficient. In these instances, a match-
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 48, No. 12, December 2006
2473
Figure 5
Differential pair as a balun
in Figure 8 with relative phase shifts indicated at the output for
clarity. Using a reference 0° phase of an LO input signal, the phase
shift at the outputs in Figure 8 would be ⫺45°, 45°, 135°, and 225°
from top output to bottom relative to the input phase.
Figure 4 Single FET balun
ing network must be implemented using passive devices such as
transmission lines, or inductors and capacitors. In this case, the
area needed will be much larger and the input matching response
will generally not be as broadband as with a CG device, although
the reflection coefficient could possibly be lower. The resistor, Rb,
is large enough that it has a very small impact on the input
reflection coefficient. The gate of transistor M1 is biased at Vdd
(1.8 V) and the gate of M2 is biased by at voltage set by the drop
across Rb (which is determined by the current through M1).
The balun used for the input RF port was simulated independently in the Spectre simulator using a Cadence extracted layout
that included parasitic capacitances and source-follower buffers at
the outputs. The amplitude and phase balance are shown in Figure
7. The difference in amplitude of the two outputs less than 0.1 dB
over the range from 1.0 to 3.0 GHz and the phase difference is less
than about 3°.
For this circuit to operate as a subharmonic mixer there must be
four LO signals with relative phase shifts of 0°, 90°, 180°, and
270°. The LO input active balun generates the 0° and 180° signals,
therefore an additional technique must by used to generate the
quadrature signals. A straight-forward technique to create the 90°
phase shifts is with resistor– capacitor polyphase filters. RC-CR
quadrature generators are a narrow-band solution with an accuracy
that is dependent on the process tolerances. The networks were
designed to create a phase shift of ⫾45° at 1 GHz by using a
resistance value of Rp ⫽ 320 ⍀ and a capacitor, Cp ⫽ 0.5 pF. The
complete LO input circuit with balun and phase shifters is shown
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2.3. Output Balun
At the output of the subharmonic mixer the signal is differential.
To convert to a single-ended signal at this point is very convenient
using an active balun since in this case the frequency of the signal
Figure 6 Cascaded common-gate/common-source balun
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 48, No. 12, December 2006
DOI 10.1002/mop
Figure 9 Output balun and buffer
Figure 7 CG-CS balun amplitude and phase balance simulations from
extracted layout circuit using Cadence Virtuoso and Spectre simulator
through the balun due to the reduced performance of the differential pair at high-frequencies, thus providing a minor amount of
filtering.
is relatively low at 100 MHz. In fact, given the low-frequency of
the signal, an active balun is even more attractive than a passive
balun due to the much smaller size of the circuit. A differentialpair with a single-ended output connected to a source follower
accomplishes the desired goal. This output balun circuit with
buffer is shown in Figure 9. The differential pair was designed so
it would perform as a balun with unity gain. Therefore, the singleended output signal from the differential pair has the same amplitude as the balanced signal at the output of the mixer. Of course,
high harmonic frequency components will be attenuated somewhat
Figure 8
DOI 10.1002/mop
2.4. Complete Circuit
The layout for the complete circuit required an area of ⬃600 ⫻
700 ␮m2 (0.42 mm2) including bonding pads and ⬃400 ⫻ 500
␮m2 (0.2 mm2) for the circuit excluding bonding pads. The
layout is relatively compact, which can be attributed to the fact
that there are no inductors in the design. Commonly, inductors
are required as part of an input matching network to improve
input reflection coefficient. In this case, since the active baluns
were designed to have good input matching, inductors, or an
LO input balun and quadrature phase shifters
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 48, No. 12, December 2006
2475
Figure 10 Output IF (100 MHz) power versus RF input power at 2.1
GHz (LO power ⫽ ⫺10 dBm at 1 Ghz)
off-chip matching network can be avoided. As aforementioned,
degeneration resistors were used rather than inductors to improve the linearity, but require significantly less space on the
integrated circuit.
3. MEASURED RESULTS
To measure the subharmonic mixer, coplanar waveguide probes
were used to contact the on-chip pads and a Rohde & Schwarz
FS300 spectrum analyzer was used to measure the output power
of the various harmonics. The power supply voltage for the
circuit, VDD, was set to 1.8 V. To measure the conversion gain
of the mixer, the LO was set to a frequency of 1 GHz and a
power of ⫺10 dBm. A 2.1 GHz RF input was used and its
power was swept from approximately ⫺30 to ⫺5 dBm. The
output power at (2.1 ⫺ 2 ⫻ 1.0) GHz ⫽ 100 MHz was observed
and the measured and simulated results are shown in Figure 10.
From this figure, the conversion gain is ⬃8 dB and, the 1-dB
Figure 11 Conversion gain versus LO power (RF power ⫽ ⫺20 dBm)
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Figure 12
RF input reflection coefficient
compression point occurs at an input RF power of approximately ⫺13 dBm. The measurements match the simulation
results closely. To find the optimal LO power that provides the
maximum conversion gain, the RF input power was held constant while the LO power was swept from ⫺20 to 0 dBm. Figure
11 shows that the gain increases relatively linearly with increasing LO power until around ⫺10 dBm at which point the
conversion gain is ⬃8 dB. From the spectrum data it was found
that an LO power of ⫺10 dBm provides the maximum conversion gain while suppressing all undesired harmonics by at least
25 dB. A full two-port calibration was performed using a
calibration substrate and an HP 8510C network analyzer was
used to measure the input reflection coefficients. Measurements
were performed for the input matching of both the RF and the
LO ports. The S11 for the RF input is shown in Figure 12. It is
clear that the active input balun provides good matching with an
input reflection coefficient less than ⫺10 dB at 2.1 GHz.
Similarly, the LO input reflection coefficient, shown in Figure
13, is approximately ⫺11 dB at the LO frequency of 1 GHz.
Figure 13 LO input reflection coefficient
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 48, No. 12, December 2006
DOI 10.1002/mop
The slight difference between the RF and the LO input matching is due to minor differences in the active balun circuits that
allows a lower input reflection coefficient at the operational
frequency of the LO at the expense of higher power consumption. The third-order intercept point (IP3) was measured by
using a two-tone RF input signal with frequencies of 2.1 and
2.11 GHz. This produced third-order intermodulation products
at 90 and 120 MHz. The results shown in Figure 14 indicate an
IIP3 of ⫺8.5 dBm and an OIP3 of ⫺0.5 dBm. The measured
double sideband (DSB) noise figure is ⬃20 dB, due primarily to
the input baluns rather than the mixer core circuit. The noise
figure could be reduced by modifying the input baluns to
eliminate the bias resistor Rb, or by using another balun topology. The power consumption for the circuit, including input and
output baluns, the mixer core, as well as the output buffer, is 36
mW. A microphotograph of the chip is shown in Figure 15.
4. CONCLUSION
A CMOS 0.18 ␮m subharmonic mixer was demonstrated that
uses active input and output baluns to generate differential
signals from single-ended, and vice versa. This configuration
could be used as part of a direct conversion receiver where
single-ended inputs and outputs are required or in a super
heterodyne receiver where a lower LO frequency is desired. By
using a subharmonic mixer, the LO frequency is designed to
have half the frequency that would be required with a fundamental mixer. Consequently, the design of the LO can possibly
be simplified and improved performance can result due to the
reduced oscillation frequency. The RF input frequency was 2.1
GHz and the LO frequency was 1.0 GHz, which results in an
output IF of 100 MHz. Active baluns at the RF and LO input
ports were designed to have active input matching and the
measured results show input reflection coefficients of less than
⫺10 dB at the LO and RF operational frequencies. The 1-dB
compression point was found to occur at ⫺13 dBm RF input
power and the conversion gain was ⬃8 dB. The circuit shows
excellent suppression of the harmonics, as well as excellent
isolation from feed-through of the RF and LO signals to the
output. If a quadrature oscillator is used for the LO it could
eliminate the need for the passive RC-CR phase-shifters at the
output of the LO balun, and potentially improve performance
through the potential increase in phase accuracy.
Figure 14 Third-order intercept point determination
DOI 10.1002/mop
Figure 15 Microphotograph of subharmonic mixer with input and output active baluns. [Color figure can be viewed in the online issue, which is
available at www.interscience.wiley.com]
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(1997), 2097–2104.
3. K. Nimmagadda and G. Rebeiz, A 1.9 GHz double-balanced subharmonic mixer for direct conversion receivers, In Proceedings of IEEE
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USA, May 20 –22, 2001, pp. 253–256.
4. Z. Zhaofeng, L. Tsui, C. Zhiheng, and J. Lau, A CMOS self-mixingfree front-end for direct conversion applications, In Proceedings of
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5. P. Upadhyaya, M. Rajashekharaiah, Y. Zhang, D. Heo, and Y.-J. Chen,
A 5 GHz novel 0.18 ␮m inductor-less CMOS sub-harmonic mixer, In
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6. V. Krizhanovskii and S.-G. Lee, 0.18 ␮m CMOS sub-harmonic mixer
for 2.4 GHz IEEE 802.15.4 transceiver, In Proceedings of IEEE 14th
International Crimean Conference on Microwave and Telecommunication Technology, Sevastopol, Crimea, Ukraine, Sept. 13–17, 2004,
pp. 141–142.
7. P. Upadhyaya, M. Rajashekharaiah, and D. Heo, A 5.6 GHz CMOS
doubly balanced sub-harmonic mixer for direct conversion—Zero
IF receiver, In Proceedings of IEEE Workshop on Microelectronics
and Electron Devices, Boise, Idaho, USA, April 16, 2004, pp.
129 –130.
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(1997), 1151–1155.
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double-balanced direct conversion mixer with an integrated wideband
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© 2006 Wiley Periodicals, Inc.
DESIGN OF COMPACT RFID READER
ANTENNA WITH HIGH
TRANSMIT/RECEIVE ISOLATION
Hae-Won Son, Jung-Nam Lee, and Gil-Young Choi
RFID System Research Team
Electronics and Telecommunications Research Institute (ETRI)
161 Gajeong-dong, Yuseong-gu, Daejeon 305–700, Korea
Received 1 May 2006
ABSTRACT: A radio frequency identification (RFID) reader antenna
with high transmit/receive isolation, which consists of two circularlypolarized radiating patches, is presented. The high isolation is
achieved by the symmetric design of the antenna geometry and
proper arrangement of four feed points on the patches, which are fed
by branch-line hybrid couplers. The experimental results show the
transmit/receive isolation of more than 36 dB in the 860 –960 MHz
RFID frequency band. © 2006 Wiley Periodicals, Inc. Microwave
Opt Technol Lett 48: 2478 –2481, 2006; Published online in Wiley
InterScience (www.interscience.wiley.com). DOI 10.1002/mop.22017
Key words: RFID reader antenna; high isolation; circular polarization
1. INTRODUCTION
Radio frequency identification (RFID) has recently attracted
much interest in supply chain management by retailers and
manufacturers. In a passive UHF RFID system, tags are powered up by a continuous-wave (CW) RF signal transmitted by a
reader, and backscatter transmission from the reader to send
back their data [1]. In a backscatter reader, the transmitted CW
signal may be directly coupled to the receiving part of the
reader to drastically degrade the receiving sensitivity. The
directly-coupled CW signal is much larger than the backscattered signal from tags, and the receiving part of the reader
should detect the weak signal close to such a strong in-band
interferer. Therefore, it is very critical to achieve high isolation
between the transmitting and receiving parts for a good performance reader.
Toward this end, the reader antenna can be configured with two
separated radiating elements for transmit (Tx) and receive (Rx),
and their separation distance is set to be large enough to give the
required Tx/Rx isolation [2]. In this type of reader antenna, the
2478
overall antenna size is highly dependent upon the separation distance between two radiating elements, and larger antenna size is
needed to achieve higher Tx/Rx isolation. In this paper, we present
a novel design of a compact backscatter reader antenna with
circular polarization (CP) and high Tx/Rx isolation. In practice, a
CP reader antenna is highly desired to ensure tag visibility regardless of tag orientation. Few papers have been reported on the CP
antenna with high Tx/Rx isolation although many papers have
been published on improving the isolation of the dual linearlypolarized antenna [3– 6].
2. ANTENNA DESIGN AND RESULTS
The geometry of the proposed reader antenna is shown in Figure
1. The antenna consists of two dual-feed radiating patches fed
by branch-line hybrid couplers. The radiating patches are circular metal plates placed over the ground plane at a height of hp.
Each branch-line coupler is printed on a thin PTFE substrate (
␧ r ⫽ 3.5, tan␦ ⫽ 0.0018) and inset between the radiating patch
and the ground plane. As shown in Figure 1, the ground body of
the proposed antenna is comprised of a pair of hexahedral
cavities surrounding the radiating patches to reduce the mutual
coupling between the patches. The cavities have a circular
aperture with the same size of the radiating patch in the
broadside direction. The overall size of the proposed antenna is
only W ⫻ L ⫻ H ⫽ 200 ⫻ 450 ⫻ 30 mm3, and this is very
compact compared to commercially available RFID reader antennas [2].
In Figure 1, two feed points (a, b) of the Tx patch are fed 90°
out of phase with respect to each other by the Tx coupler. There are
two possible Tx ports (T1, T2). The LHCP (left hand circular
polarization) wave is radiated when port T1 is used as a Tx port,
while the RHCP (right hand circular polarization) wave is radiated
for port T2. In the same manner, two feed points (c, d) of the Rx
patch are fed by the Rx coupler with two possible Rx ports (R1,
R2), and RHCP wave is received by port R1 and LHCP wave by
port R2.
Figure 2 shows the equivalent circuit of the proposed antenna.
Two equivalent four-port networks of the branch-line coupler and
one equivalent 4-port network representing the mutual coupling
between four feed points (a, b, c, d) are cascaded. The scattering
matrix 关SC] for the ideal 3 dB branch-line coupler is as follows
when all ports are matched:
冤
S11
S21
关S 兴 ⫽ S
31
S41
C
S12
S22
S32
S42
S13
S23
S33
S43
冥 冤
冥
S14
0 j 1 0
1 j 0 0 1
S24
S34 ⫽ ⫺ 冑2 1 0 0 j .
S44
0 1 j 0
(1)
The scattering matrix 关SM] for the equivalent four-port network,
representing the mutual coupling between four feed points (a, b, c,
d), is expressed as follows when all ports are matched:
冤
冥
0 S ab S ac S ad
S ba 0 S bc S bd
关S 兴 ⫽ S
S cb 0 S cd .
ca
S da S db S dc 0
M
(2)
From Eqs. (1) and (2), the transmission coefficients of SR 1T 1 and
SR 2T 1 are given as follows:
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 48, No. 12, December 2006
DOI 10.1002/mop