RF Test Methods for Balanced
Receivers for Swept Source Optical
ARCHIVES
Coherence Tomography
by
ByungKun Lee
[SB EE, MIT, 2011]
Submitted to the Department of Electrical Engineering and Computer Science
in Partial Fulfillment of the Requirements for the Degree of
Master of Engineering in Electrical Engineering and Computer Science
at the Massachusetts Institute of Technology
May 21, 2012
©2012 Massachusetts Institute of Technology
All rights reserved.
Author:
Department of Electrical Engineerina and Computer Science, May 21, 2012
Certified by:
Prof. James G. Fujimoto, Thesis SupdbtiSor, Signed by Dorothy A. Fleischer, May 21, 2012
Accepted by:
Prof. Dennis M. Freeman, Chairman, Masters of Engineering Thesis Committee
RES
RF Test Methods for Balanced Receivers for
Swept Source Optical Coherence Tomography
by
ByungKun Lee
[SB EE, MIT, 2011]
Submitted to the Department of Electrical Engineering and Computer Science
in Partial Fulfillment of the Requirements for the Degree of
Master of Engineering in Electrical Engineering and Computer Science
at the Massachusetts Institute of Technology
May 21, 2012
©2012 Massachusetts Institute of Technology
All rights reserved.
Abstract
Optical coherence tomography (OCT) has risen as a clinical standard of diagnosis and
management of ocular diseases since its development in 1991 by the MIT group and the
collaborators. Since current cutting-edge OCT technology based on frequency-swept lasers
has achieved scanning rate over 1,000,000 axial scans per second, the imaging speed is
limited by the detection and analog-to-digital conversion stages. In order to match the rapid
advancement of OCT imaging speed, a variety of balanced photoreceivers have been
developed.
A low-cost setup for systematic performance evaluation of the receivers in radio
frequency (RF) range up to 2GHz is presented. The test procedure, including measurements
of gain, bandwidth, and harmonic distortion, is automated by National Instruments Virtual
Instrument Software Architecture (NI-VISA) programming using USB and GPIB interface.
Since the test equipment has parasitic response, quasi-calibration using a fast biased detector
is necessary. Detailed description of the equipment and the test protocol is included as well as
the performance comparison of the available receiver products and prototypes.
2
1 Introduction
1.1 Optical Coherence Tomography
Optical Coherence Tomography (OCT) is an emerging modality in the field of biomedical
imaging. Analogously to ultrasound B-mode scan, OCT performs real-time cross-sectional
and three-dimensional (3D) imaging of internal structure of various subjects in vivo at high
speed and microscale resolution, using light rather than acoustic waves. Since its first
demonstration in 1991 by Huang et al. [1], numerous leading research groups have improved
OCT in terms of the quality of imaging and the broadness of application. The following
sections present a brief introduction on the theoretical background of OCT as well as its
applications in clinical diagnosis and disease management.
1.1.1 OCT in Clinic
OCT is a powerful non-invasive imaging method of living tissues with consistent
repeatability. OCT plays an important role when conventional biopsy is hazardous or
impossible: tissues such as the eye, arteries, and nervous tissues are the most widely used
applications. Moreover, OCT can be used as a previewing method when standard excisional
biopsy has low sensitivity. For example, if histology, a standard biopsy method for cancer
diagnosis, misses the lesion, a false negative occurs; since excisional biopsy is usually timeconsuming, OCT plays an important role of guiding the standard biopsy to reduce the number
of biopsies required for a diagnosis.
One of the earliest and most popular applications of OCT is human retinal imaging [2].
OCT is the key method for monitoring retinal diseases such as glaucoma and age-related
macular degeneration (AMD) which are two of the leading causes of blindness [3]. Figure
1(a)-(e) show OCT images of a human retina and anterior segment of the eye, each presenting
3
6mm (2000 pixels)
I~
~3.5mim
(500 pixels)
6mm (2000 pixels)
7mm (4500 pixels)
Figure 1 [4]. Examples of ophthalmic OCT images. (a) 3D volumetric OCT data set of optic nerve head
consisting of 500x500 transverse pixels acquired at 100 kHz axial scan rate in 2.6 seconds. (b) Macula
and (c) optic nerve head cross-sectional OCT images acquired at 100 kHz consisting of 2000 axial scans
over 6 mm. (d) OCT cross-sectional image of the anterior angle (average of 2 images). (e) A long
imaging range system configuration with 7.5 mm range enables viewing the anterior segment, including
the cornea, iris, and front of the lens.
different views [4]. Current OCT imaging technique provides cross-sectional images up to 12pm axial resolution and wide-field 3D images as well as fundus camera view images.
OCT imaging in tissues other than the eye was initiated after the recognition that light
with longer optical wavelengths is less scattered and can increase imaging depths. Current
major applications of non-ocular OCT include intravascular imaging for detection of arterial
diseases and human endoscopy for guiding histopathology in various organs such as the
breast [5], the stomach, and the intestine [6]. Figure 2 [7] shows an example of clinical
applications of OCT other than in ophthalmology.
1.1.2 Coherence Gating and Low-Coherence Interferometry
The theory of OCT is briefly revisited in the next three sections. Let us begin with a classic
optical measurement called low-coherence interferometry, or white light interferometry,
which is essentially the origin of OCT. In low-coherence interferometry, a broadband light
4
Figure 2 [6]. 3D-OCT images of a normal gastro-esophageal junction (GEJ). (a) En face projection
OCT image at a depth of 350 pm. Regions with gastric mucosa and squamous mucosa show distinct
features. (b) Cross-sectional OCT image along the probe pullback direction shows the GEJ and normal
squamous epithelium clearly. Scale bars: 1 mm.
source with short coherence length is used to generate coherence gating, which enables
precise determination of the axial position of a reflective object. In the 1980s, low-coherence
interferometry was used to measure optical echoes and backscattering in optical fibers and
waveguides [8-10]. Axial eye length measurement by Fercher et al. [11] is the first biological
application of low-coherence interferometry. Since then, numerous variants of low-coherence
interferometry were developed for non-invasive measurements of biological tissues [12, 13].
Interferometry techniques perform correlation measurement of two optical signals coming
from one light source, when one signal is scattered back from the sample and the other signal
travels a known distance in the reference path. By detecting the intensity of the combined
signal, interferometry measures the field, rather than intensity, of the signal scattered back
from the sample. Figure 3(a) shows a schematic diagram of a Michelson interferometer,
where the incident light is split into the reference beam ER(t) and sample beam Es(t). First,
consider the case when both the sample arm and the reference arm have a mirror at the end. If
the source is monochromatic, namely E, (z, t) = Eoej'*-kz), the reference field and the sample
5
Reference
MirrorV
Source
ER
E
zR
~
El
Z
Sample
z =0Reference
Source(E
Eber
s
Detector I oc
Coupler
Sample
Detector
ER +Es
(b)
(a)
Figure 3. Schematic diagrams of a time-domain OCT system. (a) TD-OCT using a Michelson
interferometer. (b) Fiber-optical implementation.
field seen at the detector are ER (t)= EOrRe
2
R)
and Es(t)=Ersej(wt-2kS), where r
2
R
and rs are the field reflectivity values of the reference arm mirror and the sample arm mirror,
respectively. The factor 1/2 comes from the beamsplitter loss. Then, the output intensity is
expressed in terms of the path length difference Az = ZS - ZR as
I=
ce|ER(t)+Es(t)| 2
=
8c|E|
[RR2+
R22RRRs cos(2kAz)].
(1)
where the power reflectivity is defined as R = r| 2 . In this case, the output intensity oscillates
periodically respect to Az and thus the measurement does not give axial distance information.
However, the axial distance information can be obtained with a polychromatic light source.
For polychromatic light, the output intensity can be obtained by integrating contribution from
each frequency components:
ceEo
I
I
1
[rRe-j2k(w)R + rse-j 2k(w):s
()e'''
{
2 dc
(2)
2
=-cE f E0 (c)j' R R2 +Rs2 + 2RRs cos [2k(9) Az|j dco
6
where k(a)) = w/c in free space. Now let us assume that the source has a spectrum whose
shape is Gaussian,
0E)1
2
= Ae('
Then, the output intensity becomes
)/(Aw).
2
12AI = ceA
e
(A
?)2
d j (R 2 + R s + 2RRRs e
(A^)2
cosI
c
11
d
(
Integrating this expression over the emission band of the source, we obtain
(Ar)2-
I
=csA_ACO~
RR2 +R,
8
2
+2RRRse
(cA,)
cos(2koAz)
(4)
where ko = k(coo) = co/c is the wavenumber associated with the central frequency. We can
see that the cross-correlation term cos(2kAz) is only visible if the reference arm position is
within the coherence length, 1c =
1ii75 c/Aco,
from the zero-path-length-difference position.
This effect, called coherence gating, allows low-coherence interferometry to precisely
determine the echo delay of the reflected light or the location where the light was reflected. In
practical imaging situation, the sample is a combination of multiple reflective layers instead
of a single perfect mirror. The output intensity is given by
I
Yce
E ()
e''
=
LRRe-j2k~~zR
S2(IA)
2
RR
cos [2ko (zsn -zR
(Re
nn(5)
DC
+2
2
(Zs.-ZR )2
! ceAVAm
ceVAwR R 2 +j
+
8
d
+ jRsnej2k()s"j
RsnlRs 2 e
2
(csw)
interference of the sample and the reference beams
cos[2ko (zsn,
-zsn
-
self-interference of the sample beam
The result of the integral consists of the DC terms and the interference terms between two
sample layers as well as the interference terms between a sample layer and the reference
mirror. Therefore, if the self-interference artifact in the sample arm is kept small, the intensity
7
of light scattered back from the sample at different axial positions or depths can be detected
by scanning the position ZR of the reference mirror. This technique directly connects to timedomain OCT (TD-OCT), the most fundamental form of OCT.
1.1.2 Time-Domain OCT
One of the earliest OCT systems are based on time-domain approach (TD-OCT). As
previously mentioned, time domain systems use Michelson interferometer with the sample
placed in one arm and a plane mirror placed in the other arm for reference. The system is then
illuminated by a polychromatic light source as in Figure 3(a) where the field at the detector
plane is the sum of the field associated with the scattered wave from the sample and the
reflected wave from the reference mirror. The image data is collected by recording the
intensity output I of the detector while the position ZR of the reference mirror is scanned to
control the echo time-delay of the light. One scan of the reference mirror position
corresponds to one axial scan (A-scan) of optical signal; the reference mirror is periodically
scanned to obtain a cross-sectional image, which is an array of A-scans along the line of
measurement. The system can be also implemented using fiber optics as shown in Figure 3(b).
One of the most critical limitations of TD-OCT is its imaging speed. Maximum imaging
speed of time-domain systems is only up to several hundreds of A-scans per second, due to
the physical limit to the scanning frequency of the reference mirror. Imaging speed is a
crucial factor for wide-field, high-resolution imaging: if the system operates at lateral
resolution of 50um, a 10mmx 10mm imaging field is equivalent to 200x200 A-scans. In TDOCT, this requires several minutes of data acquisition time, which is unrealistic in clinical
applications. In order to overcome the fundamental speed limit of the TD approach and
achieve microscale lateral resolution, Fourier-domain approach was developed as the next
generation technology.
8
1.1.3 Fourier-Domain OCT
The principle behind Fourier-domain OCT (FD-OCT) is Wiener-Khinchin theorem, which
states that the autocorrelation and the spectral power density of a signal are related by Fourier
transform [14]. In FD-OCT, the Fourier-domain power spectrum is measured and
transformed back to the autocorrelation function or the interferogram, the output intensity
respect to path length difference, using the discrete Fourier transform (DFT). FD-OCT
overcomes the mentioned physical limitations of TD-OCT by avoiding the direct scanning of
the reference arm length and thereby achieves higher imaging speed and sensitivity.
The implementation of FD-OCT can be based on either spectrometer or wavelength-swept
light source. In spectral/Fourier-domain OCT (SD-OCT), a broadband light is used, and the
differential components of the intensity spectrum are simultaneously collected by a detector
array placed at the output of a spectrometer [15-17]. In swept source/Fourier-domain OCT
(SS-OCT), the spectral components are spread in time instead of space: while the
wavenumber of a monochromatic laser source is linearly sweeping, the spectral components
are sequentially measured by a photoreceiver [18-21] as shown in Figure 4.
(c)
(a) Reference
Detector output
Sample
Beam
spitr
AL
(b)
tine
(d) Fourier transform
P
Frequency- Depth
Figure 4. Outline of SS-OCT. (a) Interferometer with path difference AL . (b) Sample-arm wave
(dotted) and reference-arm wave (dashed) are time delayed. (c) Interference signal frequency
proportional to AL . (d) Fourier transform of beat signal measures AL.
9
Although SD-OCT was arguably the main-stream technology of OCT imaging in the late
1990s and early 2000s, SS-OCT has lately risen as a superior technology with faster speed
and longer imaging depth, thanks to the recent advancement of swept-laser sources [22].
While the imaging depth of SD-OCT systems depends on the spectrometer resolution and the
detector cell width, the imaging depth of SS-OCT systems is related to the frequency
linewidth of the swept-laser source. Since the linewidth of currently available swept lasers
can be much narrower than a typical spectrometer resolution, SS-OCT systems generally
offer a much longer imaging depth. Furthermore, SS-OCT has a superior sensitivity because
it is free from spectrometer loss and the quantum efficiency of a photoreceiver is
fundamentally higher than a CCD or CMOS array.
Despite the advantages of SS-OCT over SD-OCT have been theoretically recognized by
the researchers for a long time, the lack of high-performance, low-cost swept lasers have
limited the development of SS-OCT systems. SS-OCT has become a more realistic option
only recently, due to the progress on the development of swept lasers.
1.1.4 Signal Detection in Swept Source OCT
Recent development of new swept source options has offered record scanning speed [23, 24]
and record imaging range [25, 26] to SS-OCT, thereby making the signal detection stage
become the limiting factor of the imaging performance. With a given digital sampling rate,
the swept source operation can be optimized for greater imaging depth, higher axial
resolution, or faster scanning speed. Since the tradeoff point of the three variables are
determined by the sampling rate, the importance of the signal detection stage including the
photoreceiver and the analog-to-digital converter (ADC) has increased with advancement of
wavelength-swept lasers. Currently available ADC options support up to 1 GSPS (109
samples per second), which sets the bandwidth requirement of the photoreceivers to 500MHz
10
in order to completely utilize the ADC sampling rate. Photoreceivers must be designed
carefully to convert the optical signal into an electrical signal as cleanly as possiblecontamination factors such as noise, higher harmonics, and parasitic reflection must be
minimized. The electronics of photoreceivers will be addressed in detail later.
1.2 Balanced Receivers for Swept Source OCT
Photodetection is an essential part of any optical imaging technique because the optical signal
needs to be somehow converted first into analog electric signal and then finally into digital
data in order to be processed by software. Analog-to-digital conversion is a common process
widely used in other fields of electrical engineering; there are a number of reliable ADC units
available in the market. However, the options of photoreceivers for SS-OCT are fairly limited
since the amplification gain must be high enough to scale small signals scattered by relatively
less reflective biological tissues, while the frequency bandwidth should also be high to match
the imaging speed of current cutting-edge OCT technology. This section introduces the
structure of dual-balanced photoreceivers and several measureable factors that evaluate the
electronic performance of the receiver.
1.2.1 Principles of Photodiode Operation
Photodiodes are placed at the very first and essential part of photodetection. An ideal
photodiode will generate a photocurrent perfectly proportional to the incident power when
properly operated with reverse bias. Photodiodes are structurally similar to regular
semiconductor diodes except they may be packaged with a window or optical fiber coupling
to allow light to reach the semiconductor junction. The spectral response of the photodiode
may be adjusted by controlling the semiconductor layer thickness or doping concentration.
Figure 5 shows a cross-section of a PN type photodiode. When the light reaches the PN
junction, electrons are pulled up onto the excited state if the photon energy is greater than
11
Cathode
Light
P-layer
Depletion layer
Figure 5. Diagrammatic description of a PN photo diode. Light with shorter wavelength tend to
excite the electrons in the P-layer and light with longer wavelength tend to penetrate deeper into
the layer and excite the electrons in the N-layer. Electric field in the depletion layer moves the
holes in the N-layer and the electrons to the P-layer.
band gap energy E9. This generates electron-hole pairs throughout the doped layers. In the
depletion layer between the P-layer and the N-layer, electric field accelerates electrons
towards the N-layer and holes towards the P-layer, resulting in a net charge flow. If the light
intensity is higher, a greater number of photons will be colliding to the PN junction thereby
generating proportionally greater number of electron-hole pairs per unit time. The
proportional constant between the incident power and photocurrent is defined as the
responsivity, or photosensitivity, expressed in Amperes per Watt (A/W). Photodiodes with
larger junction area has greater responsivity since a greater portion of the incident power will
be collected by the semiconductor junction. Modem photodiodes such as PIN photodiodes
and avalanche photodiodes have different types of semiconductor junction to achieve faster
response or greater current gain.
Real world photodiodes have several factors that might contaminate the signal. The
junction of the photodiode always has a finite effective capacitance which may result in an
amplification resonance when an op-amp is introduced in the photoreceiver circuit. Moreover,
photodiodes working under reverse-bias voltage generally has a leakage current named as
12
dark current even when there is no incident light. The main source of dark current is the
random generation of electron-hole pairs by the strong electric field inside the depletion layer.
The dark current generates a fix-pattern noise which can be removed by background
subtraction, but the shot noise associated to the dark current still creates temporal noise.
1.2.2 Dual-Balanced Detection
Typical photoreceivers that operate in RF range are single-ended: the intensity is converted
into electric current by single photodiode, and then converted into voltage by a
transimpedance amplifier, or sometimes merely by passive circuit elements. Figure 3 shows
examples of schematic circuit design of a single-ended photoreceiver. Electric signals
acquired by single-ended detection suffer from the noise generated by the light source and the
interferometer system as well as the receiver electronics:
out =S+uncorr
The uncorrelated noise
nuncorr
(6)
+ncorr
includes inevitable noise factors [27] such as shot noise, excess
noise, and receiver thermal noise, while the correlated noise no, includes noise that
commonly exists in the two channels such as the intensity fluctuation of the light source.
ZR
Swept
Source
Reference
50/50
Coupler
50/50 Coupler
Dual-Balanced Detector
Sample
Figure 6. Typical SS-OCT setup for dual-balanced detection. Optical circulators are
introduced to extract complementary outputs.
13
For cases such as interferometry where complementary signals are available, a technique
known as dual-balanced detection [28, 29] is applicable: if optical circulators are introduced
as shown in Figure 6, complementary signals can be obtained at the two output ports,
whereas one signal is directed back to the source in typical Michelson configuration. Dualbalanced photoreceivers take the two complementary signals and amplifies their difference
into electric voltage. There are two main advantages of dual-balanced detection over singleended detection. The primary advantage is the suppression of the correlated noise by the
subtraction of the two input signals:
V.,= S + nuncorr + ncorr
=
--
S-
uncorr
+ ncorr
(7)
2(S + nuncor)
The correlated noise should be completely removed in theory if the power levels are exactly
matched. Furthermore, the amplification gain increases by a factor of 2, since the oscillatory
signals are both present in the two inputs with different signs. This decreases the requirement
of the reference arm power, thereby reducing the uncorrelated noise power relative to the
signal power as well. Abbas and Chan [29] theoretically explained the improved noise
performance
of dual-balanced receivers
as compared to single-ended receivers
and
demonstrated the improvement with experimental results.
Most SS-OCT systems at MIT use dual-balanced photoreceiver products provided by
Thorlabs, inc. and recent prototypes developed by Thorlabs engineers. Since the prototype
receivers usually come without a detailed performance evaluation data, the need for
systematic RF test methods for Thorlabs photoreceivers arose and motivated this project. The
following sections define and briefly describe the electronic performance characteristics such
as gain, bandwidth, and harmonic distortion.
14
1.2.2 Gain and Bandwidth
Gain and bandwidth are the two most representative characteristics of any analog amplifier.
For photoreceivers, the amount of amplification is represented by either the overall
conversion gain or the transimpedance gain defined as follows:
-
(output voltage amplitude)
(optical power amplitude)
(8)
(output voltage)
(photocurrent)
The responsiveness of the receivers to a rapidly-varying signal is quantified by the bandwidth,
the difference between the receiver's cutoff frequencies, where the amplitude frequency
response falls down to half of its maximum. If power frequency response in decibels is given,
the cutoff frequencies correspond to 6dB decrease from the maximum response. For
photoreceivers and signal amplifiers, the lower end of the frequency response is usually close
to DC; therefore, the term bandwidth refers to the cutoff frequency itself.
In the process of designing analog electronics, one of gain and bandwidth is traded off to
achieve the other. It is well known that for single-pole operational amplifiers, the product of
gain and bandwidth is almost constant [30]. The designing goal of the photoreceiver for SS-
Figure 7. A schematic diagram of a primitive transimpedance amplifier.
15
OCT will be therefore to maximize the amplification gain while maintaining the bandwidth
over 500MHz.
1.2.3 Harmonic Distortion
The amplification stage can introduce additional noise or harmonic distortion to the output
signal. The noise is introduced by various designing factors such as improper signal paths and
insufficient power supply noise filtering and thus can be mostly reduced by careful placement
of electronics and simple noise filtering techniques such as bypassing and decoupling [31].
Meanwhile, the harmonic distortion is mainly due to the nonlinear characteristic of the
amplifiers near saturation and thus can be minimized only by appropriate choice of the
amplifier model or special sampling techniques such as automatic zeroing [32] and correlated
double sampling [33].
The nonlinearity of the amplifier results in an unwanted modification of the harmonic
contents of the signal, thereby creating higher-order harmonics in the frequency domain.
Figure 8 illustrates how higher-order harmonic components are generated by a nonlinear
transfer function. In addition, some of the higher-order image artifacts may appear reversed
due to digital aliasing since the maximum frequency bandwidth of the data acquisition card is
\J
Time
Time
out,
Output Signal
Input Signal
Nonlinear
Transfer Function
Freauencv ~ Depth
Figure 8. Nonlinear distortion generating higher order harmonics.
16
limited by the Nyquist condition. For example, if the position of the primary image is deeper
than a half of the Nyquist limit, the reversed secondary image will be overlaid on the primary
image. In most cases, this effect is prevented since ADC has internal anti-aliasing filter.
Nonlinearities should be carefully avoided in SS-OCT because harmonic distortion
appears as higher-order-image artifacts occurring at integer multiples of original image depth.
The artifacts are especially visible for high-reflectivity layers such as the retinal nerve fiber
layer (RNFL) and retinal pigment epithelium (RPE) since harmonic distortion is generally
more severe for larger signals.These artifacts can be mistaken as real structures by clinicians,
thereby introducing a risk of diagnostic errors.
The amount of nonlinear harmonic distortion is measured by total harmonic distortion
(THD) defined as the ratio of the power in the higher harmonics to the power in the
fundamental frequency component:
THD =
"
P,,
fund
(dBc)~
Pg
fn d
4
(9)
where Pnd is the power carried by the fundamental and P denotes the power carried by the
nth harmonic. The unit for THD is dBc, decibels relative to the carrier. In general, only first
three harmonic terms are included because harmonics of order higher than four are negligible.
2 Measurement Setup
In this chapter, elements in the test setup used for RF evaluation of Thorlabs balanced
photoreceivers
are introduced. The setup can be divided into two parts: laser diode
modulation and RF analysis. Laser diode modulation setup generates intensity modulated at
radio frequency, the test input signal for the photoreceiver. The driving voltage of the laser
diode is an RF input signal DC-biased by a typical laser diode driver. The output of the
17
photoreceiver is then plugged into RF analysis equipment to measure quantities such as gain,
bandwidth, and THD. The following sections describe the setup in detail.
2.1 Laser Diode Modulation
2.1.1 Semiconductor Laser Diodes
Semiconductor laser diodes are a cost-effective option for optical signal generation. A bias
current applied to diode creates optical gain by inducing recombination of electrons and holes.
Since the lasing power linearly increases with respect to the bias current when the bias
exceeds the lasing threshold, arbitrary intensity signal can be emitted from the laser by
adding an AC component to a certain diode current over the threshold. Typical operation
curve of a semiconductor laser diode is shown in Figure 9. In our measurement, we feed a
DC-biased constant frequency signal into the diode to measure the frequency response of the
photoreceiver.
The diodes used for our measurement are 131 Onm InGaAsP diodes widely applied in fiber
communication. In our first round of measurements, Thorlabs 2.5mW laser diode LPS-1310FC was used to generate the test input signal. As described later, the Thorlabs diode turned
out to have severe parasitic resonance around 600MHz which obstructs the measurement.
P
AP
nslope
Al--j
Figure 9. Typical operation curve of a laser diode. The diode starts lasing
when the population inversion exceeds the threshold.
18
Laser Diode Modulation Frequency Response
_
S
-2
Thodabs
LPS-1310-FC
OLD344-F4-AFC
Optocom
C
U-1
0
2W
4W
600
SM
10D
12W 14M
160
18M
2[M
Freuqency (MHz)
Figure 10. Exterior appearance of the laser diodes and the measured frequency response of the
modulation setup. The Optocom laser diode (b) clearly has less parasitic fluctuation than the
Thorlabs laser diode (a) at frequencies higher than 700MHz.
Hence, we decided to purchase Optocom OLD3448-F4-AFC, a 2mW high-speed laser diode
for optical communication. Since the new laser diode has less parasitic resonance, we expect
that the measurements performed with it to show increased precision. Figure 10 shows the
exterior and the frequency response of the two laser diodes.
2.1.2 High-Frequency Sine Generator
Our evaluation of photoreceivers requires test signal frequency ranging from 10MHz to
2GHz. Since typical function generators for electronics testing cannot generate such high
frequency, we have a high-frequency sign wave generator Agilent N5181A MXG which
generates up to 3GHz designated for RF electronics testing. The signal generator supports
000 000 00
e
T0
a
Figure 11. The display screen of the Agilent N5181A MXG which can generate high-frequency
sine waves up to 3GHz and low-pass filters for removing harmonics.
19
amplitudes from IpV (-1 OdBm) to 1.4V (13dBm), which turned out to be well suited to the
input range of the laser diode.
The Agilent signal generator, however, does not have excellent harmonic performance.
The output signal includes harmonic distortion as large as -30dBc, which is well beyond the
standard acceptable harmonic distortion of -40dBc for SS-OCT (ref). If the test signal already
includes some harmonic distortion, it is unable to distinguish the photoreceiver's linear
response to the harmonics from the signal generator from the nonlinear response of the
photoreceiver. In order to perform a more accurate measurement of harmonic distortion, a set
of low-pass filters were introduced to remove the harmonics introduced by the signal
generator. Different values of cutoff frequencies are required to measure harmonic distortion
over a wide range of frequency because the second harmonic must be suppressed well while
the fundamental is passed. Four different low-pass filters whose cutoff frequencies are at
200MHz, 350MHz, 450MHz, and 600MHz were used in our measurement.
2.1.3 DC-Biasing
The RF output of the signal generator needs to be combined with a DC bias so that the laser
diode bias current is always above threshold. The DC current is provided by a typical laser
DC
3
RF
2 RF &DC
IJ
L
(b)
(a)
Figure 12. The schematic circuit diagram (a) and the external packaging (b) of the bias-tee.
20
diode controller (Thorlabs LDC 210) and combined with the RF signal by a passive threeport component known as the bias-tee. Conceptually, the bias-tee can be viewed as a
combination of a capacitor that allows AC but blocks the DC bias and an ideal inductor that
blocks AC but allows DC. The schematic diagram and the exterior appearance of a bias-tee is
shown in Figure 12. Since the bias-tee is a passive circuit element, the input and the output
can be arbitrarily chosen. For example, a bias-tee can be used either to break up a DC-biased
RF signal into DC and RF components or to combine a DC and RF signals into a DC-biased
RF signal.
In our measurement setup, a high-quality bias-tee that allows RF signals up to 4GHz
(Mini-Circuits ZFBT-4R2GW) is used. The output of the signal generator is plugged into the
RF port, and the laser diode controller output is plugged into the DC port, in order to get the
DC-biased RF output at the RF+DC port. The output is directly connected to the laser diode
to generate the intensity signal.
2.2 RF Signal Analyzers
In order to analyze frequency-dependent characteristics of the photoreceivers, two different
kind of RF instruments, the network analyzer and the spectrum analyzer, are introduced. The
two instruments have different capabilities and thus different applications. Basic functions of
the analyzers are described here.
2.2.1 Network Analyzer
The network analyzer is an electrical instrument that measures frequenct-dependent scattering
parameters of an electrical circuit. Most network analyzers operate at high frequencies, from
10kHz to 100GHz. The most distinctive feature that separates network analyzers from
spectrum analyzers is the signal generator included in the instrument. The frequency response
21
is measured by finding the complex ratio of the output signal of the circuit to the reference
signal, while the frequency is swept over certain range. In most cases, the RF signal
generated by the network analyzer is split into two wires, and one wire is directly fed back to
the reference input port. There are two types of network analyzers, vector network analyzer
and scalar network analyzer, where vector network analyzers comprise the majority. In vector
network analyzers, both magnitude and phase of the frequency response can be monitored,
while only magnitude can be obtained by scalar network analyzers.
In our measurement setup, the output of the network analyzer is fed into the RF port of the
bias-tee to modulate the laser diode current. The laser diode effectively converts the input
electric signal into light intensity. Then, the laser diode output is converted back to an electric
signal by the photoreceiver and the output is recorded by the network analyzer. Since a
substantial amount of DC voltage can damage the input port of the network analyzer, another
bias-tee is introduced at the photoreceiver output in order to filter out the DC component.
While linear system characteristics such as DC gain, bandwidth, and phase delay can be
measured with the vector network analyzer, it does not have the capability to measure
Bias-Tee
RDc'
Network Analyzer
OUT
R
DC
L_
DCLaser
Laser
Diode
90:10
Coupler
Driver (DC)RE
Detector
------Bias-Tee
(dualbalanced or
single-DCC
ended)
RF+
RF
IN
Figure 13. Schmatic diagram of a gain and bandwidth measurement
setup using the network analyzer.
22
nonlinear characteristics such as harmonic distortion because the input and the output
frequencies of the analyzer are synchronized. Moreover, in our experimental setup, the
available network analyzer (Agilent 4395A) only provided lOkHz to 500MHz frequency
range, which is insufficient for evaluation of recent prototypes whose bandwidth reaches over
1GHz. For nonlinear measurements and high-frequency measurements, we used a different
instrument known as the spectrum analyzer.
2.2.2. Spectrum Analyzer
The spectrum analyzer measures the magnitude of the input signal with respect to the
frequency. The key difference between the spectrum analyzer and the network analyzer is
that the spectrum analyzer does not have reference input, and thus does not have phase
measurement capabilities as well. However, since the entire frequency domain of the signal is
monitored, properties such as noise density and harmonic distortion can be measured.
A sweep-tuned spectrum analyzer measures the magnitude spectrum by downconverting
the input signal so that a certain portion of the spectrum is aligned at the center frequency of a
band-pass filter. The downconverting sine wave is generated by a voltage-controlled
Bias-Tee
Sga
DC
LaserDriver
Diode
90:10
Couoler
Laser
(DC)
(dual-
E
balanced
or singleended)
BisTeAnalyzer
RF+
RF
DC D
D
Figure 14. Schmatic diagram of a gain and bandwidth measurement
setup using the spectrum analyzer.
23
oscillator, enabling the frequency to sweep over a continuous range. Different portion of the
spectrum is passed by the band-pass filter for different downconverting frequency.
One important parameter in this process is the resolution bandwidth, which refers to the
bandwidth of the band-pass filter. As the name implies, the resolution bandwidth determines
the frequency resolution of the spectrum-lower resolution bandwidth allows for the
discrimination of two closely spaced frequency components. The resolution bandwidth also
accounts for the noise floor because broader band-pass filter allows more frequency
components of noise. Therefore, the noise level in the signal is often represented by the noise
density defined as
N
Af,
(10)
(W/Hz)
W
where P ,, is the power associated with the noise and Afe
denotes the resolution bandwidth.
Moreover, there is a tradeoff between the frequency resolution and how fast the analyzer
display can update the full frequency range under consideration. This tradeoff can be
described by the following relation between the sweep time ts, the resolution bandwidth,
and the frequency sweep range span Afspn :
Afpan
(Afes)
2
t,
~
""2
-(11)
Another important bandwidth parameter is the video bandwidth which determines the
bandwidth of the low-pass filter that removes noise in the measured spectrum before it gets
displayed on the analyzer screen. Note that this low-pass filter is not a standard linear filter
because it filters high-frequency noise in the Fourier domain. Usually, the video bandwidth is
set to be equal to the resolution bandwidth for optimal display of the spectrum. If the video
bandwidth Afid is lower than the resolution bandwidth, the sweep time is given by
24
t
span
(12)
sw es fid
The spectrum analyzer is used for measurements of two different quantities in our setup:
the frequency response up to 2GHz and the harmonic distortion. While the signal generator
mentioned in 2.1.2 inputs a sine wave to the laser diode, the spectrum analyzer measures the
power spectrum peak at the fundamental frequency and the higher harmonic frequencies. The
fundamental peak power is measured with respect to the frequency for frequency response
measurement, whereas the second to fourth harmonic peak power is recorded as well as the
fundamental peak power in harmonic distortion measurement. The spectrum analyzer models
used in our setup are Rohde & Schwarz FSEA which spans from 20kHz to 3.5GHz and
Agilent/HP 8594E which allows frequency range between 9kHz and 2.9GHz.
Figure 15. The Agilent/HP 8594E spectrum analyzer screen during a measurement. Resolution
bandwidth and video bandwidth are set to 10kHz, while the frequency span is 1MHz.
2.3 Automation
Measurements with the network analyzer are easy in a sense that all data points are verified in
one sweep. However, the spectrum analyzer must repeatedly sweep to update the spectrum
for each sampling point. Monitoring the spectrum and recording the data by eyeballing for all
25
sampling points
is
clearly time-consuming
and
ineffective approach.
Utilizing the
programming function of the RF signal generator and the spectrum analyzer, an automated
series of measurement can be performed with minimal amount of manual adjustment. In this
section, some details of the automation method using the National Instruments VISA
framework and the GPIB-to-USB interface are discussed.
2.3.1 NI VISA Programming
A majority of electronic instruments support a framework known as National Instruments
Virtual Instrument Software Architecture (VISA) that allows the user to configure, program,
and troubleshoot instrumentation systems which may include various types of interfaces such
as IEEE-488, VXI, PXI, Serial, Ethernet, and USB. VISA provides a programming interface
that links the hardware instrumentation to the development environments such as LabVIEW,
LabWindows/CVI, Measurement Studio for Microsoft Visual Studio, and MATLAB.
In MATLAB VISA programming, the user must create instrument objects first to
communicate with the instruments. Once the objects are defined, interacting with the
instruments involves three primary actions: read, write, and query. Reading is to simply copy
the data from the instrument's buffer to the computer and writing is sending a command to
the instrument to prepare certain data or take certain action, while making a query is simply
writing and reading with a short time interval in between. All messages are encoded in ASCII
text. The complete list of VISA commands used in the measurements is included in the
Appendices as well as the full MATLAB code.
2.3.2 GPIB-to-USB Interface
Although most of the automated test instruments which are recently manufactured primarily
support Universal Serial Bus (USB), instruments designed before the invention of USB only
support IEEE-488, an older type of digital bus more commonly called as General Purpose
26
Interface Bus (GPIB). For our case, both spectrum analyzer models only supports GPIB,
while the Agilent signal generator supports USB. Since most currently available laptop
computers do not have a GPIB port, an interface that converts GPIB signal into USB format
or vice versa became necessary. GPIB-to-USB or USB-to-GPIB conversion is not a simple
physical connector change. The converter interface needs to re-register the digital signal
because GPIB has 24 parallel pins while USB has only four pins.
Fortunately, there was a National Instruments GPIB-to-USB interface already existing in
the lab. The NI GPIB-USB-HS can handle data transfer rates up to 1.8MB/s for standard
IEEE-488.1 bus and 7.7MB/s for high-speed IEEE-488.1 bus, both of which surpasses well
beyond our requirements. There was no technical problem with using the interface because
the interface automatically detected the instruments.
Figure 16. Natiuonal Instruments GPIB-USB-HS connected to the spectrum analyzer.
3 System Calibration
In the previous chapter, various types of circuit elements and instruments involved in the
measurement were described. This chapter discusses on how to calibrate the instruments in
order to make accurate measurements under the existence of parasitic circuit elements and
how to convert the power levels of the electrical test signal to the power of an optical signal.
27
3.1 Parasitic Element Calibration
3.1.1 Laser Diode Parasitic
In any type of electronics testing, there exists a certain degree of parasitic resonance
introduced by various causes such as the capacitance of the pads and the traces on the circuit
board. In our setup, the parasitic capacitance of the laser diode package was a dominant
problem. The declining and fluctuating behavior Since the parasitic effect can be modeled as
a combination of passive circuit elements such as resistors, capacitors and inductors, it can be
represented by a linear transfer function. Thus, the overall transfer function Hff that converts
the signal generator output Vg into the photoreceiver output V.,. can be considered as the
cascade of three transfer functions as shown below:
V,((j)
(13)
= He (Ijw)V(g ()
Hff = H. HldH
Where H,,c ,
Hld
, and Hpa, represent the photoreceiver, the laser diode operation curve, and
the parasitics in the laser modulation stage, respectively. Figure 17 illustrates in detail how
the signal is converted at each step of our photoreceiver evaluation setup. Note that only AC
components are considered because the DC components are filtered by the bias-tee at the
output. Assuming that the laser diode operation curve is perfectly linear,
Hid (jo)
can be
replaced by the slope efficiency n1d = Ad / Ald
When parasitics elements are present, the receiver frequency response H,,. (jO) gets
shaped by the laser diode parasitic response Hp,,,(jco). Such shaping of the frequency
response curve introduces substantial amount of error and must be calibrated for an accurate
measurement. Most dominant parasitic effect in our setup is the low-pass filtering by the
parasitic capacitance of the Thorlabs laser diode, resulting in the attenuation of the frequency
28
Pl= HisH,,V,
+}
V,
Parasitics
LD
=n
par,,Vg
Fiber Coupled
PD
Transinpedance
Amplification
c= H.H H,, V
-
Vi
(DC)
= HrecndHVg
Photoreceiver
Figure 17. Schmatic model of laser modulation and photodetection. LD denotes the laser diode
and PD denotes the photodiode inside the photoreceiver. Variables represent AC components only.
response in higher frequencies. Without proper calibration, this effect leads to a significant
decrease in measured bandwidth. Calibration is even more important for harmonic distortion,
because the harmonic distortion depends on both the frequency and the amplitude of the
photoreceiver input. Calibration methods are described in detail in the next section.
Moreover, a combination of parasitic capacitance, resistance and inductance can create a
very sharp parasitic resonance, which may appear constantly as a sharp dip in all
measurements. This is a serious problem cannot be solved by calibration because
measurement is impossible at the sharp dip. In our setup with the Thorlabs LPS-1310-FC,
there is a - 10dB dip placed in between 550 and 650MHz, while the precise location of the dip
is sensitive to the geometrical alignment of the elements. Although the reason for the
inconsistent location of the dip is unclear, we infer that the resonance is slightly affected by a
small change in the electrical connections or the grounding.
29
3.1.2 Quasi-Calibration Using Fast Single-Ended Receiver
Since the parasitics are inseparable from the elements, perfect calibration of the system is
impossible. Instead, we can perform an approximate calibration with a fast photoreceiver
known to have an even frequency response throughout the frequency range of our interest.
Single-ended, biased receivers suit particularly well for this purpose because they have large
bandwidth due to the absence of an amplification stage. This quasi-calibration method
measures the effect of parasitic elements by having the receiver transfer function Hec (jo) to
be nearly constant. Let us consider the case when we have an ideal single-ended detector with
known flat amplitude frequency response IH,
(i)I = G
and want to perform a calibrated
measurement of the amplitude frequency response IH,,. (jo)| = A(a)) of a dual-balanced
receiver. In a calibrated measurement of amplitude frequency response, the overall amplitude
transfer function is recorded first with the single-ended receiver:
Heff single( CO) =
(14)
Gnld H,,, (jm) .
Next, the measurement is repeated for the dual-balanced receiver, yielding
H effbalanced(01 =)Ao)n ld
(15)
Hpa,(CO).*
Then, calibrated amplitude frequency response of the dual-balanced receiver can be
calculated by finding the ratio of the two overall transfer functions,
Heff ,balanced
IHeff.single
A(]co)
.()1
(j)A(j)=
Heff,balanced (co)(
G
)
.Heffsingle
(16)
We can refer to the specifications of the single-ended receiver to find an approximate value
of the single-ended receiver conversion gain G. A method for measuring G more accurately is
presented in section 3.2.2. If the amplitude response A(co) is given, the conversion gain and
the bandwidth of the dual-balanced detector can be determined by the following:
30
BW(
Gbalanced =-iA
lim A(c)
4
2
(7
The single-ended detector used for our setup is Electro-Optics Technology (EOT) ET3500F which promises fairly even amplitude frequency response up to 10GHz. The slope
efficiency of the photodiode for 1310nm light is 0.86A/W, which gives the conversion gain
of G = 43V / W for 500 impedance matching environment. The amplitude frequency
responses of several dual-balanced detectors before and after calibration are featured in
chapter 4.
3.2 Input Power Calibration
3.2.1 Calibrated Input Voltage Profile
Nonlinear properties such as harmonic distortion depend on the amplitude of the input signal
as well as the frequency. Therefore, in order to monitor the harmonic distortion with respect
to both the input power and the frequency, the amplitude of the optical signal going into the
photoreceiver must be precisely controlled. In our measurement, we modulated the laser
diode with different input signal amplitude for different frequencies to achieve constant
optical power amplitude at the laser diode output.
Since the purpose of our measurement is to test the performance of the photoreceivers for
SS-OCT systems, the definite value of the optical power amplitude must be specified for
comparison with a real OCT signal. The testing range of the optical power amplitude should
be similar to the interference fringe amplitude. As shown in equation (1), if the light coming
P
2
1
back from the reference arm has power PR = ce|E I RR = -" RR and the light scattered back
8
1
from the sample arm has power Ps =- ce
8
0
4
2
31
R =
P
-
4
Rs, the fringe amplitude is given by
4
cc E2
RRRs = 0
2
RRRs = 2 PRPS [27]. Typical fringe amplitude for OCT imaging
applications ranges from about 1pW to 50pW, while ophthalmic imaging systems tends to
have smaller signal since the illumination power is limited stricter than for endoscopic
imaging systems for eye safety issues. In our measurement, harmonic distortion was
measured for power amplitude values of 10, 20, 30, 40, and 50gW.
As briefly discussed in section 2.1.2, a handful of low-pass filters are connected to the
signal generator to remove harmonic contents in the input signal for accurate measurement of
harmonic distortion. Therefore, in order to calibrate the input power for constant optical
power amplitude, we need to consider low-pass filter insertion loss in addition to parasitic
loss. It is helpful to record a calibrated input voltage profile which gives an appropriate input
voltage level to generate constant optical power amplitude at a given frequency. For a desired
value of optical power amplitude Pd, the calibrated input voltage profile can be calculated as
V"'"'(
P
)= nI|H,,.(j w)) L,,
(o)
=i-
H
single
GP
Id
(1j)
L, (O)
(18)
where L1lf (co) <1 is a function of frequency representing the insertion loss of the low-pass
filter. Lf can include information about multiple low-pass filters if the filter is replaced with
another one with different cutoff frequency in the middle of a measurement session. The
product H
(jw) L 1 f,(w) can be measured at once by recording the raw frequency
response of the single-ended receiver with the low-pass filter placed at the signal generator
output; alternatively, one can separately measure the low-pass filter loss and multiply it with
the raw frequency response of the single-ended receiver originally used for parasitic element
calibration.
32
As mentioned in 3.1.2, an approximate value of the conversion gain G of the single-ended
receiver can be calculated based on the data given by the specifications sheet. However, the
specified gain value is a rough estimate and the actual product can have small fabrication
errors. For precise calibration of parasitic effect and input voltage profile, G must be
independently measured for our receiver unit. The following section describes the method of
finding the value of G.
3.2.2 Determination of Single-Ended Receiver Gain
The greatest difficulty in measuring the conversion gain of the single-ended detector is the
lack of an instrument that can measure optical power modulation amplitude precisely. If we
can measure the modulation amplitude P or set it to a certain known value by controlling the
input voltage, the conversion gain can be simply calculated as
G
Vcsingle
(19)
PId
where V.esi,,,
and
Pd
are the amplitudes of the single-ended photodetector output and the
laser diode output. Typical optical powermeters display time-averaged power so they are
incapable of measuring power modulation. However, although direct measurement of the
modulation amplitude is impossible, there is an indirect method for setting the amplitude to a
known value. Since the laser diode stops lasing when the diode current is lower than the
threshold, nonlinear clipping occurs at the bottom end of the output signal if the modulation
amplitude is larger than the DC term of the optical power. This results in a rapid increase of
the higher harmonics with increasing modulation amplitude. Using this effect, we can set the
modulation amplitude to the desired value in the following steps:
1. While monitoring the laser diode output power with a powermeter, find the DC diode
current which sets the output power equal to the desired modulation amplitude
33
Pd .
2. Keep the DC diode current at the same value and connect the laser diode output to the
single-ended receiver. Slowly increase the RF modulation amplitude while watching
the second harmonic of the receiver on the spectrum analyzer screen.
3. Stop increasing the RF amplitude when the second harmonic starts to increase rapidly
and record the RF power of the fundamental displayed on the spectrum analyzer.
4. Convert the RF power on dBm scale to the voltage amplitude on linear scale to
calculate V,,e,,n,. The voltage must be doubled, considering the 3dB attenuation
effect of 502 impedance matching.
The experimental value of G is calculated for each laser diode because the diodes have
different spectral characteristics. For Thorlabs LPS-1310-FC, the measured values are
Pjd =760pW and V,,,,,g,
ET-3500F
V1,singe
is
= 27.42mV; therefore, the estimated conversion gain of the EOT
G=36.08V/W.
For Optocom
OLD3448-F4-AFC,
Pd = 2 19lW
and
=87.4OmV, which gives G=39.89V/W. The measured values of the conversion
gain are slightly smaller than the theoretical estimate G = 43 V/W calculated in 3.1.2. This
can be caused by small fabrication error, fiber coupling loss, or impedance mismatch at the
receiver output.
ld
Figure 18. Schematic view of nonlinear clipping and clipped photoreceiver output signal seen
by an oscilloscope.
34
4 Test Results
With the experimental setup discussed in chapter 2 and calibration methods described in
chapter 3, we evaluated the RF performance of the Thorlabs dual-balanced photoreceivers
used for SS-OCT systems. The photoreceiver specifications provided by the manufacturer is
listed in Table 1. Since the designer only specified the transimpedance gain, the gain factor
between the photodiode current and the receiver output voltage, the overall conversion gain
was calculated by multiplying the transimpedance gain by photodiode slope efficiency.
Although we expect that the result of our measurement is consistent with the reported
specifications, the reported specifications can be incorrect for the prototype receivers since
they are theoretical estimates given by the designer.
Model Name
PDB460C
PDB470C-SP1
PDB480C-SP1
PDB480C
PDB480C-SP1-D
Transimpedance
Gain (V/A)
30000
20000
6800
10000
20000
Photodiode Slope
Efficiency (A/W)
0.90
0.86
0.86
0.86
0.86
Conversion . Bandwidth
Gain (V/W)
27000
200MHz
17200
330MHz
5400
1.5GHz
8600
1.5GHz
17200
1.5GHz
Table 1. Reported specifications of the photoreceivers. Only transimpedance gain was notified by the designer.
The conversion gain is calculated by multiplying the transimpedance gain by the photodiode slope efficiency.
4.1 Frequency Response
4.1.1 Results with Thorlabs LPC-1310-FC
The first calibrated frequency response measurements up to 2GHz were performed with
Thorlabs LPC-13 10-FC. The gain and bandwidth values presented in Table 2 in comparison
with the reported values were calculated based on equations (16) and (17), while the singleended receiver conversion gain G =36.08 V/W was used. Normalized amplitude responses
of the photoreceivers before and after calibration are shown in Figure 19 and Figure 20.
35
Model Name
Reported Gain
(V/W)
Measured Gain
(V/W)
PSB460C
PDB470C-SP1
PDB480C-SP1
PDB480C
PDB480C-SP1-D
27000
17200
5400
8600
17200
27400
18600
8000
7800
20000
Measured
Bandwidth
230MHz
370MHz
1.77GHz
1.53GHz
1.93GHz
Reported
Bandwidth
200MHz
330MHz
1.5GHz
1.5GHz
1.5GHz
Normalized Frequency Response, Logarbythmic Scale
Normalized Frequency Response
0
2
-1
0
-2
0 -2
0
...
.. ..................
-7
....
......
- --PB460C
---
PDB470C-SPI
100
150 200
250 300
-
-8
(Single-Endei
-- ET-3500F
50
PDB460C
PDB470C-SPI
PD84WC-SPI
---
-8
350 4
450 500
0
Frequency (MHz)
so
1M
150
2
2
350
4"
4W
S
Frequency (MHz)
Figure 19. Frequency response curves measured with LPS-1310-FC and the network analyzer
before and after calibration.
Normalized Calibrated Frequency Response
Normalized Frequency Response
0
-5
0
-5
-10
---.......
-15
0
-10
*-15
"-20
-20
-25
--
DB400C-SPI
.- -PDBJ0C
0
200
400
6W
0
1000 1200 1400
(MHz)
Frequency
- ~--PDB40C-SPI
-25
PDB480C-SPI-D
ET-35MF(Single)
---
Iwo
160
-30
203)
0
200
PDB480C
PDB48DC-SPI-D
400
600
80
1000 1200 1400
Frequency
(MHz)
1600
100 2000
Figure 20. Normalized frequency response measured with LPS-13 10-FC and the spectrum analyzer,
before and after calibration. -10dB parasitic resonance dip at 620MHz is present in the raw
frequency response. Since the resonance is sharp, there is a remnant of the dip after calibration.
Table 2. Gain and bandwidth measured with Thorlabs LPS-1310-FC, in comparison with reported values.
The measured values roughly match the specifications reported by the designer. Again, the
reported values are not necessarily accurate because they are not independently measured but
36
estimated from electronics specifications provided by the manufacturer. It is interesting that
PDB480C-SPI appears to have higher gain than PDB480C, although the reported values
suggest the opposite.
4.1.2 Results with Optocom OLD3448-AFC
The same measurement was repeated with another laser diode, Optocom OLD3448-AFC. As
mentioned in 2.1.1, the original purpose of this laser diode is optical communication and thus
expected to have less parasitic resonance. For conversion gain value, G = 39.89 V/W was
used. The experimental results are listed in Table 3 and the amplitude response plots are
presented in Figure 21 and Figure 22.
Model Name
Reported Gain
(V/W)
Measured Gain
(V/W)
PDB460C
PDB470C-SPI
PDB480C-SPI
PDB480C
PDB480C-SPI-D
27000
17200
5400
8600
17200
26600
17700
8500
7300
19500
Measured
Bandwidth
230MHz
370M11z
1.77GHz
1.53GHz
1.93GHz
Reported
Bandwidth
200MHz
330MHz
1.5GHz
1.5GHz
1.5GHz
Table 3. Gain and bandwidth measured with Optocom OLD3448-AFC, in comparison with reported values.
Normalized Frequency Response, Logarhythmic Scale
Normalized Frequency Response
n.
-2
02 -4
... . ... .
-
S-6
-7
-
PDB460C
PDB47OC-SPI
PDB480C-SPI
(Single-Ended)
ET-350OF
-
-8
0
50
100
150
200
PDB460C
-PDB47OC-SP
PDB480C-SPI
-.-.
250
300 350
4W
U
0
450
50
100
150 200
250
30
350
4W
450
5U0
Frequency (MHz)
Frequency (MHz)
Figure 21. Normalized frequency response curves measured with OLD3448-AFC and the network
analyzer before and after calibration.
37
Normalized Frequency Response
0 --.-- ----- --
Normalized Calibrated Frequency Response
.-.-------.. -- -------.. ---------- --- --- --
---
-
0
----
.----.---.
-.-.-.
- ------.
--------------
. - -.
---...
-...........
---.----
--- -
-35
-40
0
- PDB4BC-SPI
PDB480C
PDB480C-SPI-D
ET-35MF (Single)
200
400
600
1(D
-
-20
- -
...
-
- - -.--
- .
-- - - -
CDA
-
--
.-.
.-
- - - - -- - -
-25 - - - - - - .2-30 -
-
-.-.
5 - --.
...
-..
...
-10 -
---
-
-..--..-.
-
-
.
100 1200
Frequency
(MHz)
-
-
-
--
-30)
14%
1660
16
20'M
U
200
PDB4..-SPI
PDB480C
PDB480C-SPD
400
m
100 1200
Frequency
(MHz)
1400
16M 15
2CE
Figure 22. Normalized frequency response measured with Optocom OLD-3448-F4-AFC and the
spectrum analyzer, before and after calibration. Parasitic resonance dip is -6dB deep and has two local
minima at 660 and 690MHz. Calibration result is much smoother than with the Thorlabs diode.
The results meet with the values measured with the Thorlabs laser diode: the measured gains
fall within ??% error range and the measured bandwidths are nearly identical. The two
measurements agree on that PDB480C-SPI has higher gain than PDB480C does. The
frequency response plots show that the fluctuation noise and the remnant of the parasitic
resonance are less visible with the Optocom laser diode than with the Thorlabs laser diode.
4.2 Harmonic Distortion
Harmonic distortion was measured with Thorlabs LPS-1310-FC as a function of optical
power modulation amplitude ranging from 10 to 50pW and frequency between 130 and
500MHz. The modulation amplitude corresponds to the peak value V, of the RF modulation,
not the peak-to-peak value V,, . As mentioned in section 3.2.1, this amplitude corresponds to
the output fringe amplitude P =
-RR,
2
in a real OCT system. The power P/2 incident to
the sample is often limited by safety criterion such as ANSI standards. The results of
harmonic distortion measurement are shown in the figures below.
38
PDB460C
10sw
-
- -- 20s
-
--
-
-
0 -10 -
40sW
- .
- --. -.
-15 - --.
--
-
-20
E
-30 -
-
.
.... ......-.- -.-..-........
-
-2 5
-5
-
-
-
- .
...--..
.. -
.
-.
... -. -.
10
1
10
- ... -.
--
220
200
240
-
... .
2M
200
Frequency (MHz)
Figure 23. Total harmonic distortion of PDB460C. The harmonics are
very high for low frequency and become lower for frequency around
bandwidth (200MHz).
PDB470C-SPI
-20
-
-
0
0
-0s
-
-304w
0LW
.50
1
20
20,
M
3
4M
3M
4W
Soo
Frequency (MHz)
Figure 24. Total harmonic distortion of PDB470C-SPI. The
harmonics are fairly low for low frequency and grow higher around
bandwidth (330MHz).
PDB48OC-SPI
-20 -
-
-
-
-
- - -.-.-.-.-
-- --.
....
-. -.
......
....-25 -.
80
-.
-
- .... .-
-.-..--
-.-
--.
-30 - ....
-..
-35 - - ..
-
-
-
-.
...
-40
-45
-
-.-.-.-
20pW
lo---30pLw
40ILW
50sW
-9
00
1
20W
250
3W
30
4W
450
Soo
Frequency (MHz)
Figure 25. Total harmonic distortion of PDB480C-SPI. Resonance of
the photodiode capacitance and the amplification circuit generates
random oscillation.
39
PDB480C
0-25
s0gw
--
o
201sLW
.......
-35.............
............
......... .......
Frequency (MHz)
Figure 26. Total harmonic distortion of PDB480C. Harmonics are very
high for low frequencies and dramatically decrease around 170MHz.
PDB480C-SPI-D
19OcOJ 15)
20'
25)
30
35)t
40s
o
4
40
500
- --
soo
Frequency (MHz)
Figure 27. Total harmonic distortion of PDB48oC-SPI-D.
Harmonics are well suppressed in the entire frequency range.
5 Conclusion
Through this project, we established methods for comprehensive testing of post-amplified
dual-balanced photoreceivers, including measurements of amplitude frequency response,
conversion gain, bandwidth, and nonlinear characteristics such as harmonic distortion. In
order to compensate parasitic effect, approximate calibration method using a fast singleended biased photoreceiver was introduced. The calibration was based on a well-structured
mathematical model and the he evaluation results of Thorlabs photoreceivers currently
existing in the lab supports that the model well suits the measurement setup.
40
RF testing instruments such as network analyzer, spectrum analyzer and signal generator
were used. Utilizing the programmability of the instruments, we were able to automate the
measurement via VISA framework and GPIB-to-USB interface. The automation of the
measurement significantly reduced the required time and effort while increasing the accuracy
and the repeatability as well. Considering the acceptable frequency range of the RF
instruments, we expect that this setup will be able to evaluate photoreceivers with larger
bandwidths available in the future.
Acknowledgements
Foremost, I would like to express my sincere gratitude to Professor James G. Fujimoto for
continuously guiding and supporting the project. His guidance and wisdom helped me
throughout the process of developing the ideas and writing the thesis.
I would like to thank my senior colleagues Dr. Benjamin M. Potsaid, Dr. Ireneusz
Grulkowski, Jonathan J. Liu, Woo Jhon Choi, and Chen D. Lu for always being responsive to
my questions. My sincere thanks also goes to Mrs. Dorothy A. Fleischer, the administration
staff at the Laser Medicine and Medical Imaging Group. Her advices and management were
truly helpful whenever it came to ordering equipment.
41
Appendix
A. VISA Command List
Action
Command
Instrument
Agilent N5181A MXG
POW:AMPL # DBM
Set power to # dBm
RF Signal Generator
FREQ # MHZ
Set frequency to # MHz
OUTP:STAT ON
Set the RF output channel on/off
OUTP:STAT OFF
Agilent/HP 8594E
CF #MZ
Set the center frequency to # MHz
Spectrum Analyzer
SP #MZ
Set the sweep span to
TS;
Acquire new spectrum
MKPK
Move the marker to the peak location
MKA?
Ask for the power at the marker
B. MATLAB Code for Automation
fresp.m-measures frequency response at one sampling point
function
tf
if nargin ==
= fresp(src,
4;
spa,
pow,
freq,
navg)
navg = 1; end
]]);
', num2str(pow), ' dBI'
num2str(freq),
' MHz']);
STATV ON' );
fwrite(src,
fwrite(src,
fwrite(src,
['POW: AMPL
fwrite(spa,
fwrite (spa,
pause (0. 5);
['CF ',
' SPE 1.MZ
['FREQ
' OTUTP
',
num2str(freq),'MZ']);
tf = 0;
for cnt = 1:navg
fwrite (spa, 'TS; ')
pause (0.05) ;
fwrite (spa,
'MKPK');
pause (0.05) ;
tf = tf + str2double(query(spa,
end
'MKA'?'))/navg;
42
pause(0.1);
fwrite(src,
'OUTP:STAT
OFF')
end
measfr.m-measures the frequency response by repeating fresp.m
delete(instrfindall);
clear
src spa
src = visa('ni',
spa = visa( 'ni
'USB0::x0957:: x
'GP10: :18::1STR
,
l01::MY50141927 ::INSTR');
fopen(src);
fopen(spa);
fwrite(spa,
'SNGLS;TS;');
pause(0.2);
freq = 2.5:2.5:500;
pow = -15;
navg = 8;
tf
= zeros(1,length(freq));
for cnt = 1:length(freq);
tf(cnt) = fresp(src, spa, pow, freq(cnt),
end
navg);
fclose (src);
fclose (spa);
display(' Measurement
finishe')
delete(instrfindall);
clear
src spa navg
harmdist.m-measures harmonic distortion at one point
function
[thd, hd]
if nargin == 4;
fwrite(src,
fwrite(src,
fwrite(src,
= harmdist(src, spa, pow, freq, navg)
navg = 1;
end
['POW:AMPL ', num2str(pow), ' dBm']);
['FREQ ',
num2str(freq),
' MHz']);
'OUTP:STAT ON'
pfund = 0;
fwrite(spa,
fwrite(spa,
pause(0.5);
['CF ',
num2str(freq),'MZ']);
'SP1 1MZ');
43
for cnt = 1:navg
fwrite(spa,
'TS;)
pause (0.05);
fwrite (spa, 'VMKPIK');
pause (0.05);
pfund = pfund + str2double(query(spa,
end
hd
=
'MKA?'))/navg;
zeros(1,3);
for cntl = 1:3
fwrite(spa,
pause (0.5);
['CF ',
num2str(freq*(cntl+1)),'MZ']);
for cnt2 = 1:navg
fwrite(spa, 'TS;')
pause(0.05);
fwrite(spa, 'MKPK');
pause(0.05);
hd(cntl) = hd(cntl) + str2double(query(spa,
end
if hd(cntl) < -70, hd(cntl ) = -Inf; end
'MKA?'))/navg;
end
pause(0.1);
fwrite(src,
'OUTP:STAT
OFE')
hd = hd-pfund;
thd = pow2db(sum(db2pow(hd)));
end
measharm.m-measures harmonic distortion by repeating harmdist.m
delete(instrfindall);
clear
sr:an
d spa
load normdata.mat
src = visa('ni',
'USB0::0x0:957::0xlF01::MY51
'GP B01:: 18::INST ');
',
spa = visa('ni
0141927
9
:INSTR');
fopen(src);
fopen(spa);
fwrite(spa,
'SNGLS; IS; 'P);
pause(0.2);
freq = 130:10:500;
stopfreq =
[200,
290,
pow = 10:10:50;
sloptical-microwatts
-electrical-200mVpp
420];
= -10dJBm,
40 OmVpp
-4dBm,
navg = 4;
hd = zeros(length(pow),length(freq),3);
44
lVpp = +4dri 2Vpp
+10dm
thd = zeros (length (pow) ,length(freq));
inpow = pow2db( (pow*34/15*le-3) .^2/50*1e3) ;
Jipw
inpo wc al ib (src, pa.,pw, pow-11 C) ;
For
For nomal
n
i ed (opt ical input
normali zeI otputl power-
refresh(src, freq(1));
for cntl = 1:length(freq)
for cnt2 = 1:length(pow)
[thd(cnt2,cntl), hd(cnt2,cntl,:)] = harmdist(src, spa,
inpow(cnt2)-para(freq(cntl)==parafreq)lpfloss(freq(cntl)==lpffreq), freq(cntl), navg);
end
if
sum(frea(cntl) == stopfreq)
display('Please
chanre the low pass f ilter
and press any key')
pause;
refresh(src, freq(cntl+1));
end
end
display('Measurement
finished')
delete(instrfindall);
clear src and spa and nav g
inpowcalib.m-generates calibrated power profile
function inpow = inpowcalib(src,spa,pow,start)
if nargin ==
3,
start = repmat(-10,[I,length(pow)]);
end
inpow = zeros(l,length(pow));
% Power AdjustmenL
for cnt = 1:length(pow)
inpow(cnt) = start(cnt);
outpow = fresp(src,spa,inpow(cnt),1);
assert(outpow > -30,
'Check if
the detector
and the laser
ar~e On') ;
while abs(outpow-pow(cnt))
> 0.005
outpow = fresp(src,spa,inpow(cnt),l);
inpow(cnt) = inpow(cnt) + pow(cnt) - outpow;
end
end
end
lpfcalib.m-for low pass filter insertion loss calibration
delete(instrfindall);
clear
src spa
load normdata.ma.t
src = visa('nL',
'IISBO: :x0957: OxIF01::MY50141927: :INSTR'
1 NSTR' );
spa = visa('ni',
' B0::
P
0
fopen(src);
45
controller
fopen(spa);
fwrite
(spa,
fwrite(spa,
fwrite(spa,
SWE
IME AUTO ON
'BWID:RES 3 kHz');
'BWID:VID I kHz');
'
pause(0.2);
freq = 130:10:500;
stopfreq = [200, 290, 420];
pow = 0;
navg = 16;
tf
= zeros(1,length(freq));
for cnt = 1:length(freq)
tf(cnt)
= fresp(src, spa, pow, freq(cnt),
navg);
if
sum(freq(cnt) == stopfreq)
display('Please
change the low pass filter and Press aiy key'
pause;
end
end
46
References
1.
2.
3.
4.
5.
6.
7.
8.
9.
10.
11.
12.
13.
14.
15.
16.
17.
18.
Huang, D., E.A. Swanson, C.P. Lin, J.S. Schuman, W.G. Stinson, W. Chang, M.R. Hee, T.
Flotte, K. Gregory, C.A. Puliafito, and J.G. Fujimoto, Optical Coherence Tomography.
Science, 1991. 254(5035): p. 1178-1181.
Shimada, M., H.W. Chang, Y. Fujishige, and K. Okuyama, Calibration of polarizationsensitive and dual-angle laser light scatteringmethods using standardlatex particles.Journal
of Colloid and Interface Science, 2001. 241(1): p. 71-80.
Kundert,
U.,
B.
Discourse and Corpus. Konfliktverlaufe:
Normen
Der
Geschlechterbeziehungen in Texten Des 17th Jahrhunderts, 2004. 33: p. 17-35.
Potsaid, B., B. Baumann, D. Huang, S. Barry, A.E. Cable, J.S. Schuman, J.S. Duker, and J.G.
Fujimoto, Ultrahigh speed 1050nm swept source / Fourierdomain OCT retinal and anterior
segment imaging at 100,000 to 400,000 axial scans per second. Optics Express, 2010. 18(19):
p. 20029-20048.
Boppart, S.A., W. Luo, D.L. Marks, and K.W. Singletary, Optical coherence tomography:
feasibilityfor basic research and image-guided surgery of breast cancer. Breast Cancer Res
Treat, 2004. 84(2): p. 85-97.
Mashimo, H., S. Desai, M. Pedrosa, M. Wagh, Y. Chen, P. Herz, P.L. Hsiung, A. Aguirre, A.
Koski, J. Schmitt, and J.G. Fujimoto, Ultrahigh resolution endoscopic optical coherence
tomography: a novel technologyfor gastrointestinalimaging. Gastroenterology, 2005. 128(4):
p. A251-A251.
Zhou, C., D.C. Adler, T.H. Tsai, H.C. Lee, L. Becker, J.M. Schmitt, Q. Huang, J.G. Fujimoto,
and H. Mashimo, Endoscopic 3D-OCT Reveals Buried Glands following Radiofrequency
Ablation of Barrett'sEsophagus. Endoscopic Microscopy V, 2010. 7558.
Takada, K., I. Yokohama, K. Chida, and J. Noda, New measurement systemfor fault location
in optical waveguide devices based on an interferometric technique. Applied Optics, 1987. 26:
p. 1603-1608.
Youngquist, R., S. Carr, and D. Davies, Optical coherence-domain reflectometry: a new
optical evaluation technique. Optics Letters, 1987. 12(3): p. 158-160.
Gilgen, H.H., R.P. Novak, R.P. Salathe, W. Hodel, and P. Beaud, Submillimeter optical
reflectometry. IEEE Journal of Lightwave Technology, 1989. 7: p. 1225-1233.
Fercher, A.F., K. Mengedoht, and W. Werner, Eye-length measurement by interferometry
with partiallycoherent light. Opt Lett, 1988. 13: p. 1867-1869.
Schmitt, J.M., A. Knuttel, M. Yadlowsky, and M.A. Eckhaus, Optical-coherencetomography
of a dense tissue: statistics of attenuation and backscattering. Physics in Medicine and
Biology, 1994. 39(10): p. 1705-20.
Izatt, J.A., M.D. Kulkarni, H.-W. Wang, K. Kobayashi, and M.V. Sivak, Jr., Optical
coherence tomography and microscopy in gastrointestinaltissues. IEEE Journal of Selected
Topics in Quantum Electronics, 1996. 2(4): p. 1017-28.
Strube, H.W., A Generalization of Correlation-Functionsand the Wiener-Khinchin Theorem.
Signal Processing, 1985. 8(1): p. 63-74.
Fercher, A.F., C.K. Hitzenberger, G. Kamp, and S.Y. Elzaiat, Measurement of Intraocular
Distances by BackscatteringSpectral Interferometry. Optics Communications, 1995. 117(1-2):
p. 43-48.
Hausler, G. and M.W. Linduer, "Coherence radar" and "spectral radar"-new tools for
dermatologicaldiagnosis. Journal of Biomedical Optics, 1998. 3(1): p. 21-31.
Wojtkowski, M., A. Kowalczyk, R. Leitgeb, and A.F. Fercher, Full range complex spectral
optical coherence tomography technique in eye imaging. Optics Letters, 2002. 27(16): p.
1415-17.
Chinn, S.R., E.A. Swanson, and J.G. Fujimoto, Optical coherence tomography using a
frequency-tunable optical source. Optics Letters, 1997. 22(5): p. 340-342.
47
19.
20.
21.
22.
23.
24.
25.
26.
27.
28.
29.
30.
31.
32.
33.
Golubovic, B., B.E. Bouma, G.J. Tearney, and J.G. Fujimoto, Optical frequency-domain
reflectometry using rapid wavelength tuning of a Cr4+.forsteritelaser. Optics Letters, 1997.
22(22): p. 1704-1706.
Lexer, F., C.K. Hitzenberger, A.F. Fercher, and M. Kulhavy, Wavelength-tuning
interferometry of intraoculardistances. Applied Optics, 1997. 36(25): p. 6548-6553.
Haberland, U.H.P., V. Blazek, and H.J. Schrnitt, Chirp optical coherence tomography of
layered scattering media. Journal of Biomedical Optics, 1998. 3(3): p. 259-66.
Yun, S.H., G.J. Tearney, J.F. de Boer, N. Iftimia, and B.E. Bouma, High-speed optical
frequency-domain imaging. Optics Express, 2003. 11(22): p. 2953-2963.
Wieser, W., B.R. Biedermann, T. Klein, C.M. Eigenwillig, and R. Huber, Multi-Megahertz
OCT High quality 3D imaging at 20 million A-scans and 4.5 GVoxels per second. Optics
Express, 2010. 18(14): p. 14685-14704.
Klein, T., W. Wieser, C.M. Eigenwillig, B.R. Biedermann, and R. Huber, Megahertz OCTfor
ultrawide-field retinal imaging with a 1050nm Fourier domain mode-locked laser. Optics
Express, 2011. 19(4): p. 3044-3062.
Jayaraman, V., J. Jiang, B. Potsaid, G. Cole, J. Fujimoto, and A. Cable, Design and
performance of broadly tunable, narrow line-width, high repetition rate 1310nm VCSELs for
swept source optical coherence tomography. Vertical-Cavity Surface-Emitting Lasers Xvi,
2012. 8276.
Jayaraman, V., J. Jiang, H. Li, P.J.S. Heim, G.D. Cole, B. Potsaid, J.G. Fujimoto, and A.
Cable, OCT Imaging up to 760 kHz Axial Scan Rate Using Single-Mode 1310nm MEMSTunable VCSELs with > 100nm Tuning Range. 2011 Conference on Lasers and ElectroOptics (Cleo), 2011.
Rollins, A.M. and J.A. Izatt, Optimal interferometer designs for optical coherence
tomography. Optics Letters, 1999. 24(21): p. 1484-6.
Kasper, B.L., C.A. Burrus, J.R. Talman, and K.L. Hall, Balanced Dual-DetectorReceiverfor
Optical Heterodyne Communication at Gbit/S Rates. Electronics Letters, 1986. 22(8): p. 413415.
Abbas, G.L., V.W.S. Chan, and T.K. Yee, A Dual-DetectorOptical Heterodyne Receiverfor
Local OscillatorNoise Suppression. Journal of Lightwave Technology, 1985. 3(5): p. 11101122.
Faulkner, E.A. and Grimbleb.Jb, Active Filters and Gain-Bandwidth Product. Electronics
Letters, 1970. 6(17): p. 549-&.
Persson, E. Power electronic design and layout techniques for improved performance and
reduced EML in Power Electronics in Transportation,1994. [Proceedings]. 1994.
Enz, C.C. and G.C. Temes, Circuit techniques for reducing the effects of op-amp
imperfections: Autozeroing, correlated double sampling, and chopper stabilization.
Proceedings of the Ieee, 1996. 84(11): p. 1584-1614.
Huang, Y.T., G.C. Temes, and P.F. Ferguson, Reduced nonlinear distortion in circuits with
correlateddouble sampling. Iscas 96: 1996 Ieee International Symposium on Circuits and
Systems - Circuits and Systems Connecting the World, Vol 1, 1996: p. 159-162.
48