0Abut2Pratik Evirici
Using the SG3525 PWM Controller - Explanation and
Example: Circuit Diagram / Schematic of Push-Pull
Converter
PWM is used in all sorts of power control and converter circuits. Some common examples include
motor control, DC-DC converters, DC-AC inverters and lamp dimmers. There are numerous PWM
controllers available that make the use and application of PWM quite easy. One of the most popular of
such controllers is the versatile and ubiquitous SG3525 produced by multiple manufacturers – ST
Microelectronics, Fairchild Semiconductors, On Semiconductors, to name a few.
SG3525 is used extensively in DC-DC converters, DC-AC inverters, home UPS systems, solar
inverters, power supplies, battery chargers and numerous other applications. With proper
understanding, you can soon start using SG3525 yourself in such applications or any other application
really that demands PWM control.
Before going on to the description and application, let’s first take a look at the block diagram and the
pin layout.
Pins 1 (Inverting Input) and 2 (Non Inverting Input) are the inputs to the on-board error amplifier. If you
are wondering what that is, you can think of it as a comparator that controls the increase or decrease
of the duty cycle for the “feedback” that you associate with Pulse Width Modulation (PWM).
This functions either to increase or decrease the duty cycle depending on the voltage levels on the
Inverting and Non-Inverting Inputs – pins 1 and 2 respectively.


When voltage on the Inverting Input (pin 1) is greater than voltage on the Non-Inverting Input
(pin 2), duty cycle is decreased.
When voltage on the Non-Inverting Input (pin 2) is greater than voltage on the Inverting Input
(pin 1), duty cycle is increased.
The frequency of PWM is dependent on the timing capacitance and the timing resistance. The timing
capacitor (CT) is connected between pin 5 and ground. The timing resistor (RT) is connected between
pin 6 and ground. The resistance between pins 5 and 7 (RD) determines the deadtime (and also
slightly affects the frequency).
The frequency is related to RT, CT and RD by the relationship:
With RT and RD in Ω and CT in F, f is in Hz.
Typical values of RD are in the range 10Ω to 47Ω. The range of values usable (as specified by the
manufacturers of SG3525) is 0Ω to 500Ω.
RT must be within the range 2kΩ to 150kΩ. CT must be within the range 1nF (code 102) to 0.2µF
(code 224). The oscillator frequency must be within the range 100Hz to 400kHz. There is a flip-flop
before the driver stage, due to which your output signals will have frequencies half that of the oscillator
frequency that is calculated using the above mentioned formula. So, if you are looking to use this for a
50Hz inverter, you require drive signals of 50Hz. So, the oscillator frequency must be 100Hz.
A capacitance connected between pin 8 and ground provides the soft-start functionality. The larger the
capacitance, the larger the soft-start time. This means that the time taken to go from 0% duty cycle to
the desired duty cycle or maximum duty cycle is larger. So, the duty cycle increases more slowly
initially. Keep in mind that this only affects initial rate of increase of duty cycle, ie, the rate of increase
of duty cycle after the SG3525 starts up.
Typical values of the soft-start capacitance lie within the range 1µF to 22µF depending on the desired
soft-start time.
Pin 16 is the output from the voltage reference section. SG3525 contains an internal voltage reference
module rated at +5.1V that is trimmed to provide a ±1% accuracy. This reference is often used to
provide a reference voltage to the error amplifier for setting the feedback reference voltage. It can be
directly connected to one of the inputs or a voltage divider can be used to further scale down the
voltage.
Pin 15 is VCC – the supply voltage to the SG3525 that makes it run. VCC must lie within the range 8V
to 35V. SG3525 has an under-voltage lockout circuit that prevents operation when VCC is below 8V,
thus preventing erroneous operation or malfunction.
Pin 13 is VC – the supply voltage to the SG3525 driver stage. It is connected to the collectors of the
NPN transistors in the output totem-pole stage. Hence the name VC. VC must lie within the range
4.5V to 35V. The output drive voltage will be one transistor voltage drop below VC. So when driving
Power MOSFETs, VC should be within the range 9V to 18V (as most Power MOSFETs require
minimum 8V to be fully on and have a maximum VGS breakdown voltage of 20V). For driving logic
level MOSFETs, lower VC may be used. Care must be taken to ensure that the maximum VGS
breakdown voltage of the MOSFET is not crossed. Similarly when the SG3525 outputs are fed to
another driver or IGBT, VC must be selected accordingly, keeping in mind the required voltage for the
device being fed or driven. It is common practice to tie VC to VCC when VCC is below 20V.
Pin 12 is the Ground connection and should be connected to the circuit ground. It must share a
common ground with the device it drives.
Pins 11 and 14 are the outputs from which the drive signals are to be taken. They are the outputs of
the SG3525 internal driver stage and can be used to directly drive MOSFETs and IGBTs. They have a
continuous current rating of 100mA and a peak rating of 500mA. When greater current or better drive
is required, a further driver stage using discrete transistors or a dedicated driver stage should be used.
Similarly a driver stage should be used when driving the device causing excessive power dissipation
and heating of SG3525. When driving MOSFETs in a bridge configuration, high-low side drivers or
gate-drive transformers must be used as the SG3525 is designed only for low-side drive.
Pin 10 is shutdown. When this pin is low, PWM is enabled. When this pin is high, the PWM latch is
immediately set. This provides the fastest turn-off signal to the outputs. At the same time the soft-start
capacitor is discharged with a 150µA current source. An alternative method of shutting down the
SG3525 is to pull either pin 8 or pin 9 low. However, this is not as quick as using the shutdown pin. So,
when quick shutdown is required, a high signal must be applied to pin 10. This pin should not be left
floating as it could pick up noise and cause problems. So, this pin is usually held low with a pull-down
resistor.
Pin 9 is compensation. It may be used in conjunction with pin 1 to provide feedback compensation.
Now that we’ve seen the function of each pin, let’s design a circuit with the SG3525 and see how it is
put to use practically.
Let’s make a circuit running at 50kHz, driving MOSFETs (in a push-pull configuration) that drive a
ferrite core which then steps up the high frequency AC and then is rectified and filtered to give a 290V
regulated output DC that can be used to run one or more CFLs.
For the turns calculation, check out my article "Ferrite Transformer Turns Calculation for HighFrequency/SMPS Inverter": http://tahmidmc.blogspot.com/2012/12/ferrite-transformer-turnscalculation.html
So here’s the circuit (click on the circuit to enlarge the image):
Let’s analyze it and see what I’ve done.
You can firstly see that the supply voltage has been provided and ground has been connected. Also
notice that VC has been connected to VCC. I’ve added a bulk and a decoupling capacitor across the
supply pins. The decoupling capacitor (0.1µF) should be placed as close to the SG3525 as possible.
You should always use this in all your designs. Do not omit the bulk capacitor either, although you may
use a smaller value.
Let’s see pins 5, 6 and 7. I’ve added a small resistance RD (between pins 5 and 7) that provides a little
deadtime. I’ve connected RT between pin 6 and ground and CT between pin 5 and ground. RD = 22Ω,
CT = 1nF (Code: 102) and RT = 15kΩ. This gives an oscillator frequency of:
As the oscillator frequency is 94.6kHz, the switching frequency is 0.5 * 94.6kHz = 47.3kHz and this is
close enough to our target frequency of 50kHz. Now if you had needed 50kHz accurate, then the best
way would have been to use a pot (variable resistor) in series with RT and adjust the pot, or to use a
pot (variable resistor) as RT, although I prefer the first as it allows for fine tuning the frequency.
Let’s look at pin 8 now. I’ve connected a 1µF capacitor from pin 8 to ground and this provides a small
soft-start. I’ve avoided using too large a soft-start as the slow duty cycle increase (and thus the slow
increase in voltage) causes problems when using CFLs at the output.
Let’s look at pin 10 now. Initially it’s pulled up to VREF with a pull-up resistor. So, PWM is disabled
and does not run. However, when the switch is on, pin 10 is now at ground and so PWM is enabled.
So, we’ve made use of the SG3525 shutdown option (via pin 10). Thus the switch acts like an on/off
switch.
Pin 2 is connected to VREF and is thus at a potential of +5.1V (±1%). The output of the converter is
connected to pin 1 through a voltage divider with resistances 56kΩ and 1kΩ. Voltage ratio is 57:1. At
feedback “equilibrium”, voltage at pin 1 is 5.1V as well as this is the target of the error amplifier – to
adjust the duty cycle to adjust the voltage at pin 1 so that it is equal to that of pin 2. So, when voltage
at pin 1 is 5.1V, voltage at output is 5.1V * 57 = 290.7V and this is close enough to our 290V target. If
greater accuracy is required, one of the resistors can be either replaced with a pot or in series with a
pot and the pot adjusted to give required reading.
The parallel combination of the resistor and capacitor between pins 1 and 9 provides feedback
compensation. I won’t go into detail into feedback compensation as it is a vast topic on its own.
Pins 11 and 14 drive the MOSFETs. There are resistors in series with the gate to limit gate current.
The resistors from gate-to-source ensure that MOSFETs don’t get accidentally turned on.
So that’s about it. You can see that this is quite an easy circuit to design. If you’ve understood all of
this, you can now design circuits with SG3525 yourself. Try to make a few, eg for 50Hz output and
with isolated feedback. If you can’t don’t worry, I’ll put up another article with a few more circuits using
SG3525 so that you become completely clear with it (if you haven’t already).
Edited by - liang2408 on Feb 04 2
Typical circuit for welding equipment shown on the following circuit diagram. Turn on delay can be controlled
accurately with Potentiometer P2. We can discharge C1 at each line zero voltage using DB1 diode bridge and R6R7 resistors. The voltage charge will be reset at each new half line cycle and the turn-on delay will be maintained
the same.
Through both potentiometers, the Transil reduces power dissipation. Here’s the circuit diagram:
Note: Transil is a transient voltage suppression diode trademarked by STMicroelectronics [Source:
STMicroelectronics Application Note]
High Efficiency Sine Wave Inverter - Part 1
The goal of this series of pages is to explain and ultimately build a high efficiency pure sine wave inverter (DC to AC converter) from
scratch. I have been working on this project for about 6 months and so far the inverter is basically working.
This inverter is not of the type that requires a transformer to step up the voltage. It is basically a monster high power audio amplifier
designed to run appliances. For this reason its output is not like most commercial sine inverters. The great majority of inverters out
there can generate a very blocky and noisy sine wave. This inverter is generating a nice and smooth high fidelity sine wave just as an
audio amplifier would.
The main working blocks of the inverter are illustrated in the diagram. The first block is the
power source. For this project I decided to go high voltage so the battery bank is 120VDC
nominal. I decided to go high voltage because of efficiency reasons. Using this voltage will
allow us to output the same power with very little current thus preventing MOSFETs from
dissipating too much energy as heat.
Another reason for choosing a high voltage power source is because to make a perfect output
sine wave we need to have a DC voltage who's amplitude is equal to the output peak to peak
voltage of the sine wave.
As you might have figured out by now, it takes 175+175 volts input to create a 120V RMS sine
wave. Using 120V as input is closer to 350V thus lowering the power losses associated with
stepping up the voltage. If the same project would have been done with a standard system voltage of 12V, 24V or 48V the losses
would have been unacceptable for the scope of this project.
The next building block is the device that will step up the voltage from 120V nominal to a regulated 360VDC. An inductive boost
converter was chosen as they are known to be able to operate at around 97% efficiency. More detailed information about this block
will be given later.
Now Ill discuss the main part of the project. The next block is the inverter. Now that we have our required input voltage for the
desired output voltage all that is left to do is to convert it to an AC wave. As we all know this could be done easily by operating the
MOSFETs in the linear region. I am not doing that because operating the MOSFETs in the linear region will give extremely low
efficiencies.
To solve this problem I encoded the 60Hz sine wave into a high
frequency square wave. This way the MOSFETs are being switched
by regular square waves. The only difference is that the square
wave carries the information of the sine wave encoded as a series
of changing pulse widths.
As you can see in the pictures to the left, the pulse width
increases proportionally to the voltage of the reference sine wave.
Click the images for a larger view.
This 'trick' is achieved by applying a generated sine wave of the
same frequency as you want your output sine wave to be into the
PWM comparator of your PWM chip. I did not used a standard technology PWM chip. what I did instead was to use op amps (tl084) to
make my own PWM chip combined with a 555 timer as a ramp generator. It was done like that because it gives me more freedom to
change every aspect of the circuit easily and consciously. I have nothing against using a readily made PWM chip like the tl494. It
should work as well but I have not done any testing with it to be sure.
The last part of the basic system is a simple low pass filter. The purpose of this filter is to filter out the high frequency that carries
the information about our precious sine wave. After the filter all that is left is the information that we inputed (the pure and clean
sine wave).
In the next part of this article I will have posted schematics describing each block in detail. The power source block (batteries) will
not be discussed any further. Thank you for reading this and any comments are welcomed.
WARNING: This information, all pictures shown here and all schematics are copyrighted material. The owner of this material,
Argenis Bilbao (myself) prohibit the use of this information for anything other than personal use or educational purposes. The
use of this information for commercial purposes without my authorization is a violation to copyrighted material and will be
prosecuted by law.
EXTREMELY High voltages are used in this project so extreme caution must be used while handling any electronics. I am not to
be held responsible for any damages caused to you or others by the use of this information. This information is provided AS IS.
High Efficiency Sine Wave Inverter - Part 3 (Inverter Board)
To recap a little bit Ill briefly explain how the sine wave inverter works. Basically this inverter
is a Class D audio power amplifier designed to work with high voltages. The inverter creates a
square wave suitable for MOSFET switching with minimal power loss as heat because the
MOSFETs will not be in the linear region. There is an on-board sine wave generator that is used
as the input signal to the “amplifier”. After filtering out the high frequency square wave, the
amplified input signal generated by the on-board oscillator remains. If you need further and
more detailed explanation please read the first article of this series HERE.
Here is the schematic for the oscillator that generates the input signal:
This is simply a quadrature oscillator. The first op-amp generates a square wave of the
required output frequency (in this case 50Hz or 60Hz). This square wave is then filtered of all
its harmonics making it a beautiful sine wave by the second op-amp that is configured as an
active low pass filter. The active filter should have a cut-off frequency just a little above of
the desired output frequency. Then another op-amp is configured as an unity gain amplifier to
act as a buffer so that the following steps don't deform the generated sine wave.
To implement the PWM (pulse width modulation) circuit a ramp generator was used in
conjunction with an op-amp configured as a comparator. The schematics of the ramp generator
is illustrated above. I will not discuss in detail how this is accomplished, there are numerous
555 timer tutorials in the internet. The 555 is operated with a current limiter to charge a
capacitor at constant current. One of the properties of the capacitors is that voltage can not
change abruptly. Since we are charging it at constant current the voltage across the capacitor
increases linearly. The 555 timer controls the time that the capacitor is going to be charged (it
sets the PWM frequency, in this case 20kHz).
Now, here is where the magic takes place. The generated ramp is fed to a comparator as well
as the sine wave generated by previous block. When the voltage of the ramp is greater than the
voltage of the sine (reference voltage) it generates a perfectly square pulse lasting until the
voltage of the ramp drop below the sine voltage.
NOTE: This image is property of:
http://www.cpemma.co.uk/pwm.html
In the illustration all the pulses are of equal
length because the reference signal is a
constant voltage but in the circuit the
pulses will be of different length because the reference is a sine wave (variable voltage).
The output signal is then fed to a 4011 CMOS NAND gate. This chip squares up the wave even
better and it also generates a wave that is 180 degrees out of phase. This makes possible
current flow in the output. If both wires (power output of the inverter) are in phase then the
voltages will be the same thus no current would be able to flow.
Now I think that this is the most important part of the circuit, the two half bridge drivers.
There are two half bridge drivers in this circuit and they essentially do the same task. The only
difference is that one side takes care of the original signal and the other side deals with the
signal that have been shifted by 180 degrees. It is necessary to use a H bridge driver because
the MOSFETs that are connected to the positive rail have their Source pin “floating” and thus
they can not be fully turned off creating a very serious condition in the half bridge. When this
happens both MOSFETs are conducting thus creating a short to ground. The IR21834 chip
handles this by making a “fake ground” at the high side MOSFET's Source pin. Then, the gate is
driven by a higher voltage to ensure that the MOSFET is fully on when a pulse from the signal
comes.
Spike filtering is done with TVS diodes (transient voltage suppressor diodes) across each
MOSFET and a very famous 0.1uF@1000V capacitor from the positive rail to ground. This
capacitor absorbs large spikes and if any voltage remains above desired limits the TVSs take
care of it by dissipating it as heat. TVS diodes work as zener diodes. When there is a voltage
too high it conducts infinite amount of current (reverse biased) to ground.
As in any other driving circuit, care must be taken about ringing. All MOSFETs have a 5.6 Ohm
resistor with a diode across it to avoid ringing. Zener diodes are used in the gates too in case
that there is a spike that exceeds the Gate to Source voltage of the MOSFETs. This zener should
dissipate that spike by clamping it. I used a 1k Ohm resistor from Gate to Source to make sure
that the MOSFET's internal parasitic capacitor discharges entirely before the next pulse comes
by.
The output from the MOSFETs is the power output, now all thats left is to filter the high
frequency out and stay with the amplified sine wave. The filter is very simple, it is just two
coils and one capacitor connected across the output of each half bridge. An illustration is
shown below.
Efficiency tests will be posted shortly. Click the image for the full schematic of the inverter
board:
As a bonus here is a picture that I took of the oscilloscope's screen. It shows the output before
and after filtering.
I want to give special thanks to Don Carroll and to Ross Wheeler for helping me out so much
with this project. Without them this project would have been impossible.
WARNING: This information, all pictures shown here and all schematics are copyrighted
material. The owner of this material, Argenis Bilbao (myself) prohibit the use of this
information for anything other than personal use or educational purposes. The use of this
information for commercial purposes without my authorization is a violation to copyrighted
material and will be prosecuted by law.
EXTREMELY High voltages are used in this project so extreme caution must be used while
handling any electronics. I am not to be held responsible for any damages caused to you or
others by the use of this information. This information is provided AS IS.
TEK FAZLI 4 SEVİYELİ EVİRİCİ
Minimum anahtarlama frekansları ile verimleri çok yüksektir ( > %98),
DEVRE ŞEMASI
Köprü tranzistorleri tetiklemek için bağımsız D.A. kaynakları trafoların ayrık sargılarından
sağlanmıştır. Optokuplör yerine IR2110 ( Yarım Köprü sürücü tümdevre ) veya IR2130 ( 3 fazlı evirici sürücü
tümdevre ) entegreleri kullanılabilir. Bu entegreler üst mosfetleri tetiklemek için kondansatör şarj ve
deşarjından faydalanırlar, trafolara gerek kalmaz. 15 Khz anahtarlama frekansında yumuşak anahtarlama
için paralel rezonans kolu ile kısmen sönümleme (snubber circuit) sağlanabilir.
DEVRE ÇALIŞIRKEN
20 Hz çıkış frekansında görülen gerilim
Çıkış trafosu 9v / 220v 50 hz değerlerine göre tasarlanmış olup, gerilim frekansının yükseltilmesiyle
birlikte reaktanstaki artışla beraber gerilimin genliğinde azalma görülmüştür. Bunun için PWM yöntemiyle
aynı zamanda gerilim genliği de ayarlanmalıdır ( 5 açı yöntemi uygulanabilir ).
6 Khz Anahtarlama Frekansında Eviricinin Çıkış Grafiği
Trafonun Sekonderinde Görülen Gerilim Sinüs Dalga
Şekli
Mikrodenetleyici B portu çıkışı tetikleme sinyallerinin tek periyotluk zaman diyagramı.
T2 , T4, T6, T8 sinyalleri bunların tümleyenidir.
KAYNAKLAR
[1]J. Rodriguez, J. S. Lai and F. Z. Peng, “Multilevel Inverters: Survey of Topologies, Controls, and
Applications,” IEEE Transactions on Industry Applications, vol. 49, no. 4, Aug. 2002, pp. 724-738.
[2] J. S. Lai and F. Z. Peng, “Multilevel Converters-A new Breed of Power Converters,” IEEE Trans. Ind.
Applicat., vol.32,pp. 509-517, May/June 1996.
[3] L. M. Tolbert, F. Z. Peng, and T. Habetler, “Multilevel Converters for Large Electric drives,” IEEE
Trans. Ind. Applicat.,vol.35,pp. 36-44, Jan./Feb. 1999.
[4] R. H. Baker and L. H. Bannister, “Electric Power Converter,” U.S. Patent 3 867 643, Feb. 1975. [5]
A. Nabae, I. Takahashi, and H. Akagi, “A New Neutral-point Clamped PWM inverter,” IEEE Trans. Ind. Applicat.,
vol. IA-17, pp. 518-523, Sept./Oct. 1981.
EK FAZLI İNVERTER
DEVRE ŞEMASI
MALZEME LİSTESİ
1 ADET
470K
1 ADET
56nF
1 ADET
10K
1 ADET
CA3524
1 ADET
10
2 ADET
1K
2 ADET
22K
1 ADET
IRFZ44
1 ADET
1K
1 ADET
100nF/400V Kondansatör
1 ADET
2x12V/220V/150W
Kondansatör
MOSFET
Transformatör
İnveterlerin Çalışma Prensibi
Şekil 2. Devre Şeması
Devrede Kullanılan Elemanlar
Eleman adı
Değeri
R1
470 Ω
R2
470 Ω
R3
220 K Ω
C1
100nF
T1 T2
BD243C
T3 T4
2N3055
TRAFO
2X12/220 Volt 120 Watt
IC
4047 multivibratör devresi
Kaynak
12 Volt 10 amper DC kaynak
CD4047 ile inverter Uygulama Devresi
Düşük güçlü bir multivibratör devresi olan CMOS 4047 entegresi devrenin kalbini teşkil
etmektedir. Entegre simetrik bir kare dalga üreteci olarak kullanılmıştır. Entegre çıkışları
akım ve gerilim olarak düşük seviyede olduğu için bu çıkışlar T1 ve T2 güç transistörleri ile
kuvvetlendirilerek trafonun 2X12 volt girişine uygulanmıştır, böylece trafonun diğer
tarafından 220 volt luk gerilim elde edilmiştir.
Malzeme Listesi
R1,R2 470Ω
R3 220kΩ
C1 100nf
T1,T2 2N3055
Trafo 2x 12 volt 120 watt
IC CD4047
CD4047 ile Yapılan 12V'tan 220V'ta DC-AC Çevirici Devresi
PCB Layout For Cheap 12V to 220V Inverter Circuit Diagram
COMPONENTS LIST
Resistors
R1 = 18k?
R2 = 3k3
R3 = 1k
R4,R5 = 1k?5
R6 = VDR S10K250 (or S07K250)
P1 = 100 k potentiometer
Capacitors
C1 = 330nF
C2 = 1000 µF 25V
Semiconductor
T1,T2 = MJ3001
IC1 = 555
IC2 = 4013
Miscellaneous
LA1 = neon light 230 V
F1 = fuse, 5A
TR1 = mains transformer, 2x9V 40VA (see text)
4 solder pins
4047 ile bir inverter
4 SEVİYELİ EVİRİCİ
300W Push-Pull Evirici
REFERENCES
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30. A. Coekin, High-Speed Pulse Techniques, Pergamon, 1975. (TK7835.C56 1975).
31. Csaki, I. Hermann, I. Ipsits, A. Karpati, and P. Magyar, Power Electronics Akademiai Kiado,
Budapest, 1979. (ISBN 963-05-1671-3).
32. B. Dewan and A. Straughen, Power Semiconductor Circuits, John Wiley & Sons, Inc., 1975.
33. K. Dubey, Power Semiconductor Controlled Drives, Prentice Hall, Englewood Cliffs, 1985.
34. Hans-Peter Hempel, Power Semiconductor Handbook, SEMIKRON, 1980. (ISBN 3-9800346-15).
35. R. G. Hoft, Semiconductor Power Electronics, Van Nostrand, New York,
1986.(TK7871.85.H65).
36. L. Kusic, Computer-Aided Power Systems Analysis, Prentice-Hall, 1986. (TK1005.K87 1986).
37. W. Lander, Power Electronics, McGraw-Hill, 1981. (ISBN: 0-07-084123-3).
38. M. Miller, Is power electronics a national priority ?, Power Conversion & Intelligent Motion
Control, March 1987.
39. Mohan, T. M. Undeeland, and P. Robbins, Power Electronics, John Wiley, New York, 1989.
40. M. D. Murphy and F. G. Turnbull, Power Electronic Control of AC Motors, Pergamon, New
York, 1988.
41. E. Newell and J. W. Motto, Introduction to Solid State Power Electronics, Youngwood:
Westinghouse Electric Corporation, 1977.
42. S. Oxner, Power FETs and Their Applications, Prentice-Hall Inc., 1982.
43. Pearman, Power Electronics: Solid State Motor Control, Reston Publishing Company, Inc.,
1980.
44. Pearman, Solid State Industrial Electronics, Reston Publishing Company, Inc., 1984. (ISBN: 08359-7041-8) (TK7881.P43).
45. Rajagopalan, Computer Aided Analysis of Power Electronic Systems, Marcel Dekker, New
York, 1987.
46. H. Rashid, Power Electronics, Prentice Hall, Englewood Cliffs, 1988.
47. H. Seidman, H. Mahrous, and T. G. Hicks, Handbook of Electric Power Calculations, 1983.
(ISBN 0-07-056061-7).
48. P. Severns and G. E. Bloom, Modern DC-to-DC Switchmode Power Converter Circuits, Van
Nostrand Reihold Company Inc..
49. E. Tarter, Principles of Solid State Power Conversion, Howard W. Sams, 1985.
50. W. Williams, Power Electronics, John Wiley, New York, 1987.
C. Power Supplies
51. Chryssis, High-Frequency Switching Power Supplies Theory and Design, McGraw-Hill, 1984.
(ISBN 0-07-010949-4) (TK868.P6C47).
52. Gottlieb, Regulated Power Supplies, third edition, Howard W. Sams & Co., Inc., 1984.
53.
54.
55.
56.
57.
58.
59.
60.
61.
62.
63.
64.
65.
66.
67.
Gottlieb, Power Supplies: Switching Regulators Inverters & Converters, 1984.
Griffith, Uninterruptible Power Supplies, Marcel Dekker, New York, 1989.
Hnatek, Design of Solid State Power Supplies, Van Nostrand, New York, 1981.
Lee (Ed.), High-Frequency Resonant, Quasi-Resonant, and Multi-Resonant Converters,
Virginia Power Electronics Center, 1989.
Lee (Ed.), Modeling, Analysis, and Design of PWM Converters, Virginia Power Electronics
Center, 1990.
Middlebrook and S. Cuk (Eds.), Advances in Switching Mode Power Conversion, vols. I & II,
TESLA Co., Pasadena, California 1983.
MOTOROLA, Switchmode Application Manual, Motorola Inc., 1981.
MOTOROLA, Linear/Switchingmode Voltage Regulator Handbook: Theory and Practice,
1981.
Pressman, Switching and Linear Power Supply, Power Converter Design, Hayden, Rochelle
Park, 1977.
Rensink, Switching Regulator Configurations and Circuit Realization, Ph.D Thesis by Loman
Rensink, California, 1979.
Severns and G. E. Bloom, Modern DC - to - DC Switch Mode Power Converter Circuits, Van
Nostrand, New York, 1985.
Sum, Switch Mode Power Conversion: Basic Theory and Design, Marcel Dekker, New York,
1984.
Wood, Switching Power Converters, Van Nostrand, New York, 1981.
UNITRODE, Unitrode Switching Regulated Power Supply Design Seminar Manual, Unitrode
Corporation, 1985.
UNITRODE, Applications Handbook, Unitrode Corporation, 1985.
D. Electronic Equipment Thermal Design, Package Design
68. N. Ellison, Thermal Computations for Electronic Equipment, Van Nostrand Reinhold
Company, New York, 1984.
69. D. Kraus and Avram Bar-Cohen, Thermal Analysis and Control of Electronic Equipment,
Hemisphere Publishing Corporation, Washington, 1983. (ISBN 0-07-035416-2)
(TK7870.25.K73).
70. S. Matisoff, Handbook of Electronics Packaging Design and Engineering, Van Nostrand
Reinhold Company, 1982.
71. S. Steinberg, Cooling Techniques for Electronic Equipment, John Wiely & Sons, Inc., 1980.
(TK7870.25.S73).
E. Noise Reduction Techniques
72.
73.
74.
75.
76.
77.
78.
79.
W. Denny, Grounding for the Control of EMI.
J. Geogopoulos, Fiber Optics and Optical Isolators.
N. Ghose, EMP Environment and System Hardness Design.
C. Hart and E. W. Malone, Lighting and Lighting Protection.
Mardiguian, Electrostatic Discharge - Understand, Simulate and Fix ESD Problems.
Mardiguian, Interference Control in Computers and Microprocessor-Based Equipment.
Mardiguian, How to Control Electrical Noise.
Morrison, Grounding and Shielding Techniques in Instrumentation, second edition, John
Wiley & Sons, Inc., 1977.
80. Morrison, Instrumentation Fundamentals and Applications, John Wiley & Sons, Inc., 1984.
81. W. Ott, Noise Reduction Techniques in Electronic Systems, Wiley-Interscience Publication,
1976.
82.
83.
84.
85.
86.
87.
88.
89.
A. Smith, Coupling of External Electromagnetic Fields to Transmission Lines.
R. J. White and M. Mardiguian, EMI Control Methodolgy and Procedures.
R. J. White, EMI Control in the Design of Printed Circuit Boards and Backplanes, 248 Pages.
R. J. White, Shielding Design Methodlogy and Procedures.
R. J. White, Electrical Filter.
R. J. White, Electromagnetic Shielding Materials and Performance.
EMC EXPO, 1986 Symposium Record, 416 Pages.
EMC Library :
vol. 1 Electrical Noise and EMI Specifications
vol. 2 EMI Test Methods and Procedures
vol. 3 EMI Control Methods and Techniques
vol. 4 EMI Test Instrumentation and Systems
vol. 5 EMI Prediction and Analysis Techniques
vol. 6 EMI Specifications, Standards, and Regulations