2017 IEEE International Conference on RFID (RFID)
Design of a Backscatter-Based Tag-to-Tag System
Yasha Karimi∗ , Akshay Athalye∗ , Samir R. Das† , Petar M. Djurić∗ and Milutin Stanaćević∗
∗ Department
of Electrical and Computer Engineering, Stony Brook University, Stony Brook, NY, USA
of Computer Science, Stony Brook University, Stony Brook, NY, USA
Email: yasha.karimi@stonybrook.edu
† Department
Abstract—Practical technologies for the Internet of Things
(IoT) must provide connectivity to all objects under a common
framework irrespective of their size or value. Power requirement,
cost of wireless devices and scalability have proved critical
bottlenecks for the universal deployment of the IoT. One approach to address these issues is the use of a communication
paradigm where the devices communicate via backscattering
and exploit harvested power from an external RF source. In a
Backscattering Tag-to-Tag Network (BTTN), the tags themselves
are able to read and interpret the backscattered communications
from other neighboring tags. In the tag-to-tag link, the BTTN
tag has to demodulate a receiving signal with a low modulation
index. In order to improve the link range, we propose a powerefficient demodulator design that enables the receiving tag to
quantify the amplitude-shift keying (ASK) modulated signal with
a modulation index as low as 0.6%. The demodulator consumes
1.21 µW at 1.1 V supply voltage at a data rate of 10 kbps.
I. I NTRODUCTION
For more than a decade, the most widely researched embodiment of backscatter communication has been the Radio
Frequency Identification (RFID) technology [1]–[3]. Extensive
work has been done on designing RFID tags and readers and
improving the data rate of the backscatter modulation [4], [5].
However, the RFID technology has been mostly limited to the
identification and localization tasks until the introduction of
computational RFID tags that have been based on the Intel
WISP platform [6], [7], a programmable RFID tag with an
on-board microcontroller with sensing inputs. Various power
management techniques for combining power harvested from
different sources as a power source in computational RFID
systems, like solar cells, have been investigated [8], [9].
A major limitation of conventional RFID systems is that
the communication link is not completely passive, i.e., passive
tags can only communicate with active readers. This proves
an obstacle to the extension of conventional RFID technology
into wireless sensor network (WSN) applications due to the
infrastructure and deployment costs involved with a network of
RFID readers. To combat this problem, networks with similar
reader-tag communication links, but in which RFID readers
are replaced with commodity devices, like Bluetooth radios
or WiFi access points, have been proposed. In these networks,
the tags can communicate with these devices by emulating the
appropriate signals according to the communication standards
such as Bluetooth [10] and WiFi [11]–[13]. These tags operate
at high data rates and require higher harvested power.
The concept of backscatter-based communication directly
between passive devices for applications where data rates can
be low and the harvested power is scarce is starting to gain
978-1-5090-4576-1/17/$31.00 c 2017 IEEE
978-1-5090-4576-1/17/$31.00 ©2017 IEEE
research interest. Tag-to-tag communication using backscatter
between two standard passive tags was explored in [14]. The
electromagnetic models that govern this communication were
discussed in [15]. These early efforts mostly focused on establishing close-range communication between standard passive
tags. While these efforts used an RFID reader to provide the
excitation signal for passive tag-to-tag communication, recent
efforts have moved away from this to using more general
excitation sources. The design presented in [16] uses TV
signals for the excitation, and it demonstrates that with a
conventional architecture of passive RFID tags, the range of
tag-to-tag communication is limited to a fraction of a meter.
In [17], the authors implement a CDMA encoding technique
to extend the range of the tag-to-tag link and demonstrate
a reading distance of 90 ft with 0 dBm incident power and
0.33 bps data rate. In [18], we have shown how to increase
the efficiency of the power harvesting circuit and demodulator
sensitivity with respect to the input power.
In this paper, we present the characterization of the
backscatter tag-to-tag link in BTTN (Backscatter Tag-to-Tag
Network) through a derivation of the modulation index of the
signal at the receiving tag. The modulation index determines
the maximum distance at which tags can communicate. With a
novel architecture of the demodulator, we demonstrate a longer
range in tag-to-tag communication networks.
II. H ARDWARE C HALLENGES IN THE D ESIGN OF BTTN
TAGS
Although the concepts of transmitting data by backscatter
modulation and receiving data using passive envelope detection have been widely explored in the context of backscatterbased systems, their application to BTTN systems gives rise
to unique challenges. Specifically, in the forward (reader to
tag) link of RFID systems, the incoming signal at the tag
has a very large modulation depth (greater than 75%). This
makes the passive demodulation at the tag straightforward to
implement. In the reverse (tag to reader) link, the backscatter
signal from the tag has a low modulation index, but it is
demodulated by an RFID reader that implements active IQ
demodulation and active cancellation of the interfering carrier
signal. As opposed to this, in BTTN systems, the receiving
tag has to perform demodulation of the incoming signal with
a low modulation index through passive envelope detection.
We derive the relationship between the modulation index of
the received signal and the parameters of the tag-to-tag link.
Assuming that the tag can harvest enough energy for the
2017 IEEE International Conference on RFID (RFID)
Exciter
20
S
Ptx=-10dBm
Ptx=-15dBm
Ptx=-20dBm
Ptx=-25dBm
Ptx=-30dBm
18
16
14
dST
dSR
m[%]
12
10
8
T
dTR
Transmit tag
R
6
4
Receive tag
2
0
0.5
Fig. 1. Backscatter tag-to-tag communication setup.
1
1.5
2
2.5
dTR[m]
operation, the communication range of the tag-to-tag link is
determined by the modulation index of the received signal that
the tag can resolve.
Fig. 2. Modulation Index as a function of distance between communicating
tags when the input power of the receiving tag is -20dBm.
A. Link budget analysis
express the modulation index of the signal received at Rx tag
as
1
m(dT R ) =
,
(3)
1 + k(dT R )
We present a link budget analysis in a typical BTTN setup
and specifically investigate the dependence of the modulation
index of the incoming signal on the distance between the transmitting (Tx) and receiving (Rx) tags. Consider a single BTTN
link shown in Figure 1. It consists of a dedicated excitation
source (S) that sends out a continuous wave (CW) signal, one
transmitting tag that backscatters this signal, and one receiving
tag which passively demodulates the backscattered signal. It
can be seen that the level of the excitation signal at the Tx
and Rx tags, respectively, depends on the distances dST and
dSR , and that the strength of the backscatter signal at the Rx
tag depends on the distance dT R . Backscattering is achieved
at the Tx tag by changing the load impedance ZL of the
antenna between two states. The power of the backscatter
signal depends on the differential radar cross section (RCS),
which can be expressed as [19],
λ2 GA 2 ∗
|Γ L,1 − Γ∗ L,2 |2 ,
(1)
4π
where λ is the wavelength of the excitation, GA is the antenna
gain, and Γ∗L,i is the conjugate antenna reflection coefficient at
modulation state i, i = 1, 2. This coefficient can be expressed
by
∗
ZL,i − ZA
Γ∗L,i =
, i = 1, 2,
(2)
ZL,i + ZA
∆σ =
where ZA is the impedance of the tag antenna. The difference
in the excitation power levels received at Tx and Rx and the
difference in the relative phase of the signals received at Rx for
the two states of the transmitting tag determine the behavior of
the BTTN link. To simplify the initial analysis, we assume that
the transmitting tag is switching between the non-reflecting
and reflecting states and that the excitation and backscatter
signals at the Rx tag are in phase. Using the Friis formula for
the RF signal propagation and ignoring reverberations, we can
where
4πdT R
k(dT R ) = 2
GA λ
r
Prx
,
Ptx
(4)
with Ptx and Prx denoting the incident powers at the Tx and
Rx tags, respectively. In Figure 2, we show the modulation
index as a function of distance between the tags for different
power levels at the transmitting and receiving tags assuming
that the link operates at 915 MHz and that the tag antennas are
dipole antennas with a gain of 2.2 dB. We notice that for long
communication ranges, the modulation index of the signals is
low.
If the excitation power at both, the Tx and Rx tags is the
same, the BTTN link is completely symmetric, i.e., the link
behavior will be identical for the communication in either
direction between the Tx and Rx, and it only depends on the
distance between the tags, and not on the excitation power.
However, when the excitation power seen by both tags is
different, which will be the case in many practical scenarios,
this behavior no longer holds. From a high level observation of
the link, it is clear that if the Tx tag is closer to the exciter, it
will receive a higher excitation signal and hence, the strength
of the transmitted backscatter signal will be proportionally
higher. So a tag closer to the exciter will be a better transmitter.
Conversely, if Rx tag is closer to the exciter than Tx tag and
receives a higher excitation signal, then the strength of the
backscatter signal it receives will be weaker relative to this
excitation signal. So a tag that is further away from the exciter
will be a better receiver. This behavior is very important to
consider when designing and characterizing BTTN systems.
2017 IEEE International Conference on RFID (RFID)
Processing Section
Low Power
1.4
200
1.2
180
Vdd
Digital Logic
Data
160
1
Demodulation Section
Vripple[mV]
Baseband Amplifier
& Comparator
Envelope Detector
Vdd
140
0.8
120
0.6
100
0.4
80
Matching
0.2
Super Capacitor
& Regulator
Voltage Multiplier
Off-Chip
Energy Harvesting Section
IC
Venv[mV]
Switch
Antenna
60
0
10 -1
10 0
10 1
40
10 2
W[ m]
Fig. 3. Block diagram of BTTN tag architecture.
CC
Fig. 5. The ripple voltage at the output of the envelope detector as a function
of the width of the transistors in the voltage doubler.
M2
Venv
Rf
+
RO
Vdem
CO
Cf
M1
-
Cg
-
Venv
Fig. 4. Voltage doubler circuit for the envelope extraction of the input RF
signal.
III. D EMODULATOR D ESIGN
A high level block diagram of a BTTN tag architecture
is shown in Figure 3. Here, we focus on the design of a
demodulator circuit that will be able to resolve the received RF
signal with a low modulation index. We propose an architecture consisting of a voltage doubler for envelope extraction and
an amplifier with integrated filtering followed by a comparator.
A. Voltage Doubler for Envelope Extraction
For extraction of the envelope of the received RF signal, a
voltage doubler, shown in Figure 4, is connected to the antenna
circuit. The voltage doubler rectifies and at the same time
increases the amplitude of the input signal. The sizing of the
transistors M1 and M2 , along with the values of the resistor
Ro and capacitor Co are determined through optimization of
the performance of the envelope detector. The average value of
the voltage at the output of the voltage doubler has to be maximized, while the ripple voltage is minimized. Additionally, the
detector should be able to track the modulated envelope. In
conventional RFID system, due to the high modulation index,
this presents a more stringent constraint on the time constant
of the envelope detector than in a tag-to-tag system designed
to detect baseband signals with a low modulation index.
We first set the values of the resistor Ro and capacitor
Co . The time constant Ro Co should be large to reduce and
block the carrier signal (>1.09 ns). On the other hand, this
value is limited by the data rate of the ASK modulated signal
Vref
Vamp -
+
Vref
+
Vcomp
Fig. 6. Processing of the baseband signal after the envelope detector: amplifier
with integrated high-pass filter followed by comparator.
(<100 µs). The value of the resistor is selected to be 100 kΩ,
while the value of the capacitor Co is 5 pF. After sizing the
resistor and capacitor, we examine the optimal sizing of the
transistors M1 and M2 . If the W/L ratio of the transistors is too
small, the voltage drop across the transistor will be large and
it will reduce the average output voltage. On the other hand,
increasing the sizing of the transistor increases the leakage
current, which reduces the output voltage and increases the
ripple voltage. The optimal sizing of the transistor is obtained
through simulations. Figure 5 shows the ripple and the output
voltage as a function of the width of the transistors. The
chosen value of 1 µm leads to a ripple voltage of 239 µV
at input power of -28 dBm. This width provides a large
output voltage and a low ripple, thereby providing a smooth,
detectable baseband signal input to the amplifier.
B. Amplifier and Comparator
After the envelope detector, the ASK modulation in the
baseband signal, due to the low modulation index, cannot be
distinguished by a comparator. Instead, the baseband signal is
first amplified with integrated high-pass filter and the amplifier
is followed by the comparator, as shown in Figure 6. With low
data rates in tag-to-tag communication systems, on the order of
10 kbps, the RC time constant of the high-pass filter has to be
of the order of 500 µs. To realize a high resistance, the Rf is
2017 IEEE International Conference on RFID (RFID)
Vdd
Vdd
+
Vin
M11
M13
VB
M1
M2
M3
M4
Vdd
-
M12
M9
Vin
VB3
M10
VB2
VB1
M7
M8
M5
M6
Vout
TABLE I
Devices
M1 − M2
M3 − M4
M5 − M6
M7 − M8
M9 − M10
M11 − M12
M13
HIGH - GAIN AMPLIFIER
W/L(µm)
37.47/0.112
9.45/0.112
0.5/0.5
0.54/0.135
0.562/0.112
0.337/0.202
0.337/0.225
implemented as a diode connected transistor. The capacitances
Cg and Cf are set to 18 pF and 165 fF, respectively, providing
a DC gain of 101.
The high-gain amplifier is implemented as a low-noise,
low-power folded-cascode amplifier shown in Figure 7 [20].
The introduction of the high-gain amplifier in the path of
the baseband signal increases the power consumption of the
modulator. As the tag is passive, the power is harvested
from the input RF signal, and the power consumption is a
critical constraint in the design of the high-gain amplifier. The
noise of the amplifier determines the minimum distinguishable
modulated signal, leading to the low-noise design.
The transistors in the high-gain amplifier, except the transistors in the biasing circuit, operate in the subthreshold
region, since this region provides the highest gain-to-power
ratio [20]. The transistor sizing is summarized in Table I.
The transconductance of the amplifier is dominated by the
transconductance of the input transistor M1 , which in the
subthreshold region of the operation is equal to
gm1 =
4kT
(1 + m),
gm1 κ
(6)
4κ(gm5 + gm11 )
,
3gm1
(7)
vn2 =
Fig. 7. A folded-cascode implementation of low-noise low-power high-gain
amplifier.
S IZING OF TRANSISTORS IN
of the amplifier is determined by gm1 and the total loading
capacitance. The stability is dependent on the biasing current
in the cascoded branch, and with reducing this current, the
phase margin of the amplifier reduces.
We assume that the noise at the output of the amplifier is
dominated by the high-gain amplifier noise. As the amplification is integrated with high-pass filtering, the flicker noise
will be dominated by the thermal noise. Considering only
the thermal noise, the input referred noise of the high-gain
amplifier is [20]
κID1
,
UT
(5)
where ID1 is the biasing current of transistor M1 , UT is the
thermal voltage and κ equals 1/n, where n is the subthreshold
slope coefficient. The gain of the amplifier is proportional to
the biasing current of the amplifying transistor through the
value of gm1 , while the output resistance is inversely proportional to the current in the cascoded branch. The bandwidth
m=
where k is the Boltzmann’s constant, T is the absolute
temperature and gm5 and gm11 are the transconductances
of the transistors M5 and M11 , respectively. Increasing the
biasing current of the amplifying transistor reduces the noise.
To reduce the input-referred noise contribution from transistors
M5 and M11 , the transistors are biased in strong inversion and
the current in the cascoded branch should be small.
We first set the ratio of the biasing current of the amplifying
transistor and the current in the cascode branch to 5 as a tradeoff between noise and stability. With this current ratio, the
factor m in (6) is 1.08. Then, the transconductance of the
input transistor, that is, the biasing current of the amplifying
transistor is set by the limit on the noise of the amplifier and
is derived in the presented analysis.
The smallest amplitude of the input signal of the amplifier
is defined by the ripple voltage in the baseband signal after
envelope extraction. From Figure 5, for the optimal width of
the transistors in the voltage doubler, the ripple voltage is
239 µV. As the lowest amplitude that can be resolved, we
choose 800 µV. An SNR of 11 dB at the output of the amplifier
leads to a bit error rate (BER) of 10−3 [21]. With a bandwidth
of B = 100 kHz, the lowest value of gm1 is
s
4kT B
Vm
Vn =
× (2.08) ≤ √
,
(8)
gm1 κ
2 × 12.6
gm1 ≥ 209.6
kT B
≥ 0.13µS.
κVm2
(9)
This means that the value of gm1 is limited by the settling
behavior of the amplifier. We set gm1 at 8.5 µS and the biasing
current of the transistor M1 at 302.8 nA.
IV. R ESULTS
We present experimental results with a conventional RFID
tag architecture that demonstrate the measured modulation
index in the tag-to-tag communication link and simulation
results of the proposed demodulator architecture.
2017 IEEE International Conference on RFID (RFID)
50
40
Gain [dB]
30
20
10
0
Fig. 8. Prototype BTTN tag with dipole antenna and conventional modulator
and demodulator implementation.
-10
10-2
10-1
100
101
102
103
104
f [kHz]
480
Fig. 10. Transfer function of the baseband amplifier with integrated high-pass
filter.
475
V env [mV]
B. Simulation Results
470
465
460
6
6.5
7
7.5
8
8.5
t [ms]
Fig. 9. Recorded voltage at the output of the envelope detector in the prototype
Rx tag.
A. Experimental Results
We have fabricated a discrete prototype of the BTTN tag,
shown in Figure 8, powered by a CR 1620 coin cell battery. It
includes a single dipole antenna and uses a discrete component
conventional architecture of modulator and demodulator. In
the demodulator, a two stage voltage multiplier, implemented
using zero bias Schottky diode HSMS-285x series from Avago
Technologies, was used for envelope extraction. The setup
similar to the one shown in Figure 1, with two prototype
tags, one acting as a Tx tag and one acting as an Rx tag
was used. On the Rx tag, the envelope detector is followed by
a high-resolution 16-bit 80 kbps analog-to-digital conversion.
The recorded signal is captured and stored in memory. In
Figure 9, the recorded signal after the envelope detection is
shown when the incident power at the Tx and Rx tags is 15dBm and the distance between the tags, dT R , is 1.5 m. We
would like to point that if a conventional demodulator is used,
the communication range in the tag-to-tag link is a fraction
of a meter, which is in accordance with the demonstrated
performance in [16].
The proposed design of the ASK demodulator for integration in the BTTN tag is simulated in the 45nm CMOS
technology. The supply voltage in this technology is 1.1 V.
The demodulator design was simulated in Keysight Advanced
Design Simulation (ADS) and Cadence Virtuoso.
We first simulated the high-gain amplifier and the baseband
amplifier with the integrated high-pass filter. The open loop
DC gain of the high-gain amplifier is 68 dB with a unity-gain
frequency of 12 MHz. The simulated gain of the baseband
amplifier is shown in Figure 10. The gain in the passband is
40.1 dB, with corner frequencies at 2.9 kHz and 50 kHz.
We simulated the complete architecture of the demodulator
comprising the envelope extraction, baseband amplification,
and quantization. The input power was held at -28 dBm
with ASK modulation at data rate of 10 Kbps. The transient
response of the demodulator at different nodes is shown in
Figure 11 for an input signal with a modulation index of
0.6%. To fully characterize the response of the demodulator,
RF signals with different modulation indexes were presented at
the input. The output voltage of the amplifier as a function of
the voltage after the envelope extraction is shown in Figure 12.
While the gain for the low modulation index is high, the
response is non-linear and saturates for larger voltages at the
input of the amplifier.
The power consumption of the demodulator is 1.2 µW,
which makes the proposed architecture suitable for BTTN
tags. The demodulator could continuously operate on the input
harvested power of -20 dBm, considering that the state-of-theart efficiency of the power harvesting circuitry at this power
level is 30% [22]. At this power level, we demonstrated that
the demodulator could resolve a modulation index of 0.6%.
From Figure 2, we can conclude that the tag-to-tag link could
operate at a distance of 2 m if the Tx tag is also receiving at
the least the same input power of -20 dBm.
2017 IEEE International Conference on RFID (RFID)
149.5
580
1
560
149
0.8
148.5
148
V comp [V]
V amp [mV]
V env [mV]
540
520
500
0.6
0.4
480
0.2
460
147.5
1.45
1.5
1.55
1.6
1.65
1.7
1.75
1.8
1.85
440
1.45
1.9
1.5
1.55
1.6
1.65
1.7
1.75
1.8
1.85
1.9
0
1.45
1.5
1.55
1.6
1.65
1.7
t [ms]
t [ms]
t [ms]
(a)
(b)
(c)
1.75
1.8
1.85
1.9
Fig. 11. Voltage transient response at the output of different building blocks of the demodulator: (a) envelope detector (b) baseband amplifier (c) comparator.
3
Vpp
[mV]
amp
10
2
10
−1
10
0
1
10
10
2
10
Vpp
[mV]
env
Fig. 12. Output of the baseband amplifier in the demodulator for different
amplitudes of the received signal after envelope detection.
V. C ONCLUSIONS
The ability of the tags to communicate with each other
through backscatter without the presence of a reader greatly
reduces the deployment cost of networks of these tags. While
the basic idea of tag-to-tag communication using backscattering and harvested RF power is not new, the state-of-theart has gone little beyond proving the basic feasibility. The
proposed design of the demodulator enables the tag-to-tag link
to operate at longer distances advancing the state-of-the-art in
backscatter-based tag-to-tag communication systems to new
levels.
ACKNOWLEDGMENT
This work was supported by the National Science Foundation (NSF) under grants CNS-1405740 and CPS-1646318.
Jihoon Ryoo developed the digital section of the tags and
helped in experimental setup.
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