IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 10, OCTOBER 2012
4329
Tapped-Inductor Buck HB-LED AC–DC Driver
Operating in Boundary Conduction Mode for
Replacing Incandescent Bulb Lamps
Diego G. Lamar, Member, IEEE, Marcos Fernandez, Student Member, IEEE, Manuel Arias, Member, IEEE,
Marta M. Hernando, Senior Member, IEEE, and Javier Sebastian, Senior Member, IEEE
Abstract—High-brightness light-emitting diodes (HB-LEDs) are
recognized as being potential successors of incandescent bulb lamps
due to their high luminous efficiency and long lifespan. To achieve
these advantages, HB-LED ballast must be durable and efficient.
Furthermore, for this specific application, ac–dc HB-LED ballast
requires a high-step-down ratio, high power factor and low cost.
This paper presents a tapped-inductor buck power factor corrector (PFC) operating in boundary conduction mode design for replacing incandescent bulb lamps. This low-cost solution presents a
suitable high-step-down ratio without galvanic isolation in order
to produce an output voltage of about 20 V from line voltage. In
addition, the tapped-inductor buck PFC maintains high efficiency
in comparison to other one stage solutions widely used to design
low-cost ac–dc HB-LED drivers (e.g., flyback PFCs). Static analysis, input current distortion analysis, and an average small signal
model of the tapped-inductor buck PFC have been implemented
in this paper both to check the validity of the proposed solution
and to provide a suitable design procedure of the ac–dc HB-LED
driver. Finally, a 12-W experimental prototype was developed to
validate the theoretical results presented.
Index Terms—AC-DC power conversion, current control, harmonic distortion, LEDs, lighting, power factor, switched mode
power supplies.
I. INTRODUCTION
N RECENT years, advances in solid-state lighting technology have changed traditional lighting solutions. Nowadays, high-brightness light-emitting diodes (HB-LEDs) are very
attractive light sources due to their excellent characteristics:
high efficiency, long lifespan, and low maintenance requirements [1], [2]. In addition, HB-LED packages are becoming
more and more robust while providing greater reliability than
traditional light sources (fluorescent lamps, incandescent lamps,
etc.). To achieve the advantages of HB-LED lamps, however,
HB-LED ballast must be both durable and efficient.
I
Manuscript received December 28, 2011; revised March 1, 2012; accepted
March 6, 2012. Date of current version May 31, 2012. This work was supported
by the Spanish Ministry of Education and Science under Consolider Project
RUE CSD2009-00046, Project DPI2010-21110-C02-0, and European Regional
Development Fund (ERDF) grants. A short version of this paper was presented
at the 2012 IEEE Applied Power Electronics Conference, Orlando, FL, with a
slightly different title. Recommended for publication by Associate Editor M.
Ponce-Silva.
The authors are with the Grupo de Sistemas Electrónicos de Alimentación (SEA), Universidad de Oviedo, 33204 Gijón, Spain (e-mail:
gonzalezdiego@uniovi.es;
fernandezdmarcos@uniovi.es;
ariasmanuel@
uniovi.es; mmhernando@uniovi.es; sebas@uniovi.es).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TPEL.2012.2190756
Since HB-LEDs are diodes, the default method for driving
them is by controlling the dc forward current through the semiconductor. If the primary energy source is the ac line, then some
type of ac–dc converter must be placed between the line and the
HB-LEDs [3], [4]. It is likewise known that the low-frequency
harmonic content of the line current must comply with specific
regulations. (EN 61000-3-2, Class C [5], [6] and the ENERGY
STAR program [7]). These regulations establish a very strict harmonic content, such that only very sinusoidal line waveforms
are able to comply with the aforementioned regulations. Therefore, the only practical method to comply with these regulations
is to use active high power factor (PF) converters, commonly
known as power factor correctors (PFCs).
When replacing incandescent bulb lamps with HB-LED luminaries, 12 HB-LEDs of 1 W are needed to produce the same
luminous flux as from a 60-W incandescent bulb. This HBLED configuration implies that low-cost electronic ballast with
a high-step-down ratio must be used. At this point, a flyback
PFC as a one stage solution is often used to design the HB-LED
ac–dc driver for this application. In this case, a sinusoidal input
current is obtained and galvanic isolation is provided. However,
as existing incandescent lamps are not isolated from the mains,
galvanic isolation is not mandatory in this application. Therefore, the flyback PFC may not be an optimum solution. In this
case, the transformer increases both the cost and size of the ac–dc
HB-LED driver. Also, its efficiency is not high enough [8]–[11].
For the aforementioned reasons, solutions other than the converters belonging the flyback family of PFCs can be used to
design low-cost ac–dc HB-LED drivers using only one stage
to convert ac to dc. In the last years, some authors have presented solutions based on the buck PFC in order to increase its
efficiency [12]–[14].
Inspired on solutions based on the buck PFC, this paper
presents a tapped-inductor buck PFC operating in boundary conduction mode (BCM) for replacing incandescent bulb lamps. As
will be shown in this paper, this solution presents a suitable highstep-down ratio without galvanic isolation in order to produce
an appropriate voltage to supply one string of HB-LEDs from
line voltage while maintaining both its simplicity (few components and one stage PFC solution) and higher efficiency than
other PFCs based on one stage (e.g., flyback PFCs). Section II
presents both the static analysis of the proposed solution and
the distortion analysis of the input current to guarantee compliance with standard regulations. The design procedure of the
proposed ac–dc HB-LED driver is described in Section III. The
0885-8993/$31.00 © 2012 IEEE
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Fig. 1.
PFC.
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 10, OCTOBER 2012
(a) Tapped-inductor buck PFC. (b) Modified tapped-inductor buck PFC. (c) Ideal input current and input voltage waveforms of the tapped-inductor buck
average small-signal analysis of the tapped-inductor buck PFC
operating in BCM is detailed in Section IV. Section V presents
the experimental results of a 12-W prototype used to verify the
theoretical analysis presented in the paper. Finally, Section VI
concludes with a summary of the main benefits and drawbacks
of this solution.
II. STATIC ANALYSIS OF THE AC–DC TAPPED-INDUCTOR
BUCK PFC OPERATING IN BCM
The buck converter is known to provide the most efficient
step-down topology at the lowest cost both for operating in continuous conduction mode and discontinuous conduction mode.
It is also a low-cost and low-size solution for designing a power
supply due to its few number of components and the use of
a one stage PFC solution. For aforementioned reasons, it is
an excellent option for designing an ac–dc HB-LED driver for
replacing incandescent bulb lamps. However, its efficiency decreases when a high-step-down ratio is needed due to the fact
that a very low duty ratio is generated at the peak value of the
input voltage. Some authors have presented solutions to solve
this problem, subsequently increasing the cost and complexity
of the ballast: two buck stages in cascade [12], [13], quadratic
and cube buck topologies [14], etc.
Fig. 1(a) shows the proposed solution for designing an HBLED ac–dc driver for replacing incandescent bulb lamps: the
tapped-inductor buck PFC. This topology is very simple and
uses few components. Furthermore, it introduces higher duty
cycles [see dTIB in Fig. 1(c)] than the traditional buck converter
[see dB in Fig. 1(c)] by means of the turn ratio (n1 and n2 )
of its tapped inductor when a high-step-down ratio is needed
[15]–[17]. Therefore, its efficiency is higher than traditional
buck PFC for high-step-down ratio applications and it will be
higher than the converters belonging the flyback family of PFCs.
In order not to penalize the efficiency at these power levels and to
obtain a good tradeoff between switching losses and conduction
losses, the tapped-inductor buck converter will operate in BCM.
In the last years, this operation mode has been widely applied in
ac–dc topologies for increasing the efficiency of the HB-LED
driver [18]–[20]. Also, as can be seen in Fig. 1(b), the topology
can be modified to connect the source terminal of the main
switch to ground.
The buck PFC is known not to shape the line current
ig (t) around the zero crossing of the line voltage (v g (t) =
v gp |sin(ω L t)|, where ω L is the line angular frequency), i.e.,
during the time intervals when the line voltage is lower than
the output voltage v O [21], [22]. These intervals, (0,θ) and (π
− θ,π) shown in Fig. 1(c), contribute to increasing the total
harmonic distortion (THD) of the input current. Therefore, both
a static analysis and an input current distortion analysis of the
tapped-inductor buck PFC operating in BCM will be carried out
in this section.
Faraday’s law applied to both the transistor ton and the diode
toff conduction periods yields
vg (t) − vO
n1 + n2
vO
φm ax (t) = toff
n2
φm ax (t) = ton
(1)
(2)
where φm ax (t) is the maximum tapped-inductor magnetic flux
in a switching period. Equations (1) and (2) are only valid when
the tapped-inductor buck PFC can shape the input current, i.e.,
θ < ω L t < π − θ. The expression of θ can be easily calculated
vO
θ = sin−1
.
(3)
vg p
Using (1), (2) and the condition of BCM (1/fS (t) = ton +
toff ), the switching frequency can be expressed as follows:
fS (t) =
1
m(n + 1)
λ(ωL t)
=
m + n| sin(ωL t)| ton
ton
(4)
where m is the ratio of v O /v gp , n = n2 /n1 and λ(ω L t) is the
normalized switching frequency. If Faraday’s law is applied to
the transistor conduction period ton , then the expression of the
peak value of input current for a specific switching period can
be rewritten as follows:
ig p (t) =
vg (t) − vO
vg p
ton =
[| sin(ωL t)| − m] · ton
L
L
(5)
LAMAR et al.: TAPPED-INDUCTOR BUCK HB-LED AC–DC DRIVER OPERATING IN BOUNDARY CONDUCTION MODE
Fig. 2.
Normalized line current versus ω L t for different values of m and n.
where L is the inductance corresponding to the complete inductor (see Fig. 1). Using (4) and (5), the average input current in a
switching period is
fS (t)
vg p
ton ig p (t) =
ig avg (t) =
λ(ωL t)(| sin(ωL t)| − m)
2
xL
m(n + 1)
vg p
(| sin(ωL t)| − m)
=
xL m + n| sin(ωL t)|
(6)
where xL = 2 L/ton is a fictitious impedance used to perform this
static analysis. As can be seen in (6), the average input current
is nonsinusoidal. Moreover, as already stated, the expression of
the average input current is only valid when the tapped-inductor
buck PFC can shape the input current, i.e., θ < ω L t < π − θ.
When the line input voltage is lower than v O , the input current
becomes zero and its THD increases. However, ig avg (t) can be
highly sinusoidal for certain values of m and n (see Fig. 2). As
can be seen in (6), the shape of ig avg (t) depends solely on m and
n. As the limits imposed by standard regulations are normalized
to the power processed [5]–[7], only certain values of n and m
are instrumental in achieving compliance. The expression of the
PF and of the THD of ig avg (t) can be obtained from (6). In this
case, transcendent equations are obtained, which must be solved
numerically. Fig. 3(a) and (b), respectively, shows the plot of
PF and THD versus m and n. Even if m increases, the distortion
of the input current remains slight for realistic values of n.
In the case of the ENERGY STAR program, Fig. 3(a) shows
that the PF is higher than 0.9 in all designs. However, for Class
C (EN 61000-3-2) regulations, the limit imposed on the third
harmonic depends on the PF and on the rms value of the first
harmonic of the line current. The rest of the limits for all harmonics are imposed by the rms value of the input current with
respect to the rms value for the first harmonic. A transcendent
inequation can be obtained for every harmonic from (6) and
the Class C limits. Fig. 4(a)–(d) shows these inequations solved
numerically as areas of compliance of Class C regulations for
the third, fifth, seventh, and ninth harmonic for different designs
(m and n). As can be seen, the third and fifth harmonics impose
the restriction of compliance with Class C regulations. Finally,
Fig. 4(e) shows the area of compliance for the entire harmonic
content for different designs.
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Fig. 3. (a) PF for different values of m and n. (b) THD for different values of
m and n.
Following the static study, xL can be calculated for a specific
value of the power processed pG by the PFC using (6):
xL =
vO2
μ
PG
(7)
where μ is
π −θ /ω L
m(n + 1)(| sin(ωL t)| − m)
| sin(ωL t)|dt.
m + n| sin(ωL t)|
θ /ω L
(8)
Fig. 5 shows the values of μ for different m and n values.
Finally, the switching frequency can be rewritten using (2), (4),
and the expression of xL
μ=
ωL
m2 π
fS (t) =
λ(ωL (t))
v2
= O λ(ωL (t)μ.
ton
2LpG
(9)
Fig. 6 shows the normalized switching frequency λ(ω L t) versus
the line angle for different m and n designs.
III. DESIGN PROCEDURE FOR A TAPPED-INDUCTOR BUCK PFC
The design procedure for a tapped-inductor buck PFC operating in BCM is very simple. The inputs for this design are
the output voltage v O the peak value of the input voltage v gp
(v gp m ax , v gp nom , and v gp m in ), the output power pG and the line
frequency. The steps can be summarized as follows:
STEP 1. Choose n according to a tradeoff between current
and voltage stress in both the power transistor and diode. As in
any PFC, the minimum duty cycle (corresponding to the peak
value of the input voltage and current) should be around 50%,
which means a reasonable tradeoff between voltage and current
stress in the aforementioned power devices.
STEP 2. Check that the tapped-inductor buck PFC design
complies with international regulations [EN 6100-3-2, Class C
with Fig. 4(e) and ENERGY STAR program with Fig. 3(a)],
i.e., m and n verification.
STEP 3. Choose the value of L. This value should be chosen
so as to guarantee both that the maximum switching frequency
(at intervals around zero crossing of the input voltage) will not
be too high and that the minimum switching frequency (at the
peak value of the line voltage) will not be too low (see Fig. 6).
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 10, OCTOBER 2012
Fig. 4. Area of compliance with Class C EN 61000-3-2 regulations for the (a) third harmonic, (b) fifth harmonic, (c) seventh harmonic, (d) ninth harmonic, and
(e) entire harmonic content.
Fig. 5.
Fig. 6.
and n.
Normalized switching frequency versus ω L t for different values of m
Fig. 7.
σ for different values of m and n.
μ for different values of m and n.
The maximum and the minimum switching frequency can be
easily calculated from (7)
fS
m ax
fS
m in
π
v2
= fS t =
(10)
= O μ
2ωL
2LpG
v2
θ
v 2 m(n + 1)
μ= O σ
= fS t =
= O
ωL
2LpG m + n
2LpG
(11)
where σ is a dimensionless parameter. Fig. 7 shows the value
of σ for different designs. The maximum switching frequency
must be calculated for mm in (v O /v gp m ax ) and the minimum
switching frequency for mm ax (v O /v gp m in ).
the average output power
IV. AVERAGE SMALL-SIGNAL MODEL OF AN AC–DC
TAPPED-INDUCTOR BUCK PFC OPERATING IN BCM
The average small-signal model of the tapped-inductor buck
PFC operating in BCM can be obtained following a similar
process to that proposed in [20]. The value of the dc component
of the output current io dcr can be easily obtained from (5) and
(6), assuming the balance between the average input power and
iO dcr =
pG
vO
vO ω L
=
μ=
vO
xL
xL m2 π
π −θ /ω L
m(n + 1)(| sin(ωL t)| − m)| sin(ωL t)|
dt.
×
m + n| sin(ωL t)|
θ /ω L
(12)
LAMAR et al.: TAPPED-INDUCTOR BUCK HB-LED AC–DC DRIVER OPERATING IN BOUNDARY CONDUCTION MODE
The value of the dc component of the input current ig dcr can
likewise be obtained from (6)
vg p ωL π −θ /ω L m(n + 1)(| sin(ωL t)| − m)
ig dcr =
dt.
xL π θ /ω L
m + n| sin(ωL t)|
(13)
Following the procedure presented in [23], (9) and (13) must
be perturbed in order to obtain the small-signal model. The
notation used in this section to describe the different variables
the usual one, i.e., upper-case letters have been used to describe
steady-state quantities, whereas lower-case letters with hats have
been employed for the perturbations of the same quantities.
Perturbing (12) and (13) yields the following expressions:
∂ig dcr ∂ig dcr v̂g p +
(14)
t̂on
îg dcr =
∂vg p P
∂ton P
∂iodcr ∂iodcr ∂iodcr v̂
+
+
v̂O . (15)
îodcr =
t̂
gp
on
∂vg p P
∂ton P
∂vO P
However, as can be seen in (12) and (13), the derivative of
both ig dcr and io dcr with respect to v gp , v O , and ton makes this
average small signal model impractical. A simplification will
hence be carried out. A perfect sinusoidal line waveform is,
therefore, assumed as the average input current. The peak value
of this simplified input current is assumed to be the peak value
of (6). Using (6) and the expression of xL , the simplified ig avg
yields
vg p ton m(n + 1)
(1 − m)| sin(ωL t)|
2L
m+n
where its dc value after replacing m with v O /vgp is
ig agv (ωL t) =
ig dc =
ton vO (vg p − vO )
(n + 1).
πL (vO + nvg p )
pg (ωL t) = ig avg (ωL t)vg p | sin(ωL t)|.
(18)
vg p ton m(n + 1)
(1 − m)(1 − cos(2ωL t)) (20)
4L
m+n
where its dc value after replacing m with v O /vgp is
iO (ωL t) =
vg p ton (vg p − vO )
(n + 1).
4L (vO + nvg p )
where
∂ig dc Ton VO2 (N + 1)2
=
∂vg p P
πL (VO + N Vg p )2
=
(21)
In order to prove the validity of the previously adopted simplification, Fig. 8(a) shows the quotient between the real value of
the average input current ig dcr and the simplified value ig dc . In
addition, Fig. 8(b) shows the quotient between the real value of
the average output current iO dcr and the simplified value iO dc .
Ton V M 2 (N + 1)2
1
=
2
πL (M + N )
ri
(23)
VO (N + 1)(1 − M )
= gig on .
πL
(M + N )
(24)
∂ig dc VO (N + 1)(Vg p − VO )
=
∂ton P
πL (VO + N Vg p )2
=
Perturbing the value of iOdc and taking into account (18)
∂iO dc ∂iO dc ∂iO dc v̂g p +
v̂O (25)
t̂on +
îodc =
∂vg p P
∂ton P
∂vO P
where
Ton (N + 1)(2Vg p VO + N Vg2p − VO2 )
∂iO dc =
∂vg p P
4L(VO + N Vg p )2
(19)
After establishing the balance between pg (ω L t) and pO (ω L t),
the expression of iO (ω L t) can be easily calculated
iO dc =
As can be seen, the simplified and real values of average currents
are very close for values of m around 0.1 (which is an interesting
value for a practical design).
Continuing with the small-signal model, (17) is perturbed
∂ig dc ∂ig dc v̂g p +
(22)
t̂on
îg dc =
∂vg p P
∂ton P
(17)
The pulsating output power (the power delivered by the PFC)
can be obtained by multiplying the output voltage v O by the
current iO (t) injected by the power stage into the output cell
made up of the bulk capacitor CO and the HB-LEDs
pO (ωL t) = iO (ωL t)vO .
Fig. 8. (a) ig d c /ig d c r values for different values of m and n. (b) iO d c /iO d c r
values for different values of m and n.
(16)
The pulsating input power pg (ω L t) can be obtained by multiplying the values of v g (ω L t) and ig avg (ω L t)
4333
=
Ton (N + 1)(2M + N − M 2 )
= giO g (26)
4L(M + N )2
∂iO dc Vg p (N + 1)(Vg p − VO )
=
∂ton P
4L(VO + N Vg p )
=
Vg p (N + 1)(1 − M )
= giO on
4L(M + N )2
(27)
Ton (N + 1)
1
=
.
4L(M + N )2
rO
(28)
∂iO dc Ton (N + 1)Vg p
=
∂vg p P
4L(VO + N Vg p )2
=
The relationships obtained between the perturbed variables
can be summarized in the small-signal circuit shown in Fig. 9(a).
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 10, OCTOBER 2012
Fig. 9. (a) Small-signal model of the ac–dc tapped-inductor buck converter operating in BCM. (b) Small-signal model of the ac–dc tapped-inductor buck converter
operating in BCM with output capacitor and HB-LEDs.
From the small-signal circuit of Fig. 9(b) with the output filter
capacitor and the HB-LEDs connected, the input–output transfer
function can be derived by inspection to give
rO
1
îLEDs
= giO g
v̂g p
rO + rd 1 + (rO rd /rO + rd )CO s
(29)
where rd is the dynamic resistance of the HB-LED string. The
control-output transfer function can likewise be derived by inspection to give
rO
1
îLEDs
.
= giO on
rO + rd 1 + (rO rd /rO + rd )CO s
t̂on
(30)
V. EXPERIMENTAL RESULTS
A. Verification of the Static Analysis
A prototype of a tapped-inductor buck PFC used as an ac–
dc HB-LED driver was built and tested. It was controlled to
operate in BCM using a commercial IC (NCL30000 by ON
Semiconductors).
The circuit was built according to the scheme given in Fig. 10,
in which the output current is controlled (instead of the output
voltage). The resistors RS , RF 1 , RCL 1 , and RCL 2 and the capacitors of CF 1 , CCL 1 , and CCL 2 define the output current
feedback loop. The converter output is connected to a string
of 7 LXK2PW14T00 HB-LEDs (Luxeon). The rated operating
conditions of this converter are v g RM S = 90–130 V, iLEDsdc =
0.5 A, pG = 12.5 W, v O = 22.5 V. Other parameters values are
L = 600 μF, Co = 1000 μF and n = 0.8.
Fig. 11 shows the line current waveforms in the prototype
shown in Fig. 10. As can be seen, these waveforms are highly
sinusoidal. Furthermore, theoretical results (in magenta) match
experimental results and the experimental values of the PF and
the THD are very close to theoretical values (see Table I). Therefore, the compliance√with international regulations is assured
(check m = 22.5/110 2 = 1.28 and n = 0.8 values at Fig. 3(a)
for ENERGY STAR regulations and Fig. 4(e) for EN 61000-3-2
Class C regulations). The rectified versions of these input current
waveforms also correspond to the average value of the current
passing through the transistor (averaged over a switching period). However, it should be noted that the switching frequency
components of this current have been removed by the EMI filter
placed at the input of the line rectifier (see Fig. 10). The values
of the EMI filter components are Lf = 1.5 mH, Cf 1 = 1.8 nF
and Cf 2 = 100 nF. Fig. 12 shows the output voltage and the
current through the string of HB-LEDs. As you can be seen, a
significative low frequency ripple appears in the output current.
Fig. 10.
IC.
Implementation of the tapped-inductor buck PFC with NCL30000
The only way to reduce this ripple is to increase the capacitance
of the output capacitor. Therefore, the electrolytic capacitor cannot be removed and the lifespan of the ac–dc HB-LED driver is
reduced.
Fig. 13 shows the efficiency of the prototype versus the line
voltage. As can be seen, the efficiency is around 90% for all
input voltages. This efficiency is greater than solutions based
on converters belonging the flyback family of PFCs even using
dissipative snubbers in order to reduce the ringing in both the
MOSFET and the diode (gray color in Fig. 10).
Fig. 14 shows the experimental waveforms of the line current obtained in analog dimming operations when the current
across the HB-LED string is 0.5, 0.4, and 0.3 A. As these figures
show, the experimental PF is very high under all operating conditions. The experimental results have also been compared with
theoretical values (in magenta). As can be seen, the theoretical
waveforms match their experimental counterparts.
The prototype was tested until both the prototype tempera√
ture and HB-LED temperature stabilized (v gp = 110 2 and
iLEDsdc = 0.5 A). The final operating temperature was reached
after 90 min of operation. During the warm-up process, the voltage across the HB-LEDs decreases. However, this decrease is
minimal: from 22.56 V at the beginning of the warm-up process
to 22.24 V at the end of this process. The PF, thus, remains constant: from 0.993 V at the beginning of the warm-up process to
0.992 V at the end of this process.
LAMAR et al.: TAPPED-INDUCTOR BUCK HB-LED AC–DC DRIVER OPERATING IN BOUNDARY CONDUCTION MODE
Fig. 11.
4335
Line current for different input voltages.
TABLE I
EXPERIMENTAL PF AND THD VERSUS THEORETICAL PF AND THD
Fig. 13.
Efficiency versus line voltage.
reasons, the efficiency of this prototype is lower than in the
previous design.
B. Verification of the Small-Signal Model
Fig. 12.
Output voltage and current through the HB-LEDs.
Finally, in order to check the validity of this solution with universal input voltage, a new tapped-inductor was designed using
the same core as in the previous design (E20 size and 3F3 material). The main characteristics of this new design are: L = 3 mH
and n = 0.25. Furthermore, the voltage rating of the MOSFET
was increased (from 400 to 600 V). However, the voltage rating
of the diode was the same as the one in the previous design
(200 V). Fig. 15 shows the line current waveforms in the prototype shown in Fig. 10 with the aforementioned modifications.
As can be seen in this new design, the waveforms are highly
sinusoidal. Moreover, theoretical results (in magenta) match experimental results and the experimental values of the PF and
the THD are very close to theoretical values (see Table II). In
this case, the compliance with√international regulations is
√also
assured (check m = 22.5/110 2 = 1.28, m = 22.5/230 2 =
0.069, and n = 0.25 values at Fig. 3(a) for ENERGY STAR regulations and Fig. 4(e) for Class C EN 61000-3-2 regulations).
Only waveforms at 220 and 265 Vrm s line input voltage are
shown because the results at 110 and 90 Vrm s input voltage are
very similar to the previous design shown in Tables I and II.
Fig. 16 shows the efficiency of this new design versus the
input voltage. Even using higher voltage MOSFETs (600-V
MOSFET) and dissipative snubbers, the efficiency is greater
than the solutions based on flyback PFCs. For aforementioned
The small-signal model can be verified by comparing the
output-voltage transient response of the experimental prototype
with that predicted by the small-signal equivalent circuit shown
in Fig. 9(b), using a resistive load instead of an HB-LED string
in order to simplify the experiment. The 12.5-W prototype used
in the previous section was used as an example with √
the following values of the parameters: Vo = 20, Vg p = 110 2, Ton =
6.2 ms, N = 1.25, L = 600 μH, rd = 3.33 Ω, and Co = 1000 μF.
Using the small circuit of Fig. 9(b), the time-domain response
for the output voltage for a step of the peak value of the input
voltage ΔVgp is
rO · rd
ΔVg p 1 − e−1/(r O r d /r O +r d )C O .
rO + rd
(31)
Similarly, for a step control change ΔTon , the time-domain
response for the output voltages is
vo (t) = giO g p
rO · rd
ΔTon 1 − e−(1/(r O r d /r O +r d )C O ) .
rO + rd
(32)
The results of the experimental prototype (in blue) and the prediction of the small-signal model (in red) are shown in Fig. 17(a)
for a step of the peak value of the input voltage of 14.14 V and
in Fig. 17(b) for a step control change of 1.8 μs. The excellent
agreement of the results confirms the validity of the small-signal
model.
vo (t) = giO on
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 10, OCTOBER 2012
Fig. 14.
Line current during analog dimming operations.
VI. CONCLUSION
Fig. 15.
design).
Line current for different input voltages (Universal input voltage
TABLE II
EXPERIMENTAL PF AND THD VERSUS THEORETICAL PF AND THD
(UNIVERSAL INPUT VOLTAGE DESIGN)
This paper presents a low-cost topology with a high-stepdown ratio based on a tapped-inductor buck converter operating
in BCM. This low-cost solution presents a suitable high-stepdown ratio without galvanic isolation so as to produce an output
voltage of about 20 V from line voltage. As has been demonstrated, the tapped-inductor buck PFC also maintains both high
efficiency and low cost and size in comparison with other PFCs
based on one stage (e.g., flyback PFC). Although the input
current demanded to the mains is not sinusoidal, the analysis
carried out shows that the converter complies with standards if
it is properly designed. However, the proposed solution presents
a main drawback: the electrolytic capacitor cannot be removed.
Nevertheless, this is the price to pay for a very low-cost solution.
All these conclusions have been proved on a 12-W experimental
prototype.
The tapped-inductor buck PFC operating in BCM, thus, seems
to be an attractive option in the case of ac–dc HB-LED drivers
for replacing incandescent bulb lamps.
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Fig. 16.
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Efficiency versus line voltage (both US and universal input voltage
Fig. 17 Time-domain response of the ouput voltage for a (a) peak value of the
input voltage step and (b) step control change.
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Diego G. Lamar (M’08) was born in Zaragoza,
Spain, in 1974. He received the M.Sc. and Ph.D. degrees in electrical engineering from the University of
Oviedo, Gijón, Spain, in 2003 and 2008, respectively.
In 2003 and 2005, he became a Research Engineer and an Assistant Professor, respectively, at the
University of Oviedo, where since September 2011,
he has been an Associate Professor. His research interests include switching-mode power supplies, converter modeling, and power-factor-correction converters.
4337
Marcos Fernandez Diaz (S’11) was born in Aviles,
Spain, in 1986. He received the M.Sc. degree in
telecommunications engineering from the University
of Oviedo, Gijón, Spain, in 2011.
He has been with the Department of Electrical
and Electronic Engineering, Computers and Systems,
University of Oviedo, for the Power Supply System Group since 2011. His research interests include
power factor corrector ac–dc converters for LED
lighting.
Manuel Arias Pérez de Azpeitia (M’10) was born
in Oviedo, Spain, in 1980. He received the M.Sc. degree in electrical engineering from the University of
Oviedo, Gijón, Spain, in 2005, and the Ph.D. degree
in the same university in 2010.
Since February 2005, he has been a Researcher
in the Department of Electrical and Electronic Engineering, University of Oviedo, developing electronic
systems for UPSs and electronic switching power
supplies. Since February 2007, he has also been an
Assistant Professor of electronics in the same University. His research interests include dc–dc converters, dc–ac converters, and
UPS.
Marta M. Hernando (M’94–SM’11) was born in
Gijón, Spain, in 1964. She received the M.S. and
Ph.D. degrees in electrical engineering from the University of Oviedo, Gijón, Spain, in 1988 and 1992,
respectively.
She is currently a Professor at the University of
Oviedo. Her main interests include switching-mode
power supplies and high-power factor rectifiers.
Javier Sebastián (M’87–SM’11) was born in
Madrid, Spain, in 1958. He received the M.Sc. degree
from the Polytechnic University of Madrid, Madrid,
in 1981, and the Ph.D. degree from the Universidad
de Oviedo, Gijón, Spain, in 1985.
He was an Assistant Professor and an Associate Professor at both the Polytechnic University of
Madrid and the Universidad de Oviedo. Since 1992,
he has been with the Universidad de Oviedo, where
he is currently a Professor. His research interests include switching-mode power supplies, modeling of
dc-to-dc converters, low-output-voltage dc-to-dc converters, and high-powerfactor rectifiers.