14-Bit, 80 MSPS, 3 V A/D Converter
AD9245
FUNCTIONAL BLOCK DIAGRAM
FEATURES
Single 3 V supply operation (2.7 V to 3.6 V)
SNR = 72.7 dBc to Nyquist
SFDR = 87.6 dBc to Nyquist
Low power: 366 mW
Differential input with 500 MHz bandwidth
On-chip reference and sample-and-hold
DNL = ± 0.5 LSB
Flexible analog input: 1 V p-p to 2 V p-p range
Offset binary or twos complement data format
Clock duty cycle stabilizer
AVDD
DRVDD
AD9245
VIN+
VIN–
MDAC1
SHA
8-STAGE
1 1/2-BIT PIPELINE
4
A/D
16
3
A/D
REFT
REFB
CORRECTION LOGIC
OTR
14
OUTPUT BUFFERS
D13 (MSB)
VREF
D0 (LSB)
APPLICATIONS
High end medical imaging equipment
IF sampling in communications receivers:
WCDMA, CDMA-One, CDMA-2000, TDS-CDMA
Battery-powered instruments
Hand-held scopemeters
Low cost digital oscilloscopes
Power sensitive military applications
GENERAL DESCRIPTION
The AD9245 is a monolithic, single 3 V supply, 14-bit, 80 MSPS
analog-to-digital converter featuring a high performance
sample-and-hold amplifier (SHA) and voltage reference. The
AD9245 uses a multistage differential pipelined architecture
with output error correction logic to provide 14-bit accuracy at
80 MSPS and guarantee no missing codes over the full operating temperature range.
The wide bandwidth, truly differential SHA allows a variety of
user-selectable input ranges and common modes, including
single-ended applications. It is suitable for multiplexed systems
that switch full-scale voltage levels in successive channels, and
for sampling single-channel inputs at frequencies well beyond
the Nyquist rate. Combined with power and cost savings over
previously available analog-to-digital converters, the AD9245 is
suitable for applications in communications, imaging, and
medical ultrasound.
A single-ended clock input is used to control all internal conversion cycles. A duty cycle stabilizer (DCS) compensates for
wide variations in the clock duty cycle while maintaining
SENSE
0.5V
REF
SELECT
AGND
CLOCK
DUTY CYCLE
STABILIZER
CLK
MODE
SELECT
PDWN
MODE DGND
03583-B-001
Figure 1. Functional Block Diagram
excellent overall ADC performance. The digital output data is
presented in straight binary or twos complement formats. An
out-of-range (OTR) signal indicates an overflow condition that
can be used with the most significant bit to determine low or
high overflow. Fabricated on an advanced CMOS process, the
AD9245 is available in a 32-lead LFCSP and is specified over
the industrial temperature range (–40°C to +85°C).
PRODUCT HIGHLIGHTS
1. The AD9245 operates from a single 3 V power supply and
features a separate digital output driver supply to accommodate 2.5 V and 3.3 V logic families.
2. Operating at 80 MSPS, the AD9245 consumes a low 366 mW.
3. The patented SHA input maintains excellent performance for
input frequencies up to 100 MHz, and can be configured for
single-ended or differential operation.
4. The AD9245 is pin compatible with the AD9215, AD9235,
and AD9236. This allows a simplified migration from 10 bits
to 14 bits and 20 MSPS to 80 MSPS.
5. The clock DCS maintains overall ADC performance over a
wide range of clock pulsewidths.
6. The OTR output bit indicates when the signal is beyond the
selected input range.
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.326.8703
© 2003 Analog Devices, Inc. All rights reserved.
AD9245
TABLE OF CONTENTS
AD9245–DC Specifications ............................................................ 3
Analog Input and Reference Overview ................................... 14
AD9245–AC Specifications............................................................. 4
Clock Input Considerations...................................................... 15
AD9245–Digital Specifications....................................................... 5
Jitter Considerations .................................................................. 16
AD9245–Switching Specifications ................................................. 6
Power Dissipation and Standby Mode .................................... 16
Explanation of Test Levels........................................................... 6
Digital Outputs ........................................................................... 16
Absolute Maximum Ratings............................................................ 7
Timing ......................................................................................... 17
Thermal Resistance ...................................................................... 7
Voltage Reference ....................................................................... 17
ESD Caution.................................................................................. 7
Internal Reference Connection ................................................ 17
Definitions of Specifications ........................................................... 8
External Reference Operation .................................................. 18
Pin Configuration and Functional Descriptions.......................... 9
Operational Mode Selection ..................................................... 18
Equivalent Circuits ......................................................................... 10
Evaluation Board ........................................................................ 18
Typical Performance Characteristics ........................................... 11
Outline Dimensions ....................................................................... 25
Theory of Operation ...................................................................... 14
Ordering Guide .......................................................................... 25
REVISION HISTORY
Revision B
10/03—Data Sheet Changed from REV. A to REV. B
Changes to Figure 33 ..................................................................... 17
5/03—Data Sheet Changed from REV. 0 to REV. A
Changes to Figure 30 .................................................................... 15
Changes to Figure 37 ..................................................................... 19
Changes to Figure 38..................................................................... 20
Changes to Figure 39...................................................................... 21
Changes to Table 10 ....................................................................... 24
Changes to the ORDERING GUIDE........................................... 25
Rev. B | Page 2 of 28
AD9245
AD9245–DC SPECIFICATIONS
Table 1. AVDD = 3 V, DRVDD = 2.5 V, Sample Rate = 80 MSPS, 2 V p-p Differential Input, 1.0 V External Reference, unless
otherwise noted
AD9245BCP
Typ
Parameter
Temp
Test Level
RESOLUTION
ACCURACY
No Missing Codes
Offset Error1
Gain Error
Gain Error1
Differential Nonlinearity (DNL)2
Integral Nonlinearity (INL)2
TEMPERATURE DRIFT
Offset Error1
Gain Error
Gain Error1
INTERNAL VOLTAGE REFERENCE
Output Voltage Error (1 V Mode)
Load Regulation @ 1.0 mA
Output Voltage Error (0.5 V Mode)
Load Regulation @ 0.5 mA
INPUT REFERRED NOISE
VREF = 0.5 V
VREF = 1.0 V
ANALOG INPUT
Input Span, VREF = 0.5 V
Input Span, VREF = 1.0 V
Input Capacitance3
REFERENCE INPUT RESISTANCE
POWER SUPPLIES
Supply Voltage
AVDD
DRVDD
Supply Current
IAVDD2
IDRVDD2
PSRR
POWER CONSUMPTION
Low Frequency Input4
Standby Power5
Full
VI
Full
Full
25°C
Full
Full
Full
VI
VI
V
VI
VI
VI
Guaranteed
±0.30
±0.28
±0.70
±0.5
±1.4
Full
Full
Full
V
V
V
±10
±12
±17
Full
25°C
25°C
25°C
VI
V
V
V
±3
±2
±6
±1
25°C
25°C
V
V
1.86
1.17
LSB rms
LSB rms
Full
Full
Full
Full
IV
IV
V
V
1
2
7
7
V p-p
V p-p
pF
kΩ
Full
Full
IV
IV
Full
25°C
25°C
25°C
25°C
Min
14
Unit
Bits
±1.2
±4.16
±1.0
±5.15
% FSR
% FSR
% FSR
LSB
LSB
ppm/°C
ppm/°C
ppm/°C
±34
mV
mV
mV
mV
3.0
2.5
3.6
3.6
V
V
VI
V
V
122
9
±0.01
138
mA
mA
% FSR
V
V
366
1.0
1
2.7
2.25
Max
mW
mW
With a 1.0 V internal reference.
Measured at the maximum clock rate, fIN = 2.4 MHz, full-scale sine wave, with approximately 5 pF loading on each output bit.
3
Input capacitance refers to the effective capacitance between one differential input pin and AGND. Refer to Figure 4 for the equivalent analog input structure.
4
Measured at AC Specification conditions without output drivers.
5
Standby power is measured with a dc input, CLK pin inactive (i.e., set to AVDD or AGND).
2
Rev. B | Page 3 of 28
AD9245
AD9245–AC SPECIFICATIONS
Table 2. AVDD = 3 V, DRVDD = 2.5 V, Sample Rate = 80 MSPS, 2 V p-p Differential Input, 1.0 V External Reference,
AIN = –0.5 dBFS, DCS Off, unless otherwise noted
Parameter
SIGNAL-TO-NOISE RATIO (SNR)
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
SIGNAL-TO-NOISE AND DISTORTION (SINAD)
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
EFFECTIVE NUMBER OF BITS (ENOB)
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
WORST SECOND OR THIRD
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
SPURIOUS-FREE DYNAMIC RANGE (SFDR)
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
Temp
Test Level
Min
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
IV
V
V
71.1
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
IV
V
V
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
IV
V
V
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
IV
V
V
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
IV
V
V
Rev. B | Page 4 of 28
AD9245BCP
Typ
Max
dB
dB
dB
dB
dB
dB
73.3
72.7
70.5
71.7
70.2
70.7
dB
dB
dB
dB
dB
dB
73.2
72.5
69.9
71.2
69.6
11.5
Bits
Bits
Bits
Bits
Bits
Bits
11.9
11.8
11.3
11.5
11.3
–76.5
–92.8
–87.6
–75.7
–81.6
–79.0
76.5
92.8
87.6
75.7
81.6
79.0
Unit
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
AD9245
AD9245–DIGITAL SPECIFICATIONS
Table 3. AVDD = 3 V, DRVDD = 2.5 V, 1.0 V External Reference, unless otherwise noted
Parameter
LOGIC INPUTS (CLK, PDWN)
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Capacitance
DIGITAL OUTPUT BITS (D0–D13, OTR)1
DRVDD = 3.3 V
High Level Output Voltage (IOH = 50 µA)
High Level Output Voltage (IOH = 0.5 mA)
Low Level Output Voltage (IOH = 1.6 mA)
Low Level Output Voltage (IOH = 50 µA)
DRVDD = 2.5 V
High Level Output Voltage (IOH = 50 µA)
High Level Output Voltage (IOH = 0.5 mA)
Low Level Output Voltage (IOH = 1.6 mA)
Low Level Output Voltage (IOH = 50 µA)
1
Temp
Test Level
Min
Full
Full
Full
Full
Full
IV
IV
IV
IV
V
2.0
Full
Full
Full
Full
IV
IV
IV
IV
3.29
3.25
Full
Full
Full
Full
IV
IV
IV
IV
2.49
2.45
Output voltage levels measured with 5 pF load on each output.
Rev. B | Page 5 of 28
AD9245BCP
Typ
Max
0.8
+10
+10
–10
–10
2
Unit
V
V
µA
µA
pF
0.2
0.05
V
V
V
V
0.2
0.05
V
V
V
V
AD9245
AD9245–SWITCHING SPECIFICATIONS
Table 4. AVDD = 3 V, DRVDD = 2.5 V, unless otherwise noted
Parameter
CLOCK INPUT PARAMETERS
Maximum Conversion Rate
Minimum Conversion Rate
CLK Period
CLK Pulsewidth High1
CLK Pulsewidth Low1
DATA OUTPUT PARAMETERS
Output Propagation Delay (tPD)2
Pipeline Delay (Latency)
Aperture Delay (tA)
Aperture Uncertainty (Jitter, tJ)
Wake-Up Time3
OUT-OF-RANGE RECOVERY TIME
1
2
3
Temp
Test Level
Min
Full
Full
Full
Full
Full
VI
V
V
V
V
80
Full
Full
Full
Full
Full
Full
V
V
V
V
V
V
AD9245BCP
Typ
Max
Unit
MSPS
MSPS
ns
ns
ns
1
12.5
4.6
4.6
4.2
7
1
0.3
7
2
ns
Cycles
ns
ps rms
ms
Cycles
With duty cycle stabilizer (DCS) enabled.
Output propagation delay is measured from CLK 50% transition to DATA 50% transition, with 5 pF load.
Wake-up time is dependant on the value of the decoupling capacitors; typical values shown with 0.1 µF and 10 µF capacitors on REFT and REFB.
N
N+1
N+2
N–1
tA
ANALOG
INPUT
N+8
N+3
N+7
N+4
N+5
N+6
CLK
DATA
OUT
N–9
N–8
N–7
N–6
N–5
N–4
N–3
N–2
N–1
N
tPD = 6.0ns MAX
2.0ns MIN
03583-B-002
Figure 2. Timing Diagram
EXPLANATION OF TEST LEVELS
Test Level
I
II
III
IV
V
VI
Definitions
100% production tested.
100% production tested at 25°C and guaranteed by design and characterization at specified temperatures.
Sample tested only.
Parameter is guaranteed by design and characterization testing.
Parameter is a typical value only.
100% production tested at 25°C and guaranteed by design and characterization for industrial temperature range.
Rev. B | Page 6 of 28
AD9245
ABSOLUTE MAXIMUM RATINGS
THERMAL RESISTANCE
Table 5. AD9245 Absolute Maximum Ratings
Parameter
With Respect to
ELECTRICAL
AVDD
AGND
DRVDD
DGND
AGND
DGND
AVDD
DRVDD
D0–D13
DGND
CLK, MODE AGND
VIN+, VIN–
AGND
VREF
AGND
SENSE
AGND
REFT, REFB
AGND
PDWN
AGND
ENVIRONMENTAL
Storage Temperature
Operating Temperature Range
Lead Temperature Range
(Soldering 10 sec)
Junction Temperature
Min
Max
Unit
–0.3
–0.3
–0.3
–3.9
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
+3.9
+3.9
+0.3
+3.9
DRVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
V
V
V
V
V
V
V
V
V
V
V
–65
–40
+125
+85
°C
°C
300
150
°C
°C
θJA is specified for the worst-case conditions on a 4-layer board
in still air, in accordance with EIA/JESD51-1.
Table 6. Thermal Resistance
Package Type
CP-32
θJA
32.5
θJC
32.71
Unit
°C/W
Airflow increases heat dissipation, effectively reducing θJA.
Also, more metal directly in contact with the package leads
from metal traces, through holes, ground, and power planes
reduces the θJA. It is recommended that the exposed paddle be
soldered to the ground plane for the LFCSP package. There is an
increased reliability of the solder joints, and maximum thermal
capability of the package is achieved with the exposed paddle
soldered to the customer board.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only and functional operation of the device at these or
any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. B | Page 7 of 28
AD9245
DEFINITIONS OF SPECIFICATIONS
Analog Bandwidth (Full Power Bandwidth)—The analog
input frequency at which the spectral power of the fundamental
frequency (as determined by the FFT analysis) is reduced by 3 dB.
Aperture Delay (tA)—The delay between the 50% point of the
rising edge of the clock and the instant at which the analog
input is sampled.
Aperture Uncertainty (Jitter, tJ)—The sample-to-sample variation in aperture delay.
Integral Nonlinearity (INL)—The deviation of each individual
code from a line drawn from negative full scale through positive
full scale. The point used as negative full scale occurs ½ LSB
before the first code transition. Positive full scale is defined as a
level 1½ LSB beyond the last code transition. The deviation is
measured from the middle of each particular code to the true
straight line.
Differential Nonlinearity (DNL, No Missing Codes)—An
ideal ADC exhibits code transitions that are exactly 1 LSB apart.
DNL is the deviation from this ideal value. Guaranteed no missing codes to 14-bit resolution indicates that all 16384 codes
must be present over all operating ranges.
Offset Error—The major carry transition should occur for an
analog value ½ LSB below VIN+ = VIN–. Offset error is
defined as the deviation of the actual transition from that point.
Effective Number of Bits (ENOB)—The effective number of
bits for a sine wave input at a given input frequency can be calculated directly from its measured SINAD using the following
formula:
ENOB =
(SINAD − 1.76 )
6.02
Signal-to-Noise Ratio (SNR)1 —The ratio of the rms input
signal amplitude to the rms value of the sum of all other spectral components below the Nyquist frequency, excluding the
first six harmonics and dc.
Spurious-Free Dynamic Range (SFDR)1—The difference in dB
between the rms input signal amplitude and the peak spurious
signal. The peak spurious component may or may not be a
harmonic.
Two-Tone SFDR1—The ratio of the rms value of either input
tone to the rms value of the peak spurious component. The
peak spurious component may or may not be an IMD product.
Clock Pulsewidth and Duty Cycle—Pulsewidth high is the
minimum amount of time that the clock pulse should be left in
the Logic 1 state to achieve rated performance. Pulsewidth low
is the minimum time the clock pulse should be left in the
Logic 0 state. At a given clock rate, these specifications define an
acceptable clock duty cycle.
Gain Error—The first code transition should occur at an
analog value ½ LSB above negative full scale. The last transition
should occur at an analog value 1½ LSB below the positive
full scale. Gain error is the deviation of the actual difference
between first and last code transitions and the ideal difference
between first and last code transitions.
Minimum Conversion Rate—The clock rate at which the SNR
of the lowest analog signal frequency drops by no more than
3 dB below the guaranteed limit.
Temperature Drift—The temperature drift for offset error and
gain error specifies the maximum change from the initial
(25°C) value to the value at TMIN or TMAX.
Output Propagation Delay (tPD)—The delay between the clock
rising edge and the time when all bits are within valid logic
levels.
Power Supply Rejection Ratio—The change in full scale from
the value with the supply at the minimum limit to the value
with the supply at its maximum limit.
Out-of-Range Recovery Time—The time it takes for the ADC
to reacquire the analog input after a transition from 10% above
positive full scale to 10% above negative full scale, or from 10%
below negative full scale to 10% below positive full scale.
Total Harmonic Distortion (THD)1—The ratio of the rms
input signal amplitude to the rms value of the sum of the first
six harmonic components.
Maximum Conversion Rate—The clock rate at which parametric testing is performed.
1
AC specifications may be reported in dBc (degrades as signal levels are
lowered) or in dBFS (always related back to converter full scale).
Signal-to-Noise and Distortion (SINAD)1—The ratio of the
rms input signal amplitude to the rms value of the sum of all
other spectral components below the Nyquist frequency, including harmonics but excluding dc.
Rev. B | Page 8 of 28
AD9245
25 REFB
26 REFT
27 AVDD
28 AGND
29 VIN+
30 VIN–
31 AGND
32 AVDD
PIN CONFIGURATION AND FUNCTIONAL DESCRIPTIONS
DNC 1
24 VREF
CLK 2
23 SENSE
DNC 3
22 MODE
AD9245
PDWN 4
21 OTR
CSP
(LSB) D0 5
20 D13 (MSB)
TOP VIEW
(Not to Scale)
D1 6
19 D12
DRVDD 16
DGND 15
D9 14
D8 13
D7 12
D6 11
17 D10
D4 9
18 D11
D3 8
D5 10
D2 7
03583-B-022
Figure 3. 32-Lead LFCSP
Table 7. Pin Function Descriptions—32-Lead LFCSP (CP Package)
Pin No.
1, 3
2
4
5 to 14, 17 to 20
15
16
21
22
23
24
25
26
27, 32
28, 31
29
30
Mnemonic
DNC
CLK
PDWN
D0 (LSB) to D13 (MSB)
DGND
DRVDD
OTR
MODE
SENSE
VREF
REFB
REFT
AVDD
AGND
VIN+
VIN–
Description
Do Not Connect
Clock Input Pin
Power-Down Function Select
Data Output Bits
Digital Output Ground
Digital Output Driver Supply
Out-of-Range Indicator
Data Format Select and DCS Mode Selection (see Table 9)
Reference Mode Selection (see Table 8)
Voltage Reference Input/Output
Differential Reference (–)
Differential Reference (+)
Analog Power Supply
Analog Ground
Analog Input Pin (+)
Analog Input Pin (–)
Rev. B | Page 9 of 28
AD9245
EQUIVALENT CIRCUITS
AVDD
DRVDD
D13-D0,
OTR
VIN+, VIN–
03583-B-005
03583-B-003
Figure 6. Equivalent Digital Output Circuit
Figure 4. Equivalent Analog Input Circuit
AVDD
AVDD
CLK,
PDWN
MODE
20kΩ
03583-B-004
03583-B-006
Figure 5. Equivalent MODE Input Circuit
Figure 7. Equivalent Digital Input Circuit
Rev. B | Page 10 of 28
AD9245
TYPICAL PERFORMANCE CHARACTERISTICS
AVDD = 3.0 V, DRVDD = 2.5 V, Sample Rate = 80 MSPS, DCS Disabled, TA = 25°C, 2 V p-p Differential Input, AIN = –0.5 dBFS,
VREF = 1.0 V External, unless otherwise noted
0
100
AIN = –0.5dBFS
SNR = 73.2dBc
ENOB = 11.8 BITS
SFDR = 92.8 dBc
–10
–20
90
SFDR (dBc)
SNR/SFDR (dBc AND dBFS)
–30
AMPLITUDE (dBFS)
SFDR (dBFS)
–40
–50
–60
–70
–80
–90
SNR (dBFS)
80
70
SFDR = 90dBc
REFERENCE LINE
60
SNR (dBc)
50
–100
–110
–120
0
5
10
15
20
25
FREQUENCY (MHz)
30
35
–20
100
–5
0
03583-B-033
SFDR (dBFS)
90
SFDR (dBc)
SNR/SFDR (dBc AND dBFS)
–30
AMPLITUDE (dBFS)
–20
–15
–10
INPUT AMPLITUDE (dBFS)
Figure 11. Single Tone SNR/SFDR vs. Input Amplitude (AIN) @ 2.5 MHz
AIN = –0.5dBFS
SNR = 72.7dBc
ENOB = 11.8 BITS
SFDR = 87.6 dBc
–10
–25
03583-B-032
Figure 8. Single Tone 8K FFT @ 2.5 MHz
0
40
–30
40
–40
–50
–60
–70
–80
–90
SNR (dBFS)
80
70
SFDR = 90dBc
REFERENCE LINE
60
SNR (dBc)
50
–100
–110
–120
0
5
10
15
20
25
FREQUENCY (MHz)
30
35
–20
–20
–15
–10
INPUT AMPLITUDE (dBFS)
–5
0
03583-B-034
Figure 12. Single Tone SNR/SFDR vs. Input Amplitude (AIN) @ 39 MHz
100
AIN = –0.5dBFS
SNR = 71.7dBc
ENOB = 11.5 BITS
SFDR = 81.6 dBc
–10
–25
03583-B-023
Figure 9. Single Tone 8K FFT @ 39 MHz
0
40
–30
40
SFDR (DIFF)
90
SFDR (SE)
–40
SNR/SFDR (dBc)
AMPLITUDE (dBFS)
–30
–50
–60
–70
–80
SNR (DIFF)
80
70
SNR (SE)
–90
60
–100
–110
–120
0
5
10
15
20
25
FREQUENCY (MHz)
30
35
50
40
03583-B-024
Figure 10. Single Tone 8K FFT @ 70 MHz
0
20
40
60
SAMPLE RATE (MSPS)
80
Figure 13. SNR/SFDR vs. Sample Rate @ 40 MHz
Rev. B | Page 11 of 28
100
03583-B-025
AD9245
100
0
AIN = –6.5dBFS
SNR = 73.4dBFS
SFDR = 86.0dBFS
–10
–20
90
SNR/SFDR (dBc AND dBFS)
–30
AMPLITUDE (dBFS)
SFDR (dBFS)
–40
–50
–60
–70
–80
–90
SFDR (dBc)
80
70
SNR (dBFS)
SFDR = 90dBc
REFERENCE LINE
60
SNR (dBc)
50
–100
–110
–120
0
5
10
15
20
25
FREQUENCY (MHz)
30
35
40
–30
40
0
–9
–6
03583-B-031
SFDR (dBFS)
90
SFDR (dBc)
SNR/SFDR (dBc AND dBFS)
–30
AMPLITUDE (dBFS)
–21
–18
–15
–12
INPUT AMPLITUDE (dBFS)
100
AIN = –6.5dBFS
SNR = 72.7dBFS
SFDR = 78.8dBFS
–20
–24
Figure 17. Two-Tone SNR/SFDR vs. Input Amplitude @ 30 MHz and 31 MHz
Figure 14. Two-Tone 8K FFT @ 30 MHz and 31 MHz
–10
–27
03583-B-029
–40
–50
–60
–70
–80
–90
80
70
SNR (dBFS)
SFDR = 90dBc
REFERENCE LINE
60
SNR (dBc)
50
–100
–110
–120
0
5
10
15
20
25
FREQUENCY (MHz)
30
35
40
–30
40
–27
03583-B-030
Figure 15. Two-Tone 8K FFT @ 69 MHz and 70 MHz
–24
–21
–18
–15
INPUT AMPLITUDE (dBFS)
–12
–9
–6
03583-B-027
Figure 18. Two-Tone SNR/SFDR vs. Input Amplitude @ 69 MHz and 70 MHz
1.0
1.5
0.8
1.0
0.6
0.4
DNL (LSB)
INL (LSB)
0.5
0
–0.5
0.2
0
–0.2
–0.4
–0.6
–1.0
–0.8
–1.5
–1.0
0
2048
4096
6144
8192
CODE
10240 12288 14336 16384
0
2048
4096
6144
8192 10240 12288 14336 16384
CODE
03583-B-028
03583-B-026
Figure 19. Typical DNL
Figure 16. Typical INL
Rev. B | Page 12 of 28
AD9245
75
100
74
–40°C
73
95
+25°C
72
SNR (dBc)
SFDR (dBc)
90
71
+85°C
70
69
68
85
–40°C
80
+25°C
+85°C
67
75
66
65
0
25
50
75
INPUT FREQUENCY (MHz)
100
70
125
0
Figure 20. SNR vs. Input Frequency
90
–20
–30
AMPLITUDE (dBFS)
SNR/SFDR (dBc)
125
03583-B-038
–10
84
SFDR (DCS OFF)
82
80
78
76
–40
–50
–60
–70
–80
–90
74
SNR (DCS OFF)
–100
72
–110
SNR (DCS ON)
35
40
45
50
55
DUTY CYCLE (%)
60
65
–120
70
0
–20
–20
–30
–30
–40
–40
AMPLITUDE (dBFS)
0
–10
–50
–60
–70
–80
03583-B-060
–60
–70
–80
–90
–100
–100
–110
–110
–120
–120
28.8
38.4
–50
–90
19.2
28.8
Figure 24. Two 32K FFT CDMA-2000 Carriers @
FIN = 46.08 MHz; Sample Rate = 61.44 MSPS
0
FREQUENCY (MHz)
19.2
FREQUENCY (MHz)
–10
9.6
9.6
03583-B-037
Figure 21. SNR/SFDR vs. Clock Duty Cycle
AMPLITUDE (dBFS)
100
0
SFDR (DCS ON)
86
0
50
75
INPUT FREQUENCY (MHz)
Figure 23. SFDR vs. Input Frequency
88
70
30
25
03583-B-036
38.4
0
9.6
19.2
28.8
FREQUENCY (MHz)
03583-B-059
Figure 22. 32K FFT WCDMA Carrier @ FIN = 96 MHz; Sample Rate = 76.8 MSPS
Rev. B | Page 13 of 28
Figure 25. Two 32K FFT WCDMA Carriers @
FIN = 76.8 MHz; Sample Rate = 61.44 MSPS
38.4
03583-B-061
AD9245
THEORY OF OPERATION
The AD9245 architecture consists of a front-end sample and
hold amplifier (SHA) followed by a pipelined switched capacitor ADC. The pipelined ADC is divided into three sections,
consisting of a 4-bit first stage followed by eight 1.5-bit stages
and a final 3-bit flash. Each stage provides sufficient overlap to
correct for flash errors in the preceding stages. The quantized
outputs from each stage are combined into a final 14-bit result
in the digital correction logic. The pipelined architecture permits the first stage to operate on a new input sample, while the
remaining stages operate on preceding samples. Sampling
occurs on the rising edge of the clock.
Each stage of the pipeline, excluding the last, consists of a low
resolution flash ADC connected to a switched capacitor DAC
and interstage residue amplifier (MDAC). The residue amplifier
magnifies the difference between the reconstructed DAC output
and the flash input for the next stage in the pipeline. One bit of
redundancy is used in each stage to facilitate digital correction
of flash errors. The last stage simply consists of a flash ADC.
The input stage contains a differential SHA that can be accoupled or dc-coupled in differential or single-ended modes.
The output-staging block aligns the data, carries out the error
correction, and passes the data to the output buffers. The output
buffers are powered from a separate supply, allowing adjustment
of the output voltage swing. During power-down, the output
buffers go into a high impedance state.
Referring to Figure 27, the clock signal alternately switches the
SHA between sample mode and hold mode. When the SHA is
switched into sample mode, the signal source must be capable
of charging the sample capacitors and settling within one-half
of a clock cycle. A small resistor in series with each input can
help reduce the peak transient current required from the output
stage of the driving source. Also, a small shunt capacitor can be
placed across the inputs to provide dynamic charging currents.
This passive network creates a low-pass filter at the ADC’s
input; therefore, the precise values are dependent upon the
application. In IF undersampling applications, any shunt
capacitors should be reduced or removed. In combination with
the driving source impedance, they would limit the input
bandwidth.
H
T
T
5pF
VIN–
CPAR
T
H
03583-B-012
Figure 27. Switched-Capacitor SHA Input
The analog input to the AD9245 is a differential switchedcapacitor SHA that has been designed for optimum performance while processing a differential input signal. The SHA input
can support a wide common-mode range (VCM) and maintain
excellent performance, as shown in Figure 26. An input
common-mode voltage of midsupply minimizes signaldependent errors and provides optimum performance.
100
95
For best dynamic performance, the source impedances driving
VIN+ and VIN– should be matched such that common-mode
settling errors are symmetrical. These errors are reduced by the
common-mode rejection of the ADC.
An internal differential reference buffer creates positive and
negative reference voltages, REFT and REFB, that define the
span of the ADC core. The output common mode of the
reference buffer is set to midsupply, and the REFT and REFB
voltages and span are defined as follows:
SFDR (2.5MHz)
90
1
REFT = ( AVDD + VREF )
2
1
REFB = ( AVDD − VREF )
2
Span = 2 × (REFT − REFB ) = 2 × VREF
85
SNR/SFDR (dBc)
T
CPAR
ANALOG INPUT AND REFERENCE OVERVIEW
SFDR (39MHz)
80
75
SNR (2.5MHz)
70
SNR (39MHz)
65
It can be seen from the equations above that the REFT and
REFB voltages are symmetrical about the midsupply voltage, and,
by definition, the input span is twice the value of the VREF voltage.
60
55
50
0.5
5pF
VIN+
1.0
1.5
2.0
COMMON-MODE LEVEL (V)
2.5
Figure 26. SNR, SFDR vs. Common-Mode Level
3.0
03583-B-039
The internal voltage reference can be pin strapped to fixed
values of 0.5 V or 1.0 V, or adjusted within the same range as
discussed in the Internal Reference Connection section.
Maximum SNR performance is achieved with the AD9245 set
Rev. B | Page 14 of 28
AD9245
to the largest input span of 2 V p-p. The relative SNR degradation
is 3 dB when changing from 2 V p-p mode to 1 V p-p mode.
33Ω
The SHA may be driven from a source that keeps the signal
peaks within the allowable range for the selected reference voltage. The minimum and maximum common-mode input levels
are defined as
VCM MIN =
VCM MAX =
2V p-p
49.9Ω
10pF
33Ω
AVDD
VIN+
AD9245
VIN–
AGND
1kΩ
VREF
2
0.1µF
1kΩ
03583-B-014
( AVDD + VREF )
Figure 29. Differential Transformer-Coupled Configuration
2
The minimum common-mode input level allows the AD9245 to
accommodate ground referenced inputs.
Although optimum performance is achieved with a differential
input, a single-ended source may be applied to VIN+ or VIN–.
In this configuration, one input accepts the signal, while the
opposite input should be set to midscale by connecting it to an
appropriate reference. For example, a 2 V p-p signal may be
applied to VIN+ while a 1 V reference is applied to VIN–. The
AD9245 then accepts an input signal varying between 2 V and
0 V. In the single-ended configuration, distortion performance
may degrade significantly as compared to the differential case.
However, the effect is less noticeable at lower input frequencies.
The signal characteristics must be considered when selecting
a transformer. Most RF transformers saturate at frequencies
below a few MHz, and excessive signal power can also cause
core saturation, which leads to distortion.
Single-Ended Input Configuration
A single-ended input may provide adequate performance in
cost-sensitive applications. In this configuration, there is a
degradation in SFDR and distortion performance due to the
large input common-mode swing (see Figure 13). However, if
the source impedances on each input are matched, there should
be little effect on SNR performance. Figure 30 details a typical
single-ended input configuration.
Differential Input Configurations
1kΩ
As previously detailed, optimum performance is achieved while
driving the AD9245 in a differential input configuration. For
baseband applications, the AD8138 differential driver provides
excellent performance and a flexible interface to the ADC. The
output common-mode voltage of the AD8138 is easily set to
AVDD/2, and the driver can be configured in a Sallen Key filter
topology to provide band limiting of the input signal.
2V p-p
0.33µF 1kΩ
49.9Ω
20pF
1kΩ
+
10µF
33Ω
0.1µF
33Ω
AVDD
VIN+
AD9245
VIN–
1kΩ
AGND
03583-B-015
Figure 30. Single-Ended Input Configuration
CLOCK INPUT CONSIDERATIONS
1V p-p
49.9Ω
499Ω
33Ω
499Ω
AD8138
1kΩ
33Ω
523Ω
0.1µF
20pF
1kΩ
499Ω
AVDD
VIN+
AD9245
VIN–
AGND
03583-B-013
Figure 28. Differential Input Configuration Using the AD8138
At input frequencies in the second Nyquist zone and above, the
performance of most amplifiers is not adequate to achieve the
true performance of the AD9245. This is especially true in IF
undersampling applications where frequencies in the 70 MHz to
100 MHz range are being sampled. For these applications,
differential transformer coupling is the recommended input
configuration. The value of the shunt capacitor is dependent on
the input frequency and source impedance and should be
reduced or removed. An example is shown in Figure 29.
Typical high speed ADCs use both clock edges to generate a
variety of internal timing signals, and as a result may be sensitive
to clock duty cycle. Commonly a 5% tolerance is required on the
clock duty cycle to maintain dynamic performance characteristics. The AD9245 contains a clock duty cycle stabilizer (DCS) that
retimes the nonsampling edge, providing an internal clock signal
with a nominal 50% duty cycle. This allows a wide range of clock
input duty cycles without affecting the performance of the
AD9245. As shown in Figure 21, noise and distortion performance is nearly flat for a 30% to 70% duty cycle with the DCS on.
The duty cycle stabilizer uses a delay-locked loop (DLL) to
create the nonsampling edge. As a result, any changes to the
sampling frequency require approximately 100 clock cycles to
allow the DLL to acquire and lock to the new rate.
Rev. B | Page 15 of 28
AD9245
140
425
JITTER CONSIDERATIONS
]
In the equation, the rms aperture jitter represents the root-mean
square of all jitter sources, which include the clock input, analog
input signal, and ADC aperture jitter specification. IF undersampling applications are particularly sensitive to jitter (see Figure 31).
375
80
60
350
40
325
20
DIGITAL CURRENT
300
10
The clock input should be treated as an analog signal in cases
where aperture jitter may affect the dynamic range of the
AD9245. Power supplies for clock drivers should be separated
from the ADC output driver supplies to avoid modulating the
clock signal with digital noise. Low jitter, crystal controlled
oscillators make the best clock sources. If the clock is generated
from another type of source (by gating, dividing, or other methods), it should be retimed by the original clock at the last step.
0.2ps
MEASURED SNR
65
SNR (dBc)
0.5ps
60
30
40
50
60
70
SAMPLE RATE (MSPS)
80
90
0
100
03583-B-035
Figure 32. Power and Current vs. Sample Rate @ 2.5 MHz
Reducing the capacitive load presented to the output drivers can
minimize digital power consumption. The data in Figure 32 was
taken with the same operating conditions as the Typical Performance Characteristics, and with a 5 pF load on each output
driver.
1.0ps
Low power dissipation in standby mode is achieved by shutting
down the reference, reference buffer, and biasing networks. The
decoupling capacitors on REFT and REFB are discharged when
entering standby mode and then must be recharged when
returning to normal operation. As a result, the wake-up time is
related to the time spent in standby mode, and shorter standby
cycles result in proportionally shorter wake-up times. With the
recommended 0.1 µF and 10 µF decoupling capacitors on REFT
and REFB, it takes approximately 1 second to fully discharge the
reference buffer decoupling capacitors and 7 ms to restore full
operation.
1.5ps
55
2.0ps
2.5ps
50
3.0ps
45
40
20
By asserting the PDWN pin high, the AD9245 is placed in
standby mode. In this state, the ADC typically dissipates
1 mW if the CLK and analog inputs are static. During standby,
the output drivers are placed in a high impedance state.
Reasserting the PDWN pin low returns the AD9245 to its
normal operational mode.
75
70
100
TOTAL POWER
CURRENT (mA)
[
SNR = 20 log 2 πf INPUT × t J
120
400
TOTAL POWER (mW)
High speed, high resolution ADCs are sensitive to the quality of
the clock input. The degradation in SNR at a given input frequency (fINPUT) due only to aperture jitter (tJ) can be calculated
with the following equation:
ANALOG CURRENT
1
10
100
INPUT FREQUENCY (MHz)
1000
03583-B-041
Figure 31. SNR vs. Input Frequency and Jitter
POWER DISSIPATION AND STANDBY MODE
As shown in Figure 32, the power dissipated by the AD9245 is
proportional to its sample rate. The digital power dissipation is
determined primarily by the strength of the digital drivers and
the load on each output bit. The maximum DRVDD current
(IDRVDD) can be calculated as
I DRVDD = VDRVDD × C LOAD × f CLK × N
where N is the number of output bits, 14 in the case of the
AD9245. This maximum current occurs when every output bit
switches on every clock cycle, i.e., a full-scale square wave at the
Nyquist frequency, fCLK/2. In practice, the DRVDD current will
be established by the average number of output bits switching,
which will be determined by the sample rate and the characteristics of the analog input signal.
DIGITAL OUTPUTS
The AD9245 output drivers can be configured to interface with
2.5 V or 3.3 V logic families by matching DRVDD to the digital
supply of the interfaced logic. The output drivers are sized to provide sufficient output current to drive a wide variety of logic
families. However, large drive currents tend to cause current
glitches on the supplies that may affect converter performance.
Applications requiring the ADC to drive large capacitive loads or
large fanouts may require external buffers or latches.
As detailed in Table 9, the data format can be selected for either
offset binary or twos complement.
Rev. B | Page 16 of 28
AD9245
TIMING
The AD9245 provides latched data outputs with a pipeline delay
of seven clock cycles. Data outputs are available one propagation delay (tPD) after the rising edge of the clock signal. Refer to
Figure 2 for a detailed timing diagram.
In all reference configurations, REFT and REFB drive the A/D
conversion core and establish its input span. The input range of
the ADC always equals twice the voltage at the reference pin for
either an internal or an external reference.
VIN+
The length of the output data lines and the loads placed on
them should be minimized to reduce transients within the
AD9245. These transients can degrade the converter’s dynamic
performance.
VIN–
+
10µF
0.1µF
SELECT
LOGIC
SENSE
0.5V
AD9245
03583-B-017
Figure 33. Internal Reference Configuration
If the internal reference of the AD9245 is used to drive multiple
converters to improve gain matching, the loading of the reference by the other converters must be considered. Figure 34
depicts how the internal reference voltage is affected by loading.
0.05
0
–0.05
ERROR (%)
A comparator within the AD9245 detects the potential at the
SENSE pin and configures the reference into one of four
possible states, which are summarized in Table 8. If SENSE is
grounded, the reference amplifier switch is connected to the
internal resistor divider (see Figure 33), setting VREF to 1 V.
Connecting the SENSE pin to VREF switches the reference
amplifier output to the SENSE pin, completing the loop and
providing a 0.5 V reference output. If a resistor divider is
connected as shown in Figure 35, the switch is again set to the
SENSE pin. This puts the reference amplifier in a noninverting
mode with the VREF output defined as follows:
+
0.1µF
VREF
A stable and accurate 0.5 V voltage reference is built into the
AD9245. The input range can be adjusted by varying the reference voltage applied to the AD9245 using either the internal
reference or an externally applied reference voltage. The input
span of the ADC tracks reference voltage changes linearly. The
various reference modes are summarized Table 8 and described
in the following sections.
INTERNAL REFERENCE CONNECTION
0.1µF
REFB
10µF
If the ADC is being driven differentially through a transformer,
the reference voltage can be used to bias the center tap (common-mode voltage).
0.1µF
ADC
CORE
The lowest typical conversion rate of the AD9245 is 1 MSPS. At
clock rates below 1 MSPS, dynamic performance may degrade.
VOLTAGE REFERENCE
REFT
0.5V ERROR (%)
–0.10
1.0V ERROR (%)
–0.15
–0.20
R2 
VREF = 0.5 × 1 +

 R1 
–0.25
0
0.5
1.0
1.5
LOAD (mA)
2.0
2.5
3.0
03583-B-019
Figure 34. VREF Accuracy vs. Load
Table 8. Reference Configuration Summary
Selected Mode
External Reference
Internal Fixed Reference
Programmable Reference
SENSE Voltage
AVDD
VREF
0.2 V to VREF
Internal Switch
Position
N/A
SENSE
SENSE
Resulting VREF (V)
N/A
0.5
Internal Fixed Reference
AGND to 0.2 V
Internal Divider
1.0
R2  (See Figure 35)
0 . 5 ×  1 +

R1 

Rev. B | Page 17 of 28
Resulting Differential
Span (V p-p)
2 × External Reference
1.0
2 × VREF
2.0
AD9245
OPERATIONAL MODE SELECTION
VIN+
VIN–
REFT
0.1µF
ADC
CORE
+
0.1µF
10µF
REFB
0.1µF
VREF
+
10µF
0.1µF
Table 9. Mode Selection
SELECT
LOGIC
R2
As discussed earlier, the AD9245 can output data in either offset
binary or twos complement format. There is also a provision for
enabling or disabling the clock duty cycle stabilizer (DCS). The
MODE pin is a multilevel input that controls the data format
and DCS state. The input threshold values and corresponding
mode selections are outlined in Table 9.
SENSE
R1
MODE Voltage
AVDD
2/3 AVDD
1/3 AVDD
AGND (Default)
0.5V
AD9245
Data Format
Twos Complement
Twos Complement
Offset Binary
Offset Binary
Duty Cycle
Stabilizer
Disabled
Enabled
Enabled
Disabled
03583-B-018
Figure 35. Programmable Reference Configuration
EVALUATION BOARD
EXTERNAL REFERENCE OPERATION
The use of an external reference may be necessary to enhance
the gain accuracy of the ADC or improve thermal drift characteristics. When multiple ADCs track one another, a single
reference (internal or external) may be necessary to reduce gain
matching errors to an acceptable level. Figure 36 shows the typical drift characteristics of the internal reference in both 1.0 V
and 0.5 V modes.
When the SENSE pin is tied to AVDD, the internal reference is
disabled, allowing the use of an external reference. An internal
reference buffer loads the external reference with an equivalent
7 kΩ load. The internal buffer still generates the positive and
negative full-scale references, REFT and REFB, for the ADC
core. The input span is always twice the value of the reference
voltage; therefore, the external reference must be limited to a
maximum of 1.0 V.
It is critical that signal sources with very low phase noise (<1 ps
rms jitter) be used to realize the ultimate performance of the
converter. Proper filtering of the input signal, to remove
harmonics and lower the integrated noise at the input, is also
necessary to achieve the specified noise performance.
The AD9245 can be driven single-ended or differentially
through a transformer. Separate power pins are provided to
isolate the DUT from the support circuitry. Each input configuration can be selected by proper connection of various jumpers
(refer to the schematics).
An alternative differential analog input path using an AD8351
op amp is included in the layout, but is not populated in production. Designers interested in evaluating the op amp with the
ADC should remove C15, R12, and R3, and populate the op
amp circuit. The passive network between the AD8351 outputs
and the AD9245 allows the user to optimize the frequency
response of the op amp for the application.
1.0
0.9
0.8
0.7
VREF ERROR (%)
The AD9245 evaluation board provides all of the support
circuitry required to operate the ADC in its various modes and
configurations. Complete schematics and layout plots follow
and demonstrate the proper routing and grounding techniques
that should be applied at the system level.
0.6
0.5
0.4
0.3
VREF = 1.0V
0.2
0.1
VREF = 0.5V
0
–40 –30 –20 –10
0
10 20 30 40 50
TEMPERATURE (°C)
60
70
80
03583-B-040
Figure 36. Typical VREF Drift
Rev. B | Page 18 of 28
GND
J1
Figure 37. LFCSP Evaluation Board Schematic—Analog Inputs and DUT
PRI SEC
PRI SEC
R10
36Ω
E 45
XOUT
R11
36Ω
R18
25Ω
0.1µF
C11
R2
XX
C18
0.10µF
GND
GND
C21
10pF
R15
33Ω
AVDD
R13
1kΩ
C23
10pF
P4
P3
R25
1kΩ
GND
AVDD
3
2
4
P1
AVDD
GND
VIN+
VIN–
R6
1kΩ
R7
1kΩ
R5
1kΩ
GND
GND
15pF C19
OR L1
FOR FILTER
GND
R4
33Ω
R36
1kΩ
R26
1kΩ
C22
10µF
GND GND
C13
0.10µF
AVDD
GND
D
R SINGLE ENDED
R3
0Ω
GND
C5
0.1µF
C26
10pF
AMPINB
R3, R17, R18
ONLY ONE SHOULD BE
ON BOARD AT A TIME
XOUTB
GND
C7
0.1µF
GND
R12
0Ω
GND
P10
C
C29
10µF
E
P7 A B
C9
0.10µF
AMPIN
C16
0.1µF
R42
0Ω
6
2 CT
4
T1
ADT1–1WT
XFRIN1 1
5
NC
3
GND
C6
0.1µF
GND
0.1µF
C12
OPTIONAL XFR
T2
FT C1–1–13
5
1
XOUT
X FRIN
2
CT
3
4
GND
XOUTB
C15
AMP 0.1µF
L1
10nH
GND
R9
10kΩ
R1
10kΩ
AVDD
GND
P11
P9 P8
P6
MODE
2
P5
31 AGND
32 AVDD
28 AGND
29 VIN+
30 VIN–
25 REFB
26 REFT
27 AVDD
GND
GND
AVDD
P14
CLK
C8
0.1µF
U4
AD9245
P13
R8
1kΩ
15
14
16
GND
D5 10
D4 9
13
D7 12
D6 11
DVDD
DGND
D9
D8
1
AVDD
6
5
4
GND
3
2.5V DRVDD
2
GND
(LSB)
12
11
10
9
6
7
8
RP1 220Ω
13
5
10
9
7
8
4
11
6
15
14
12
5
16
13
4
2
3
15
14
1
16
MODE PIN SOLDERABLE JUMPER:
5 TO 1: TWOS COMPLEMENT/DCS OFF
5 TO 2: TWOS COMPLEMENT/DCS ON
5 TO 3: OFFSET BINARY/DCS ON
5 TO 4: OFFSET BINARY/DCS OFF
03583-B-050
D0X
D1X
D2X
D4X
D3X
D6X
D5X
D8X
D7X
D9X
D10X
D12X
D11X
DRX
D13X
H4
MTHOLE6
H3
MTHOLE6
H2
MTHOLE6
H1
MTHOLE6
2
3
RP2 220Ω
GND
1
P2
SENSE PIN SOLDERABLE JUMPER:
E TO A: EXTERNAL VOLTAGE DIVIDER
E TO B: INTERNAL 1V REFERENCE (DEFAULT)
E TO C: EXTERNAL REFERENCE
E TO D: INTERNAL 0.5V REFERENCE
DRVDD
GND
(MSB)
OVERRANGE BIT
3.0V
1
1 DNC
2 CLK
D12 19
D11 18
D10 17
VDL
2.5V
AVDD
VREF 24
SENSE 23
MODE 22
OTR 21
D13 20
3 DNC
4 PDWN
Rev. B | Page 19 of 28
5 D0
6 D1
7 D2
8 D3
VAMP
5.0V
EXTREF
1V MAX E1
AD9245
LSB
MSB
Figure 38. LFCSP Evaluation Board Schematic—Digital Path
Rev. B | Page 20 of 28
GND
R19
50Ω
AMP
2CLK
IN
1
U1
2OE
24
OUT
23
2Q7
22
GND
21
2Q6
20
2Q5
19
VCC
18
2Q4
17
2Q3
16
GND
15
2Q2
14
2Q1
13
1Q8
12
1Q7
11
GND
10
1Q6
9
1Q5
VCC 8
7
1Q4
6
1Q3
5
GND
4
1Q2
3
1Q1
2
1OE
1
2QB
GND
R35
25Ω
R40
10kΩ
GND
C35
0.10µF
C28
0.1µF
R41
10kΩ
VAMP
POWER DOWN
USE R40 OR R41
2DB
26
2D7
27
GND
28
2D6
29
2D5
30
V
31 CC
2D4
32
2D3
33
GND
34
2D2
35
2D1
36
1D8
37
1D7
38
GND
39
1D6
40
1D5
41
VCC
42
1D4
43
1D3
44
GND
45
1D2
46
1D1
47
1CLK
48
25
AMP IN
CLKLAT/DAC
GND
D0X
DRVDD
D2X
D1X
D4X
D3X
GND
D5X
D7X
D6X
GND
D8X
D10X
D9X
DRVDD
D11X
GND
D12X
DRX
D13X
CLKAT/DAC
74LVTH162374
R33 RPG2 5
25Ω
INLO 4
INHI 3
PWDN 1
RGP1 2
GND
GND
DRVDD
GND
GND
DRVDD
GND
GND
R34
1.2kΩ
U3
AD8351
GND
VAMP
DRY
6 COMM
7 OPLO
9 VPOS
8 OPHI
10 VOCM
C44
0.1µF
R38
1kΩ
GND
R14
25Ω
VAMP
R39
1kΩ
C45
0.1µF
C24
10µF
R17
0Ω
R16
0Ω
GND
GND
GND
MSB
GND
C17
0.1µF
C27
0.1µF
GND
DRY
GND
DR
AMPIN
AMPINB
11
9
7
3
5
1
13
13
15
15
17
17
19
19
21
21
23
23
25
25
27
27
29
29
31
31
33
33
35
35
37
37
39
39
11
12
7
9
10
8
1
4
6
HEADER 40
3
5
2
14
14
16
16
18
18
20
20
22
22
24
24
26
26
28
28
30
30
32
32
34
34
36
36
38
38
40
40
12
10
8
4
6
2
03583-B-051
GND
AD9245
C4
10µF
GND
Figure 39. LFCSP Evaluation Board Schematic—Clock Input
Rev. B | Page 21 of 28
J2
GND
R29
50Ω
C43
0.1µF
ENC
ENCX
GND
R30
1kΩ
R31
1kΩ
VDL
R27
0Ω
R28
0Ω
VDL
VDL
E43
E44
E35
E51
E52
VDL
E31
VDL
E50
CLK
ENC
C33
C14
0.1µF 0.001µF
ANALOG BYPASSING
C32
0.001µF
CLOCK TIMING ADJUSTMENTS
GND
ENCODE
C25
10µF
GND
AVDD
FOR A BUFFERED ENCODE USE R28
FOR A DIRECT ENCODE USE R27
AVDD
C3
10µF
DUT BYPASSING
C10
22µF
VDL DRVDD
R20
1kΩ
GND
GND
R24
1kΩ
GND
R21
1kΩ
GND
E53
GND
R32
1kΩ
C41
0.1µF
DRVDD
C30
0.001µF
5
9
10
12
13
3A
3B
4A
4B
2B
1 1A
2 1B
4 2A
C31
0.1µF
GND
U5
14
8
11
6
7
3
C34
0.1µF
4Y PWR
3Y
2Y
1Y
74VCX86
DIGITAL BYPASSING
C2
22µF
VDL
GND
ENCX
C36
0.1µF
R23
0Ω
C1
C39
0.001µF 0.1µF
CLKLAT/DAC
R37
25Ω
Rx
DNP
VDL
DR
R22
0Ω
GND
C49
0.001µF
LATCH BYPASSING
C47
0.1µF
SCHEMATIC SHOWS TWO GATE DELAY SETUP.
FOR ONE DELAY, REMOVE R22 AND R37 AND
ATTACH Rx (Rx = 0Ω).
C38
0.001µF
C48
0.001µF
GND
VAMP
C20
10µF
C40
0.001µF
03583-B-052
C46
10µF
C37
0.1µF
AD9245
AD9245
03583-B-055
03583-B-053
Figure 42. LFCSP Evaluation Board Layout, Ground Plane
Figure 40. LFCSP Evaluation Board Layout, Primary Side
03583-B-056
03583-B-054
Figure 43. LFCSP Evaluation Board Layout, Power Plane
Figure 41. LFCSP Evaluation Board Layout, Secondary Side
Rev. B | Page 22 of 28
AD9245
03583-B-057
03583-B-058
Figure 44. LFCSP Evaluation Board Layout, Primary Silkscreen
Figure 45. LFCSP Evaluation Board Layout, Secondary Silkscreen
Rev. B | Page 23 of 28
AD9245
Table 10. LFCSP Evaluation Board Bill of Materials
Item Qty. Omit1 Reference Designator
C1, C5, C7, C8, C9, C11, C12,
18
C13, C15, C16, C31, C33, C34,
C36, C37, C41, C43, C47
1
C6, C18, C27, C17,
8
C28, C35, C45, C44
C2, C3, C4, C10, C20, C22,
8
C25, C29
2
2
C46, C24
C14, C30, C32, C38,
3
8
C39, C40, C48, C49
4
3
C19, C21, C23
Device
Package
Value
Chip Capacitor
0603
0.1 µF
Tantalum Capacitor
TAJD
10 µF
Chip Capacitor
0603
0.001 µF
5
6
Chip Capacitor
0603
10 pF
1
C26
Chip Capacitor
0603
10 pF
9
E31, E35, E43, E44, E50, E51,
E52, E53
Header
EHOLE
J1, J2
SMA Connector/50 Ω
SMA
L1
Inductor
0603
2
7
2
8
1
Recommended
Vendor/Part Number
Jumper Blocks
E1, E45
10 nH
Coilcraft/0603CS10NXGBU
9
1
P2
Terminal Block
TB6
Wieland/25.602.2653.0,
z5-530-0625-0
10
1
P12
Header Dual 20-Pin RT Angle
HEADER40
Digi-Key S2131-20-ND
Chip Resistor
0603
0Ω
R4, R15
Chip Resistor
R5, R6, R7, R8, R13, R20, R21,
R24, R25, R26, R30, R31, R32, Chip Resistor
R36
R10, R11
Chip Resistor
0603
33 Ω
0603
1 kΩ
0603
36 Ω
Chip Resistor
0603
50 Ω
220 Ω
11
5
12
2
13
14
14
2
15
R3, R12, R23, R28, Rx
6
1
R16, R17, R22, R27, R42, R37
R29
1
Supplied
by ADI
R19
16
2
RP1, RP2
Resistor Pack
R_742
17
1
T1
ADT1-1WT
AWT1-1T
Digi-Key
CTS/742C163220JTR
Mini-Circuits
18
1
U1
74LVTH162374 CMOS Register TSSOP-48
19
1
U4
AD9245BCP ADC (DUT)
CSP-32
Analog Devices, Inc.
X
20
1
U5
74VCX86M
SOIC-14
Fairchild
21
1
PCB
AD92XXBCP/PCB
PCB
Analog Devices, Inc.
X
X
22
1
U3
AD8351 Op Amp
MSOP-8
Analog Devices, Inc.
23
1
T2
MACOM Transformer
ETC1-1-13 1-1 TX
MACOM/ETC1-1-13
24
5
R1, R2, R9, R38, R39
Chip Resistor
0603
SELECT
25
3
R14, R18, R35
Chip Resistor
0603
25 Ω
0603
10 kΩ
26
2
R40, R41
Chip Resistor
27
1
R34
Chip Resistor
1.2 kΩ
28
1
R33
Chip Resistor
100 Ω
Total 82
34
1
These items are included in the PCB design, but are omitted at assembly.
Rev. B | Page 24 of 28
AD9245
OUTLINE DIMENSIONS
5.00
BSC SQ
0.60 MAX
25
24
PIN 1
INDICATOR
0.50
BSC
4.75
BSC SQ
TOP
VIEW
0.50
0.40
0.30
12° MAX
1.00
0.85
0.80
PIN 1
INDICATOR
0.60 MAX
32
1
3.25
3.10 SQ
2.95
BOTTOM
VIEW
17
16
9
8
0.25 MIN
3.50 REF
0.80 MAX
0.65 TYP
0.05 MAX
0.02 NOM
SEATING
PLANE
0.30
0.23
0.18
0.20 REF
COPLANARITY
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VHHD-2
Figure 46. 32-Lead Frame Chip Scale Package [LFCSP]
(CP-32-1)
Dimensions shown in millimeters
ORDERING GUIDE
AD9245 Products
AD9245BCP-801
AD9245BCPRL7–801
AD9245BCPZ-801, 2
AD9245BCPZRL7-801, 2
AD9245BCP-80EB1
Temperature Range
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Package Description
Lead Frame Chip Scale Package (LFCSP)
Lead Frame Chip Scale Package (LFCSP)
Lead Frame Chip Scale Package (LFCSP)
Lead Frame Chip Scale Package (LFCSP)
Evaluation Board
Package Outline
CP-32-1
CP-32-1
CP-32-1
CP-32-1
1
It is recommended that the exposed paddle be soldered to the ground plane for the LFCSP package. There is an increased reliability of the solder joints, and the maximum thermal capability of the package is achieved with the exposed paddle soldered to the customer board.
2
Z = Lead Free.
Rev. B | Page 25 of 28
AD9245
NOTES
Rev. B | Page 26 of 28
AD9245
NOTES
Rev. B | Page 27 of 28
AD9245
NOTES
© 2003 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners.
C03583-0-10/03(B)
Rev. B | Page 28 of 28