High Voltage Auxiliary Power Supply with the
Simplified Power Circuit Topology for the DC Trains
Dmitri Vinnikov
Juhan Laugis
Department of Electrical Drives and Power Electronics
Tallinn University of Technology
Ehitajate tee 5, 19086, Tallinn, Estonia
dm.vin@mail.ee
Department of Electrical Drives and Power Electronics
Tallinn University of Technology
Ehitajate tee 5, 19086, Tallinn, Estonia
laugis@cc.ttu.ee
Abstract— The paper proposes a simplified power circuit
topology for the auxiliary power supplies (APS) with output
powers up to 50 kW to be used in 3.0 kV DC commuter trains.
Focus is on the implementation of 6.5 kV IGBTs to improve
power density, system integrity as well as the reliability of the
APS. For instance, a topology proposal, an analysis and a
comparison with other types is presented. Some problems of
converter control and protection as well as hardware design
considerations are discussed. Efficiency estimation of the new
topology and some recommendations of loss minimization are
given.
Keywords- 6.5 kV IGBT, half-bridge isolated DC/DC converter,
rolling stock, efficiency, high-voltage switch
I.
requirement has been successfully implemented in the series
connection of conventional IGBTs (Fig. 1) with the
considerable lower blocking voltages as compared to the
requirements. Actually, connecting IGBTs in series allows fast
high-power/high-voltage semiconductor switches (HV switch)
with operating voltages of several kilovolts to be realized.
Depending on the voltage blocking capability of the single
transistors to be connected in series (1.2 kV, 1.7 kV, 3.3 kV or
others), the total number of interconnected IGBTs varies from
6 to 2 (see TABLE I). Such combined high-voltage switches
can operate with the nominal DC-link (UDC) voltage level of
3600 V (±20%).
INTRODUCTION
Development of power converters for the electric rolling
stock is a typical application of modern power electronics.
Among the different railway electrification systems, the most
challenging one for a power electronics designer is 3000 V
DC. Such railway tractions are currently used in Belgium,
Italy, Poland, Spain, the northern Czech Republic, Slovakia,
Slovenia, western Croatia, South Africa and in the former
Soviet Union countries (Russia, Ukraine, Latvia, and Estonia).
To guarantee safe operation for all operation points (input
voltage range 2200...4000 V DC), selection criteria for the
voltage blocking capability of the switching devices used in
rolling stock converters must be twice the nominal catenary
voltage. Regarding this design rule, converters based on the
IGBT technology for the catenary voltages of 3.0 kV DC are
only possible with IGBTs with the Uds values not lower than
6.0 kV. Up to today, this blocking voltage capability
Fig. 1. Example of APS with n series-connected IGBTs.
TABLE I.
EVALUATION OF DIFFERENT CANDIDATE IGBTS FOR THE HV SWITCH
Nominal
Max. blocking
Min. number of
value of UDC,
voltage UCEmax,
IGBT type
series connected
for HV switch
kV
kV (± 20%)
1.2 kV IGBT
600
1.2
6
1.7 kV IGBT
900
1.7
4
3.3 kV IGBT
1800
3.3
2
Series connection of elements for proper voltage blocking
capability leads to auxiliary voltage-balancing circuits (voltage
clamping) to be implemented on the high-voltage primary
inverter or in the inputs of cascaded converters, increasing the
complexity and decreasing the overall reliability of the
designed system.
II.
Another possibility to cope with the high-voltage blocking
capability problem is the cascaded converter topology (Fig. 2).
In that case, a number of identical two-level isolated DC/DC
converters are connected in series at their inputs to achieve the
desired voltage blocking capability of primary inverters. The
number of cascaded converters varies from 6 to 2, which once
again depends on the voltage blocking capability of IGBTs
used in primary inverters. Output leads of these converters are
connected in parallel, thus the output current is distributed
between the converters. It gives an extra opportunity to use
IGBT switches and secondary rectifier diodes with the reduced
operation current, thus implementing higher switching
frequencies. Another advantage of cascaded converter
topology is the effect of “distributed magnetic cores”, i.e.
using a number of isolation transformers instead of one bulky
unit enables a more compact and space-saving design of a
power converter. The same assumption applies to the output
filter chokes as well.
POWER SCHEME SIMPLIFICATION POSSIBILITIES
The 6.5 kV IGBT modules (EUPEC, ABB, IXYS, DYNEX,
etc.) recently implemented are basically designed for 3.0 kV
DC rolling stock applications with their high demands on
reliability concerning thermal cycling capability. A single
IGBT has the voltage blocking capability two times the
nominal catenary voltage level, which copes with the
requirements for the rolling stock power electronics. Such
transistors give an attractive possibility of using a simple 2level inverter topology, achieving better efficiency, power
density and reliability compared to older designs.
Fig. 3. 6.5 kV IGBT family (200 A, 400 A and 600 A; by EUPEC) [2].
Today’s state-of-the-art 6.5 kV IGBT modules are available
in three basic configurations: with 200 A, 400 A and 600 A
collector current (Fig. 3). While for the traction drives with
their MW-power ranges it is essential to use the parallel
connection of HV-IGBTs, for such middle-power applications
like onboard APS, the use of single 6.5 kV IGBT modules
becomes economically feasible.
III.
Fig. 2. Example of APS with n series-connected converters.
In general, both presented topologies combine some serious
disadvantages mainly related to the dramatically increased
number of components, specifically for:
• low efficiency that can be estimated by
k
η=
U in ⋅ I in − ∑ p x ( a x )
x =1
U in ⋅ I in
,
(1)
where Uin is an input voltage value, Iin is an input
current, px is the power loss of device x, and ax is the
relative on-state time (a=0...1) of device x [1];
• decreased long-term reliability;
• increased complexity of control and protection
circuits;
• increased dimensions and weight of the converter.
SIMPLIFIED TOPOLOGY FOR THE HIGH-VOLTAGE APS
Although such terms like simplicity, low cost of ownership,
small dimensions and low weight are the main driving forces
in the modern power electronic converter development, special
railway norms and requirements involve own orders. While
compliance with the high input voltage level was discussed in
the sections above, such item as compliance with the safety
requirements must be emphasized separately. Briefly, it means
that the high voltage input and low voltage output stages of
APS need to be galvanically isolated. Since there is a need of
the isolation transformer, certain adjustments are to be made in
the design of the power circuit of the APS, i.e. no
transformerless DC/DC converter topologies are accepted for
such demanding applications. Thus, the general field of choice
is basically limited by most widespread two-level topology - a
full-bridge (FB) isolated DC/DC converter.
Another possible design - a half-bridge (HB) isolated
DC/DC converter topology has always been referred to as a
very attractive topology for different low-power applications
like telecommunication facilities, fuel cell based power
generation systems, compact power supplies and other
applications with the power range of 1...2 kW. The advantages
of the HB topology are obvious:
• only two primary switches are required - being based
on new 6.5 kV IGBT, a drastically simplified power
circuit design can be achieved;
• half-bridge inverter output voltage rating is reduced to
half of the input - simplified isolation transformer;
• no centre-tapped transformer is required for the input
stage.
In the present paper HB topology will be first examined as
a candidate topology for the auxiliary power supply (APS) to
be used in 3.0 kV DC commuter trains. It is evident that for
the APS with an output power up to 50 kW, the half-bridge
topology and two 200 A 6.5 kV IGBTs (one for the TOP and
the other for the BOT switch) is sufficient to fulfill all the
design requirements (Fig. 4).
2200...4000 V DC
R IN
IIN
HV IGBT
MODULE 1
U in/2
C1
ISOLATION
TRANSFORMER
0
U in
DT
TT
Cf
HV IGBT
MODULE 2
TX
U in/2
C2
DB
TB
A. Control of a high-voltage half-bridge inverter
Half-bridge inverter is very simple in construction, but
special attention must be paid to its control and protection
functions. The input leads of APS are connected to the traction
supply grid with voltage tolerances from 2200 V DC (Uin(min))
up to 4000 V DC (Uin(max)). The most demanding operation
point is at the minimum input voltage and at the rated load
conditions (maximum duty cycle operation). It is essential to
prevent even short-time simultaneous conduction of TT and
TB switches in these demanding conditions - it leads to the
short circuit across the supply voltage and to the destruction of
the converter. It means that the maximum on-state time ton(max)
of each switch in the half-bridge must be set at 80% of a halfperiod to ensure that this does not happen. Pulse width at the
maximum input voltage is minimal and may be determined as
U in (min)
2200
⎛T ⎞
t on (min) =
t on (max) =
t on (max) = 0.44⎜ sw ⎟ (2)
U in (max)
4000
⎝ 2 ⎠
Operation voltage ranges and inverter switch on-state times
of the half-bridge topology in this application are presented in
TABLE II. Simulated isolation transformer supply voltage
waveforms with the different converter input voltages and at
rated load are presented in Fig. 5.
TABLE II.
OPERATION VOLTAGE RANGES VS. INVERTER SWITCH ON-STATE TIME
2200 V
4000 V
Converter input voltage Uin
Switch on-state time ton
0.8(Tsw/2)
0.44(Tsw/2)
1.2KV
0.8KV
D2
D1
D4
D3
0.4KV
0.0KV
U REC
-0.4KV
Lo
-0.8KV
Co
-1.2KV
3.3ms
U OUT
350 V DC
40...150A
3.6ms
V(1,2)
4.0ms
ton
D=0.8(Tsw/2)
4.4ms
4.8ms
5.2ms
5.6ms
6.0ms
6.3ms
5.6ms
6.0ms
6.3ms
Time
(a)
2.5KV
2.0KV
Fig. 4. Power circuit layout of the proposed APS.
The half-bridge isolated DC/DC converter configuration
consists of two equal capacitors C1 and C2 connected in series
across the input voltage source (Uin), providing a constant
potential of one-half Uin at their junction (Fig. 4). [3] The HV
IGBT switches TT (top switch) and TB (bottom switch) are
turned on alternately and are subjected to a voltage stress equal
to that of the input voltage. Concerning another two-switch
DC/DC converter topology, a double-ended forward converter,
an advantage of the half-bridge is that its secondary produces a
full-wave output rather than a half-wave output. Thus, the
square-wave frequency in the half-bridge converter is twice
that of the forward converter and the associated output filter
components will be smaller. In addition, considering specific
railway demands for better reliability and maintainability of the
power electronic converters, the half-bridge topology with its
reduced component number seems to be a more preferable
choice compared to the full-bridge topology, for instance.
1.0KV
0V
-1.0KV
-2.0KV
-2.5KV
3.3ms
V(1,2)
3.6ms
4.0ms
tonD=0.44(Tsw/2)
4.4ms
4.8ms
5.2ms
Time
(b)
Fig. 5. Simulated isolation transformer supply voltage waveforms for the
maximum (a) and minimum (b) duty cycle operation.
The measured waveforms of the isolation transformer
supply voltage during the different operation conditions are
presented in Fig. 6. It can be noticed that the regular voltage
spikes occur during the transistor turnoff. These spikes are
mostly caused by the leakage inductance of the isolation
transformer. At the instant of turnoff, current in the transistor
falls rapidly at a rate dI/dt causing a positive-going spike of
amplitude Esp=Llk(dI/dt) at the bottom end of the leakage
inductance Llk. These leakage inductance spikes, which are so
troublesome in several topologies (forward and push-pull
topologies [4]) are basically clamped to the Uin by the
freewheeling diodes DT and DB (see Fig. 4).
(a)
(b)
For the converter weight-space optimization the switching
frequency fsw of HV IGBT in this application was 1 kHz.
Average losses of half-bridge inverter in that case are presented
in Fig. 7.
B. Isolation transformer and output stage
The isolation transformer of a rolling stock APS is
responsible for providing the necessary I/O galvanic isolation
required by special safety norms. For the 3.0 kV DC-fed
converters, the required isolation barrier is 10.2 kV/1 min. Due
to the capacitors providing a mid-voltage point, the isolation
transformer during the operation sees a positive and negative
voltage with the amplitude value of only half the input voltage
(i.e. 1100...2000 V, see Fig. 5). This results in twice the
desired peak flux value of the core, because the transformer
core is operated in the first and third quadrant of the B-H loop
and experiences twice the flux excursion of a similar forward
converter core. This is an advantage of the half-bridge
topology over the double-ended forward topologies, where the
half-bridge primary transformer winding has half the turns for
the same input voltage and power.
With the selected switching frequency of the 6.5 kV IGBT
of 1 kHz the isolation transformer can be regarded as the
bulkiest component in the whole converter stack. Considering
the design, in this demanding application, toroidal transformers
outpace the laminated ones. But using the toroidal transformers
for high power densities, the minimization of core losses by
design optimization and material development is important. For
this project, the Gammamet toroidal magnetic cores, made
from 25 μm thick ribbon of soft magnetic nanocrystalline alloy
on Fe-basis were investigated.
The real advantage of the selected magnetic cores Gm14DC
are their high initial permeability and very low core losses [5].
The rate of specific core power dissipation for the selected core
type (Gm14DC) can be found by (3).
PD′ ( core ) = 0.75 f 1.4 BM1.7
Fig. 6. Measured isolation transformer supply voltage waveforms for the
maximum (a) and minimum (b) duty cycle operation.
Here it must be stated that for rolling stock APS converters
with throughput power ratings up to 50 kW the 200 A 6.5 kV
IGBT module is fully compliant. However, the switching
dynamics of the 6.5 kV IGBTs is significantly reduced (as
compared to 1.2 kV or 1.7 kV IGBT modules) due to the effect
of the absolute voltage vs. current. However, switching can be
controlled to achieve a good compromise between allowable
switching frequency and switching power loss.
IGBT turn-on
losses, 600 W
(39%)
FWD losses,
400 W (26%)
(3)
With the assumed maximum flux density of 0.4 T the
average core power dissipation for the 50-kW isolation
transformer is about 25 W. Estimated winding losses for this
particular design are not higher than 150 W (see Fig. 8).
Magnetic core
losses, 24 W
(17%)
Primary winding
losses, 69 W
(47%)
Secondary
winding losses,
52 W (36%)
Fig. 8. Isolation transformer power loss distribution
IGBT
conduction
losses, 135 W
(9%)
IGBT turn-off
losses, 400 W
(26%)
Fig. 7. Average losses of half-bridge inverter (2 x IGBT FZ200R65KF1) at
1 kHz switching frequency
Half-bridge converter output voltage can be defined by the
following equation [4]:
2 ⋅ ton .
(4)
U OUT = U REC ⋅
Tsw
Thus, the amplitude of the rectifier output voltage (UREC in
Fig. 4) can be estimated as
U REC = U OUT ⋅
Tsw .
2 ⋅ ton
(5)
For the described application, the converter output voltage
must be carefully regulated to 350 V DC despite the voltage
fluctuations on the input side. Boundary switch on-state times
and the corresponding amplitude values of UREC are shown in
TABLE IV. Simulated waveforms of UREC for different
converter operation points are presented in Fig. 9.
TABLE IV.
AMPLITUDE VALUES OF UREC FOR DIFFERENT OPERATION POINTS
2200 V
4000 V
Converter input voltage Uin
Switch on-state time ton
0.8(Tsw/2)
0.44(Tsw/2)
Amplitude value of UREC
438 V
795 V
500
UREC [V]
400
300
200
100
0
IREC [A]
-100
12.0ms
12.4ms
12.8ms
13.2ms
13.6ms
14.0ms
14.4ms
14.8ms
15.2ms
15.6ms 16.0ms
Time
ton=0.8(TSW /2)
(a)
900
UREC [V]
800
600
400
200
0
IREC [A]
-100
12.0ms
12.4ms
12.8ms
ton=0.44(TSW/2)
13.2ms
13.6ms
14.0ms
14.4ms
14.8ms
15.2ms
15.6ms 16.0ms
Time
(b)
Fig. 9. Simulated waveforms of UREC for different operation points.
It is shown in Fig. 9 how the output rectifier diodes for the
converters with such a wide input voltage range must be
selected to satisfy the two basic criteria:
• maximum possible repetitive reverse voltage URRM
(operation with high input voltage level is considered here),
• maximum possible average forward current IFAVM and
minimized forward voltage drop, UF (considering operation
with minimal input voltage and maximal duty cycle).
The selection of rectifying elements greatly influences the
efficiency and the performance of the whole converter. To
illustrate possible rectifier diode options, general
specifications of on-market available diode types were
compared in TABLE V (based on the production of company
IXYS [6]).
TABLE V.
EVALUATIVE COMPARISON OF DIFFERENT RECTIFIER DIODES
UF
trr
IF
URRM
1.6...2.2
1.2......2.2
56...1800
Silicon diode
0.5...15 us
kV
V
A
Silicon Schottky
200 V
0.3...0.7 V
25 ns
200 A
diode
SiC Schottky
600 V
1.5 V
20 ns
11 A
diode
1.2...1.6
180...453
FRED diode
0.6...1.8 V 40...450 ns
kV
A
The widely used common silicon rectifier diode has good
voltage and current ratings. However, an unacceptably high
recovery time cancels their benefits in high frequency
applications. The new challenge is the replacement of the
traditional silicon diodes by more advanced ones. Thus,
Schottky diodes or fast switching diodes (fast recovery,
ultrafast recovery, etc.) have already achieved remarkable
benefits. Both of the presented types have essentially lower
voltage drops in the on-state condition. For instance, the
0.3…0.7 V voltage drop of a Schottky diode (see TABLE V)
and about 0.6…1.8 V voltage drop in the case of fast recovery
diodes help to improve the efficiency of the rectifier stage by
15...35 %. New SiC-based diodes still suffer from material
problems and the additional disadvantages like high forward
voltage drop UF=1.5...5 V are outpace such advantageous
moments as improved switching dynamics and high possible
operation temperatures [6], [7]. Another problem is the
temperature mode of the output rectifier. Schottky barrier
diode is extremely sensitive to temperature because of its high
leakage characteristics over its operating range. The PN
junction diode (typically Fast Recovery Epitaxial Diode,
FRED) has comparatively lower leakage at higher temperature
than the Schottky diode, but the forward voltage is
considerably higher. The high forward voltage translates to a
higher power loss in the diode, thus creating a larger amount
of heat within the diode. This also lowers the overall
efficiency of the power supply. Thus, by exchanging the
reduced chances of thermal runaway caused by using Schottky
diodes by replacing it with PN junction diodes, the designer
reduces the overall efficiency of the system. For power supply
designers, efficiency is a very undesirable trade-off for
thermal stability [8].
Thus, for the high-frequency operation with voltages about
1000...1100 V (transient overvoltages are considered here) and
output currents of about 150 A, fast recovery diodes are an
absolutely dominant solution. For the presented project FRED
diodes were selected as more economical and technically
feasible.
IV.
EFFICIENCY ESTIMATION OF THE PROPOSED
CONVERTER
Switchmode converter losses are generated in passive
(resistors, capacitors, inductors, transformers, etc.) and active
components (transistors, diodes, etc.). Losses in passive
components are usually small compared with other types.
Thus, the losses in the switching elements are decisive. Losses
of primary high-voltage IGBTs are basically the function of
the load current, operation voltage and frequency. Losses of an
output rectifier are load current dependant. The efficiency of
the developed converter may be estimated by help of (1). The
“worst case” efficiency of the converter can be estimated at
the maximum input voltage and at rated load. With the
switching frequency of 1 kHz and ton=0.22 ms the efficiency is
about 93%, which is quite a good result for such devices with
high-frequency hard-switching. The breakdown of losses is
presented in Fig. 10. Actually, losses in input resistor Rin are
not considered here but they give an extra 1...2%.
Isolation
transformer
losses
5%
Filter losses
13%
(b )
Other losses
2%
Rectifier
diode losses
19%
Fig. 11. Cascaded inverter with 28 conventional IGBTs (a) compared to
proposed half-bridge inverter (b) with two 6.5 kV IGBT modules.
IGBT losses
61%
Fig. 10. Total losses breakdown of the proposed converter at maximum input
voltage and rated load.
In the design of high-power converters with high-voltage
IGBTs, main attention must be paid to the proper selection of
a switching frequency of IGBTs. Here such parameters as
compactness, EMI level, audible hum, and others must be
analyzed. In the majority of applications, designers use the
hard-switching mode of HV IGBT, which is more optimal in
terms of power circuit complexity and maintainability.
However, implementing soft-switching, the IGBT losses can
be reduced significantly, down to 8% (turn-on) or 30% (turnoff) of the hard-switching level. [9] Overall, this gives an
effect of 20-30% reduced IGBT losses, which means that the
cooling effort and therefore costs can be considerably reduced.
Otherwise, the switching frequency of IGBTs can be increased
to obtain the same power dissipation per IGBT as with the
hard-switching.
V.
(a )
TABLE VI.
COMPARISON OF DIFFERENT POWER CIRCUIT TOPOLOGIES
Cascaded
Simplified
converters or
HV IGBT
ser. switches
half-bridge
Reliability
+
Number of parts
+
Ruggedness of switching elements
+
+
Cost for power semiconductor modules
+
Effort for control and measurement
+
Decentralization of heat sources
+
“+” advantage of the system
“-” drawback of the system
For practical investigations an optically isolated HV-IGBTbased 50 kW isolated half-bridge DC/DC converter prototype
has been designed and tested.
REFERENCES
[1]
[2]
[3]
[4]
[5]
CONCLUSIONS
This paper has proposed a simplified power scheme
topology to be used in 3.0 kV DC commuter trains. The
implementation of new 6.5 kV IGBT transistors in the primary
half-bridge inverter helps to solve common problems of
converter power circuit complexity and reliability (Fig. 11).
TABLE VI shows a survey of important facts about APS
topologies compared. Although application of the new HV
IGBT has its own pros and cons, further optimization of power
scheme layouts and IGBT control strategies will ensure a
novel quality level of power electronic converters for rolling
stock.
[6]
[7]
[8]
[9]
D. Vinnikov, „Research, Design and Implementation of Auxiliary Power
Supplies for the Light Rail Vehicles”, Ph.D. dissertation, Dept. El.
Drives Pow. Elec., Tallinn Univ. Tech., Estonia, 2005.
T. Schuetze, H. Berg, O. Schilling, "The New 6.5kV IGBT. Module: A
Reliable Device for Medium Voltage. Applications", PCIM 2001,
Nuremberg, Germany.
Brown, Jess; Davies, Richard; Williams, Dilwyn and Bernacchi, Jerry.
“High-Efficiency Half-Bridge DC-to-DC Converters with Secondary
Synchronous Rectification,” PCIM 2001, Nuremberg, Germany.
A. I. Pressman, Switching Power Supply Design. Second Edition,
McGraw-Hill, 1998.
Magnetic Properties of Tape Wound Encapsulated Cores
GAMMAMET®14DC, GAMMAMET Research & Production
Enterprise, 2006.
IXYS semiconductors databook - 2004/2005.
Morisette, D.T., Cooper, J.A., Melloch, Jr., Dolny, M.R., Shenoy, G.M.,
Zafrani, M. and Gladish, J. Static and dynamic characterization of largearea high-current-density SiC Schottky diodes. IEEE Transactions on
Electron Devices. Vol. 48, issue 2, pp.: 349 - 352, Feb. 2001.
Yueqing Wang, Qingyou Zhang, Jianping Ying, Chaoqun Sun,
Prediction of PIN diode reverse recovery, Power Electronics Specialists
Conference, 2004. PESC’04. 2004 IEEE 35th Annual Volume 4, 2004
Page(s):2956 - 2959 Vol.4
Schwarzer, U.; De Doncker, R.W. „Characterization of 6.5 kV IGBT
modules for hard- and soft-switching operation in medium voltage
applications”, in Proc. Twentieth Annual IEEE Applied Power
Electronics Conference and Exposition, 2005. APEC 2005. pp.: 329 335, vol. 1.