RF MEMS-Based Tunable Filters
James Brank, Jamie Yao, Mike Eberly, Andrew Malczewski, Karl Varian,
Charles Goldsmith
Raytheon Systems Company, P. O. Box 660246 MS 35, Dallas, Texas 75266
Accepted 17 May 2001
ABSTRACT: This paper overviews the application of RF MEMS switches in tunable filters
as well as circuit developments for bandpass filters covering 110 MHz to 2.8 GHz. RF MEMS
have several desirable features, including small size, low power requirements, and low loss.
The basic operation of Raytheon’s RF MEMS capacitive membrane switch is described. An
overview of the technique used to integrate the switch into a variable capacitor structure with
sixteen capacitance states is provided. Variable capacitor structures are used to construct
multipole lumped bandpass filter designs, each with sixteen states. Finally, measured data
from two representative five- and six-pole bandpass filters are presented. Characterization
data demonstrates that the insertion loss for the five-pole filter using on-chip inductors was
between 6.6 and 7.3 dB, and between 3.7 and 4.2 dB for the six-pole filter using off-chip
inductors. © 2001 John Wiley & Sons, Inc. Int J RF and Microwave CAE 11: 276–284, 2001.
Keywords: MEMs; microelectromechanical system; tunable capacitors; varactors; membrane
capacitor; tunable filter; tunable bandpass filter
I. INTRODUCTION
Filters are the basic building blocks within frequency converting systems such as receivers and
tuners. At microwave frequencies (1 GHz and
above), filters are composed of high-Q resonators
such as printed transmission line, suspended rods,
or dielectric pucks. Depending on the media
used to create these resonators, excellent performance can be achieved with Qs in the hundreds
for printed lines to tens of thousands for dielectric resonators. The need for frequency tunability
within broadband receiving and transmitting systems usually necessitates switching of multiple
fixed-tuned circuits. The use of tunable filters and
resonators can significantly simplify complexity
and reduce losses within complex multiband systems. Unfortunately, there is not yet a tunable
Correspondence to: James Brank
Contract grant sponsor: Raytheon.
Contract grant number: DARPA F 30602-97-C-D1 8la,
© 2001 John Wiley & Sons, Inc.
276
resonator component that affords the high performance achieved by fixed resonators. YIG filters
come the closest to having very good filter selectivity, but at the expense of being bulky, requiring
significant quiescent current, and being expensive.
To date, diode varactor-tuned circuits, though
simple and requiring little bias current and size,
have not met the expectations of most modern
receiver requirements in terms of loss. As such,
inexpensive and high performance tunable resonators have become one of the “holy grails” of
receiver components.
The advent of microelectromechanical systems (MEMS) for radio-frequency (RF) applications provides new possibilities for achieving
the desired characteristics of a tunable resonator.
RF MEMS devices, a new paradigm in the construction of electronic devices, created mechanical structures on the microscale. Being constructed entirely of low-loss metals and dielectrics,
these mechanical structures inherently have low
loss.
RF MEMS-Based Tunable Filters
Development of RF MEMS switches has been
under way seriously since about 1995 from several
industrial and university groups. These devices
have distinguished themselves as having very low
loss, requiring practically no power consumption,
and having very high linearity. The application
of RF MEMS has already proven to provide
revolutionary (rather than evolutionary) improvements in electronic switching performance for
phase shifters at microwave and millimeter-wave
frequencies.
This paper explores the use of RF MEMS
capacitive switches in the application of tunable filters. Since these devices are operated in a
bistable manner, with either a high or low capacitance, they are a natural device for accomplishing
digital frequency selection within a filter. The
capacitive membrane switch is used to create a
multibit variable capacitor which serves as a digital varactor. This varactor is in turn used within
resonators and coupling circuits to create tunable,
lumped-element LC filters for receiver front-end
applications.
II. MEMS VARIABLE CAPACITOR
CONSTRUCTION/ ELECTRICAL
PERFORMANCE
To create variable capacitors, fixed MIM capacitors are combined in series with RF MEMS capacitive switches as shown in Figure 1. This creates a
two-state capacitor whose value is set by the series
combination of the fixed cap and the capacitance
of the RF MEMS. The minimum value of the twostate capacitor is limited by the off-capacitance of
the MEMS, and the maximum value is limited
by the on-capacitance of the RF MEMS. Generally, the value of the fixed cap is kept below
the on-capacitance of the MEMS switch to minimize the effect of MEMS variation. Combinations
of these two-state capacitors with fixed capacitors
allow construction of variable capacitor structures,
ref. [1].
Top View
Membrane
Undercut
Access
Holes
Signal
Path
Lower
Electrode
Dielectric
Post
Cross Section
Dielectric
High resistivity silicon
Figure 2. Views of the RF MEMS capacitive switch.
[Color figure can be viewed in the online issue, which is
available at www.interscience.wiley.com.]
The basic RF MEMS capacitive switch is
shown in Figure 2. The structure is basically a
parallel-plate capacitor with a movable top plate.
Applying a voltage between the membrane (top
plate) and electrode (bottom plate) creates an
electric field. When the field is strong enough,
the membrane will flex downward and contact the
dielectric. A simple electrical model of the switch
is shown in Figure 3. Typical on capacitance
(membrane down) is 3 pF, and off capacitance
(membrane up) is 30 fF, [2, 3].
The fixed capacitor used were metal-insulatormetal (MIM) capacitors, as shown in Figure 4.
Both top and bottom plates of the capacitor are
gold, with a thin silicon nitride dielectric layer
(r = 68). Average capacitance, resistance, and
Q at 1 GHz for four MIM test structures are
shown in Table I. It should be noted that when
measuring high-Q devices it can be difficult to
extract accurate values for Q. The Q values are
RF
MEMS
RSE
CON/OFF
RSH
Figure 1. Schematic of a two-state variable capacitor
using RF MEMS.
Electrode
Buffer Layer
RSE
Fixed
Cap
277
RSH
Figure 3. Simple schematic of the RF MEMS capacitive switch. [Color figure can be viewed in the online
issue, which is available at www.interscience.wiley.com.]
278
Brank et al.
Control
Lines
C4
C3
C2
C1
C0
Figure 5. Schematic of a four-bit variable capacitor
using RF MEMS.
Figure 4.
Layout of a MIM capacitor.
very sensitive to small errors in R and can be
difficult to extract. Other measurements of similar capacitors yielded Qs that varied by as much
as ±100%. This large variation depends upon
the specific measurement conditions, such as the
calibration method used or the condition of the
RF probes.
The schematic of a four-bit variable capacitor is shown in Figure 5. It consists of five fixed
capacitors, four of which are in series with an RF
MEMS capacitive switch. A layout of a four-bit
variable cap is shown in Figure 6. Depending
on which combination of switches are actuated,
the capacitance across the variable capacitor can
be set. In the design of a variable capacitor, the
fixed capacitors are designed to give even steps
of capacitance between the minimum and maximum required values of capacitance. The variable
capacitor layout shown in Figure 6 incorporates
some subtle improvements over the previous versions of variable capacitors. The fixed capacitor
values are shown in Table II. The variable capacitor structure is “straightened out” in order to
place the larger capacitor states closer to the
RF signal path. This was done to minimize the
series inductance. Reducing the series inductance
of the capacitors is especially important in the
TABLE I. Average Measured Resistance, Capacitance,
and Q for MIM Capacitor Test Structures
Nominal C pF
0.56
1.06
5.11
10.28
R, ohms
(@ 1 GHz)
1.60
0.60
0.10
0.13
Q
(@ 1GHz)
179.08
251.74
302.40
120.19
Figure 6. Photograph of a variable capacitor. [Color
figure can be viewed in the online issue, which is available at www. interscience.wiley.com.]
larger capacitors, as unwanted self resonance can
degrade the frequency response of a filter. A side
benefit of this rearrangement is that the digital control lines can easily be routed to the RF
MEMS devices without having to cross over any
of the RF paths.
Measured data for a typical variable capacitor
is shown in Figure 7. The capacitance increases
in even steps except for the step between steps 7
and 8. The C4 capacitor value was slightly too
large, which caused the gap in capacitance values. The capacitance is very flat versus frequency,
which is due to the high self-resonant frequency
of the structure.
TABLE II. Fixed Capacitor Values for
the Circuit Shown in Figure 6.
C0
C1
C2
C3
C4
3.08 pF
0.231 pF
0.498 pF
1.18 pF
3.88 pF
Capacitance, pF
RF MEMS-Based Tunable Filters
7.00
6.50
6.00
5.50
5.00
4.50
4.00
3.50
3.00
2.50
2.00
0.050
279
then the variable capacitors must be roughly capable of tuning
Cmax
= α2
Cmin
0.100
0.150
0.200
0.250
Frequency, GHz
Figure 7. Capacitance versus frequency for the sixteen
states of the variable capacitor.
III. TUNABLE FILTER TOPOLOGY
Of the many filter topologies available, only a few
are amenable to construction of a tunable filter.
The capacitively coupled LC resonators bandpass
filter, shown in Fig. 8, was chosen as the baseline design [4 5]. The redundancy of the elements
allows an extra degree of freedom in the choice
of component values, which allowed the inductance to stay constant as the center frequency is
tuned.
Choosing the component values is challenging,
and several design and simulation iterations are
necessary. The designer is given electrical requirements, such as frequency range over which the
filter must tune, bandwidth, insertion loss, and
number of tunable states, that must be satisfied.
Unfortunately, it may be impossible to simultaneously satisfy all requirements. For example, if a
constant bandwidth is required, the insertion loss
of the filter will vary across the tuning range. If a
constant insertion loss is required, the bandwidth
must vary across the tuning range [6].
Choice of the inductor is determined primarily by the Q of available devices, as in a fixed
filter design. An added constraint is that if a
tunable filter using fixed inductors has a tuning
range of
α=
Cs01
fmax
fmin
L1
IV. TUNABLE FILTER REALIZATIONS
The designs presented here incorporate the
improvements in variable capacitor design
Cs12
Cp1
Figure 8.
The implications can be seen in the following
figures. Two tunable filters were designed. Both
were five-pole capacitively coupled 0.1 dB Chebyshev bandpass filters with sixteen states and fixed
180 MHz bandwidth but with different tuning
ranges. Both filters used a fixed 2 nH inductor. Using a spreadsheet, the series and shunt
capacitances of Figure 8 were calculated for
each frequency step. These values are shown in
Figures 9 and 10. The starting point for these
calculations is given in ref. [1].
The filter of Figure 9 had a tuning range of
885 to 986 MHz, or an 11% tuning range. The
series and shunt capacitor values versus tuning state have very slight curvature, and can be
approximated quite well with a straight line. This
works well with the variable capacitor structure
described in Section III. Step size for this filter will be even, and bandwidth will be relatively
constant across the tuning range.
The filter of Figure 10 had a tuning range of
996 to 2068 MHz, or a 108% tuning range. The
series and shunt capacitor values versus tuning
state have a noticeable curvature, and a linear
approximation is not as good. In practice, the
variable capacitors will be designed to have the
correct value at the maximum and minimum tuning ranges, and vary linearly with state between
the extremes. Step size for this filter will be
uneven, with larger steps at the high end of the
tuning range. As the values of the coupling capacitors are somewhat flat, the bandwidth will be
roughly constant across the tuning range, with
a slightly larger bandwidth in the center of the
band.
L2
Cs23
Cp2
L3
Cs34
Cp3
L4
Cs45
L5
Cp4
Schematic of a capacitively coupled five-pole bandpass filter.
Cs56
Cp5
280
Brank et al.
14
Capacitance, pF
12
10
Cp1, Cp5
8
Cs12,Cs45
6
Cp2, Cp4
Cs23, Cs34
4
Cp3
2
Cs01, Cs56
0
885
935
985
Center Frequency, MHz
Figure 9. Variation of capacitance with tuning state for a five-pole filter, fc = 885–986 MHz. [Color figure can be
viewed in the online issue, which is available at www. interscience.wiley.com.]
Capacitance, pF
described earlier, as well as processing improvements related to the metalization. These designs
have improved electrical performance, have a
better control structure, and demonstrate the
ability to incorporate these devices into higher
level assemblies. Improved wafer processing also
reduced the filter insertion loss and improved
device quality.
To date, seventeen different tunable filters
using RF MEMS have been built. They range in
complexity from simple one-pole structures with
9
8
7
6
5
4
3
2
1
0
996
twelve MEMS devices to six-pole filters with 139
MEMS devices. Both on-chip and off-chip inductors have been studied. The filter frequencies
covered 70 MHz to 2.8 GHz. As space is limited,
only two representative designs will be discussed.
UHF Tunable Filter
A five-pole 0.1 dB Chebyshev bandpass filter
design, denoted as the UHF filter, is presented
Cp1, Cp5
Cs12, Cs45
Cs23, Cs34
Cp3
Cp2, Cp4
Cs01, Cs56
1496
1996
Center Frequency, MHz
Figure 10.
l=2057 um
w=125 um
0.075 pF
0.12 pF
0.12 pF
0.24 pF
0.24 pF
0.32 pF
1.62 pF
0.71 pF
0.52 pF
0.52 pF
0.71 pF
8.55 pF
5.32 pF
Schematic for the UHF filter.
0.22 pF
7.21 pF
3.43 pF
0.17 pF
1.87 pF
6.96 pF
0.80 pF
2.13 pF
0.39 pF
0.89 pF
0.42 pF
1.62 pF
7.69 pF
0.17 pF
1.87 pF
0.80 pF
0.22 pF
3.43 pF
6.96 pF
Figure 11.
0.15 pF
0.54 pF
1.22 pF
1.62 pF
2.13 pF
7.21 pF
l=2057 um
w=125 um
0.29 pF
0.65 pF
0.32 pF
5.32 pF
0.54 pF
1.22 pF
l=2057 um
w=125 um
0.075 pF
0.15 pF
0.17 pF
8.55 pF
l=2057 um
w=125 um
0.06 pF
2.16 pF
0.65 pF
1.62 pF
l=2057s
w=125 um
0.06 pF
0.39 pF
0.29 pF
Variation of capacitance with tuning state for a five-pole filter, fc = 996–2068 MHz.
RF MEMS-Based Tunable Filters
281
Figure 12. Layout of the UHF five-pole filter. Die size is 3.5 mm by 14 mm. [Color figure can be viewed in the
online issue, which is available at www. interscience.wiley.com.]
VHF FILTER
A six-pole 0.1 dB Chebyshev bandpass filter,
denoted as the VHF design, is presented here.
This filter had a center frequency tuning range of
110 MHz to 160 MHz with a variable bandwidth
from 37 MHz to 58 MHz. A lumped element
design was derived for the maximum and minimum tuning frequencies with constant resonator
inductances of 27 nH. Off-chip inductors were
used in this case because the required value of
inductance was too large to incorporate on-chip.
Also, the improved Q of the off-chip inductors
improved the insertion loss compared to the onchip designs. As with the previous design, four-bit
variable capacitors were designed to cover the
required tuning ranges based on the lumped
designs. Some of the larger capacitor states used
multiple MEMS devices to switch large capacitancies. The schematic and layout are shown in
Figure 14 and Figure 15.
The measured insertion loss and return loss
of the five-pole filter is shown in Figure 16.
Center frequency insertion loss across all tuning states was from 3.7 to 4.2 dB. Return loss
0
10
-5
5
-10
0
-15
-5
-20
-10
-25
-15
-30
-20
-35
-25
Return Loss (dB)
Insertion Loss (dB)
here. This filter had a center frequency tuning
range of 885 MHz to 986 MHz with a constant
bandwidth of 180 MHz. As described earlier, the
capacitively coupled LC resonator design was
chosen. A lumped element design was derived
for the maximum and minimum tuning frequencies with constant resonator inductances of 2.9
nH. On-chip inductors were used to demonstrate
the potential for integration of entire filters on
a chip, with the path to ground provided by ribbon bonds to the carrier plate. Four-bit variable
capacitors were designed to cover the required
tuning ranges based on the lumped designs. The
schematic and layout are shown in Figure 11 and
Figure 12.
The measured insertion loss and return loss
of the five-pole filter is shown in Figure 13. The
filter tuned from 880 to 992 MHz, with the center frequency insertion loss across all tuning
states from 6.6 and 7.3 dB. Measured bandwidth
varied between 168 and 174 MHz. Return loss
was better than 10 dB for all tuning states. The
bandwidth and passband shape stayed relatively
constant as the center frequency of the filter was
tuned.
-30
-40
0.6
0.7
0.8
0.9
1
1.1
1.2
1.3
1.4
Frequency (GHz)
Figure 13. UHF frequency response. [Color figure can be viewed in the online issue, which is available at www.
interscience.wiley.com.]
282
Brank et al.
90.0 pF
12.
99
0
pF
6.4 1.5 0.6 9.9
90 60 20 20
pF pF pF pF
127.45 pF
127.45 pF
27
nH
27
nH
43.290 pF
43.290 pF
14.620 pF
14.620 pF
5.20 pF
5.20 pF
24.150 pF
24.150 pF
12.
99
0
pF
6.4 1.5 0.6 9.9
90 60 20 20
pF pF pF pF
12.
99
0
pF
6.4 1.5 0.6 14.
90 60 20 71
pF pF pF 0
pF
71.73 pF
71.73 pF
27
nH
27
nH
28.34 pF
28.34 pF
10.78 pF
10.78 pF
4.15 pF
4.15 pF
20.92 pF
20.92 pF
12.
99
0
pF
6.4 1.5 0.6 14.
90 60 20 71
pF pF pF 0
pF
12.
99
0
pF
6.4 1.5 0.6 9.9
90 60 20 20
pF pF pF pF
was better than 15 dB for all tuning states. The
passband shape stayed relatively constant as the
center frequency of the filter was tuned, while the
bandwidth increased as the center frequency was
tuned higher. This resulted in the filter having
less insertion loss at the highest tuning state than
at the lowest tuning state.
When compared to a conventional switchedfilter bank, the advantages of RF MEMS-based
filters are remarkable. Analysis indicates that compared with a typical switched-filter bank, use of
RF MEMS tunable filters allow a 60X reduction in
size, 150X reduction in weight, and a 10X reduction in the number of RF support switches. A
single MEMS-based filter can have sixteen tunable states, replacing sixteen fixed frequency filters. The low power requirements of RF MEMS
can reduce filter assembly power requirements 8X.
Such size, weight, power, and circuit complexity
reductions are crucial in modern communications
designs.
V. CONCLUSION
Two tunable bandpass filters designs using RF
MEMS were demonstrated. Insertion loss for the
five-pole UHF filter with on-chip inductors was
measured to be between 6.6 and 7.3 dB, and
between 3.7 and 4.2 dB for the six-pole VHF filter with off-chip inductors. Both filters exhibited
good return loss across the tuning range. Passband shape was also maintained across the tuning
range. With their high degree of integration, RF
MEMS show great potential for weight, power
consumption, and size reduction. Ongoing design
and process improvements will reduce the insertion loss further, as well as extend the operating
frequency range.
127.45 pF
ACKNOWLEDGMENTS
127.45 pF
27
nH
27
nH
43.290 pF
43.290 pF
14.620 pF
14.620 pF
5.20 pF
5.20 pF
24.150 pF
24.150 pF
90.0 pF
Figure 14.
VHF filter schematic.
Raytheon, RF MEMS group, DARPA contract number
F30602-97-C-0186.
REFERENCES
1. Charles L. Goldsmith, Andrew Malczewski, Zhimin
Jamie Yao, Shea Chen, and David Hinzel, RF
MEMS variable capacitors for tunable filters, International Journal of RF and Microwave ComputerAided Engineering, 9 (1999), 362–374.
2. Zhimin Jamie Yao, Shea Chen, Susan Eshelmann, David Denniston, and Charles Goldsmith,
Micromachined low-loss microwave switches, IEEE
microelectromechanical systems, 8 (1999), 129–134.
RF MEMS-Based Tunable Filters
283
0
10
-5
5
-10
0
-15
-5
-20
-10
-25
-15
-30
-20
-35
-25
-40
0.05
Return Loss (dB)
Insertion Loss (dB)
Figure 15. Layout of the VHF five-pole filter. Die size is 4 mm by 16 mm. [Color figure can be viewed in the online
issue, which is available at www. interscience.wiley.com.]
-30
0.1
0.15
0.2
0.25
0.3
Frequency (GHz)
Figure 16. VHF frequency response. [Color figure can be viewed in the online issue, which is available at www.
interscience.wiley.com.]
3. Charles L. Goldsmith, Zhimin Jamie Yao, Susan
Eshelmann, and David Denniston, Performance
of low-loss RF MEMS capacitive switches, IEEE
microwave and guided waves letters, 8 (1998),
269–271.
4. Anatol I, Zverev, Handbook of filter synthesis,
Wiley, New York, 1967.
5. G.L. Matthaei, L. Young, and E.M.T. Jones,
Microwave filters, impedance-matching networks,
and coupling structures, McGraw-Hill, New York,
1964.
6. Thomas R. Cuthbert, Broadband direct-coupled
and matching RF networks, TRCPEP Publications,
Greenwood, AR, 1999.
BIOGRAPHIES
James Brank received his Bachelor’s
degree in Electrical Engineering from
Texas A&M University in 1982. He
received his Master’s degree in Electrical Engineering from Southern Methodist
University in 1987. From 1983 to 1987
he was employed at E-Systems, Garland
Division where he performed integration
and test of electronic warfare receivers.
From 1987 to 1988 he worked for Raytheon in Bristol, Tennessee where he designed components for the Standard Missile
2 program. From 1989 to the present, he has been employed by
Raytheon Systems Company (formerly the Defense Electronics Group of Texas Instruments) where he has been involved in
a wide variety of projects, ranging from the design of X-band
radar modules to the application of phased array antenna technology for cellular telephone Smart Antennas. Currently he is
designing low loss MEMS tunable bandpass filters.
Zhimin J. Yao received her Ph.D. from
the School of Materials Science and Engineering at Georgia Institute of Technology in 1995. She then worked as a post
doctoral research associate at the School
of Electrical Engineering, Cornell University for one year. Her research emphasis
was on silicon bulk micromachining. Dr.
Yao is currently working at Rockwell Science Center in Thousand Oaks, CA. Her research interests
284
Brank et al.
include design, fabrication and characterization of microelectromechanical systems.
Michael Eberly has been a member of
the IEEE since 1988 as a student member. He earned his BSEE from the University of South Florida in 1992. He was
with the United States Navy, on active
duty, from 1992 to 1994, transferring then
to the Naval Reserve. He has continued
with the Naval Reserve to the present and
currently holds the rank of Lieutenant. In
1994, he began studying part time for his Masters degree, while
teaching basic courses in Electrical Engineering Technology at
Tampa Technical Institute. In 1996, he was selected to study as
the Texas Instruments Fellowship Student at the University of
South Florida. He worked for Raytheon since the summer of
1997 as an engineering intern as part of the previously mentioned fellowship and permanently since August 1998. He was
awarded his Master’s Degree in Electrical Engineering in the
fall of 2000. He is currently working in the Applied Research
Laboratory at Raytheon on RF/MEMS.
Andrew Malczewski was born in Warsaw, Poland in May, 1973. He earned a
Bachelor’s degree in Electrical Engineering from the University of Texas at Arlington in 1996. Since 1996, he has been
involved in the design and development
of microwave and millimeter-wave circuits
for Raytheon Systems Company (formerly
the Defense Electronics Group of Texas
Instruments). He is also involved in the development of RF
MEMS technology for receiver and antenna applications. He
is presently pursuing his Master’s degree in Electrical Engineering. Karl Varian photo and biography not available.
Karl Varian photo and biography not available.