Design a simple high-voltage half-bridge switched
mode power supply (SMPS) by Daniel Marks
Draft document April 19, 2016
Building a mains-powered SMPS involves high voltages, mains voltages, and high currents that
are lethal. You should not attempt to follow the design in this document unless you are
thoroughly familiar with the dangers that working with mains voltage and high voltages entails.
This document does not inform the reader of all such dangers, and so the reader should have
previous experience at working with these risks before attempting to follow what is in this
document. The author takes no responsibility for damage to life, property, or any injury, and the
experimenter assumes all risk. The purpose of this document is to share and summarize the
findings of the author in his experiences of constructing a SMPS.
This document is for informational purposes only. The mention of any products or companies in this
document are for informational purposes only and do not constitute an endorsement on part of the
author.
Introduction
Modern MOSFETs (Metal oxide semiconductor field effect transistors) and IGBTs (Insulated gate
bipolar transistors) are can conduct tens to hundreds of amperes of current and sustain source-drain
(MOSFET) or emitter-collector (IGBT) voltages of hundreds to thousands of volts. Because of this,
switched-mode power supplies (SMPS) can be built that convert kW of power from AC to DC, and
with an efficient power factor. This opens up much opportunity to the hobbyists who wishes to exploit
SMPS for experimentation, e.g. for high voltage experiments, Tesla coils, fusion devices, welding,
induction heating, etc. This document summarizes some of the issues in constructing a simple SMPS
based on a half-bridge topology, including the half-bridge, gate drive, and transformer construction.
Most of this discussion also can be applied to building a full-bridge SMPS as well.
Mains frequency transformers (50 ot 60 Hz) require very large cores to achieve sufficiently high
inductance at this low frequency. The reactance of the secondary must be comparable or greater than
that of the load, and as a result the core is typically a very large “E” shape with high magnetic
permeability soft iron core laminations. A SMPS is able to use a much smaller transformer because the
SMPS operates in the 10 to 300 kHz range of frequencies, and because the reactance of the windings
scale with frequency, achieves a comparable reactance with a smaller core. As the eddy current and
hysteresis losses of soft iron would be too great that the higher frequencies, typically ferrite core
transformers are used, which are much less conductive and therefore do not conduct significant eddy
currents, so that hysteresis loss is the major contributor to loss in the ferrite core.
Because SMPS transformers are lighter and less expensive than mains frequency transformers, SMPS
are replacing mains frequency transformers in most applications, especially those requiring high power.
Unfortunately, this means that the mains frequency transformers that are currently used in high voltage
experimentation, such as neon sign transformers (NST) and microwave oven transformers (MOT), are
going to become increasingly difficult to find. This document was written to help the high voltage
hobbyist transition to the same SMPS technology that is becoming dominant for other power
electronics applications. The very popular Mazzilli/Royer flyback driver is used widely for this
purpose, however, it has a problem directly using mains voltages, and has not scaled to over 1 kW of
power. Furthermore, as LCD televisions are becoming dominant, the CRT flyback transformers
typically used with the Mazzilli driver will become increasingly scarce as well. The half bridge
configuration, which is already used in high current welders and solid state Tesla coils, is a good
starting point for those who wish to work with a directly mains-powered high voltage source.
Furthermore, winding a flyback transformer, while tedious, is not out of reach of the hobbyist,
especially if large enough ferrite cores can be obtained.
The high voltage SMPS in this document is unregulated, and is principally designed for charging
capacitors for Tesla coils, Jacob's ladders, or other high voltage experimentation. It is current-limited
by an inductor, not by duty cycle or frequency modulation like a conventional SMPS would be. It is
designed to be as simple to construct as possible but be usable for the same sorts of applications that a
MOT and NST would be used for. This means that the same sorts of safeguards that would ordinarily
be on a commercial SMPS are not built into the SMPS in this document. There is a primary-side
overcurrent detector, however, the secondary side current is limited by an inductor on the primary side,
which must be designed to hold the current within limits even if the secondary windings are shorted.
The voltage of the secondary can be very high if the secondary windings are open. The making and
breaking of arcs is very hard on a power supply and can potentially cause voltage and current spikes
that cause the transistors to exceed their safe operating limits, causing them to short out and be
destroyed, even with snubbers and protection circuitry. Therefore, again, the experimenter is urged
to pay attention to the potentially high voltages and currents involved in this SMPS, and to take
precautions as any high voltage, non-current limited source would require. One shock can kill.
The half bridge
The half bridge is an attractive topology for building a switched-mode power supply. It uses a pair of
transistors, one of which connects one side of the load to the positive supply, and the other the negative.
The load is switched alternately between the maximum and minimum voltage by switching on one or
the other transistor, but never both at the same time. The other side of the load is held at a nearly
constant voltage that is nominally at half the supply voltage. This side is connected to a capacitor
voltage divider between the positive and negative supply, and the capacitors also act to block DC
current flow should the half bridge stop oscillating. The inductance of the transformer primary and any
additional inductors and the capacitors for a series resonant LC circuit. The half bridge is typically
operated at a frequency well above the resonance frequency of the LC circuit so that the inductance
limits the current in the load.
+V
Load
Gate drive signal
Current
limiting
inductor
+V
Transformer
Each transistor alternately
switches one side of the load
to +V and ground
Split capacitors hold the other
side of the load at a V/2
nominal voltage.
Basic half-bridge circuit
A full bridge is similar to a half-bridge, except that both sides of the load are connected to transistor
pairs, and both sides of the load are alternately switched between the positive and negative supplies,
with one side being connected to positive when the other is connected to negative, and vice versa.
Because of this, a full bridge can deliver a voltage equal to the supply voltage across the load, while a
half bridge only delivers a voltage equal to half the supply voltage. As a result, a full bridge can deliver
double the power to the load for the same transistor current. The primary disadvantage of the full
bridge is that it requires four transistors rather than two.
The Royer or push-pull topology is popular because the Mazzili variant is able to self-oscillate,
achieving zero-voltage-switching (ZVS) and therefore low switching losses. Its primary disadvantage
is that if oscillation should cease, the load, which is a parallel LC resonator rather than a series circuit,
presents a short-circuit. The series inductor with the load, which limits current while oscillating, is a
short circuit when oscillation ceases. Therefore protection is required to shut the circuit down if a short
circuit occurs and current levels are exceeded. Another disadvantage is that as each of the transistors
delivers power through a separate winding, the voltage produced by the magnetization of the
transformer works against the other transistor, so that the voltage stresses on each transistor is doubled.
Furthermore, volt-second imbalance can occur if the two transistors are on for unequal times, saturating
the core. Nevertheless, it has proved an attractive transformer driver for under 1 kW loads because it is
easy to build and works reliably using transistors such as the IRFP260.
A flyback converter, while easy to build, produces large stresses on the switching element, and is low
efficiency. Furthermore, it magnetizes the transformer in only one direction, so that the total flux
excursion is half of the Royer, half, or full bridge, so it does not deliver as much power per cycle.
Therefore it is usually unsuitable for power supplies delivering more than 200 W because of the high
power dissipation in the switching element.
The two-switch forward converter has some advantages over the half-bridge, including no possibility of
shoot-through, less voltage stress on the switches, resetting of the flux in the transformer each cycle if
the duty cycle is less than 50%, and better recycling of the energy stored in the transformer. However,
it delivers power on up to only one half of the cycle and magnetizes the core only one way rather than
using the full flux deviation of the core, so that a more substantial transformer is required to deliver the
same amount of power as a half bridge. Nevertheless, these are advantages that make it a viable
alternative to a half bridge in many cases.
Designing the half bridge and the transformer
The essential features of the transformer can be designed using a few equations. Generally, a toroidal
ferrite formed from two U-shaped pieces is used, made of a material such as Ferroxcube 3C85 or 3C90,
Epcos N85, N95, or N96, or many others that will work. Surplus Sales of Nebraska
(www.surplussales.com) often has inexpensive surplus ferrite cores for transformers, from which the
author obtained ferrite cores. Information Unlimited (www.amazing1.com) also has prebuilt flyback
transformers or ferrite cores. Television flyback transformers are also available from many sources
including Goldmine Electronics (www.goldmine-elec.com) which may be useful to experiment with,
however, the diodes in most television flyback transformers fail after conducting more than 300 W of
power, as these were never intended for high power applications, and so the current limiting inductance
should be designed for the lower power. The important feature is that the ferrite core must have a large
enough cross-section for the desired power, and support switching frequencies up to 200 kHz.
The toroid is split into two U-shaped halves, so that the secondary windings can be wound onto a
bobbin, and then the toroid pieces slipped into the bobbin and joined into a magnetic circuit. Use a Ushaped toroid split into two halves with straight sections so that the bobbin that can be inserted between
the two halves when ready.
To start, you must know:
·
·
·
·
·
·
·
·
The cross-sectional area A in m2 of the straight sections of the ferrite toroid that are slipped
inside the bobbin.
The length of the magnetic circuit Le which is the distance around the perimeter of the toroid.
The saturation field Bsat of the ferrite (usually 0.2 Tesla is a good guess).
The relative permeability me of the core material at the onset of saturation, usually 1500 is a
good guess for many ferrite materials.
The minimum working frequency of the switching converter f.
The rectified voltage VR that is applied to the toroid.
The desired secondary voltage VS.
The desired power P.
If using rectified mains without a voltage doubler , the rectified voltage VR can be determined from the
mains voltage by VM.
The
factor coverts RMS mains voltage to the peak voltage, and the half factor accounts for the fact
that a half-bridge divides the source voltage in half. If a voltage doubler is used on the mains voltage,
then use
. To estimate the minimum number of turns on the primary NP using Faraday's
law before the saturation magnetization is reached:
so that
and volts per turn is
.
The number of secondary turns NS is given by the ratio
The peak current to achieve the power P is given by
To calculate the needed inductance of the primary LP (with the secondary open circuit)
The inductance of the transformer may be reduced by increasing the gap size between the two halves.
If on the other hand the inductance is too small to limit the current, additional primary windings need to
be added, with a corresponding increase in the number of secondary turns. Alternatively, the frequency
of oscillation may be increased to maintain the peak field in the core to within saturation limits. As the
frequency is increased, the amount of power dissipated by the power transistors (IGBT/MOSFET)
increases, and the hysteresis loss of the ferrite increases. Therefore this approach can be used only a
limited amount. Many ferrite transformer materials are practically limited to 200 kHz operating
frequency such as Ferroxcube 3C85 and 3C90. Older materials such as 3C85 also have higher
hysteresis losses. The hysteresis losses heat up the ferrite core, and if the ferrite core reaches the Curie
temperature, typically between 200ºC and 300ºC, the ferrite loses its ability to be magnetized, so the
ferrite core must be maintained well below this temperature, usually below 120ºC. Materials such as
Ferroxcube 3F3/3F4/3F5 can be used at a higher frequency, but cost significantly more. In general, the
range of operating frequencies is between 20 to 100 kHz.
Load voltage
Load current
Magnetic field
Bsat VR
IP
0
1/2f
1/f
time
-IP
-Bsat -VR
Idealized graph of the voltage, current, and magnetic field during a half bridge cycle
The above calculations assume that there is perfect coupling between the primary and the secondary,
that is, all of the flux lines in the primary winding also circulate through the secondary winding.
However, often there is significant leakage of flux lines in a high voltage transformer. In a low voltage
transformer, the primary and secondary coils are concentric so that there is the maximum coupling
between the two. However, a high voltage coil requires strong galvanic isolation between the primary
and the secondary wires so that there is no arcing between them, and the core itself which is slightly
conductive. Therefore the primary and secondary are wound on opposite sides of the core with a
significant gap between them. Unfortunately this results in significant flux leakage.
The flux leakage results in leakage inductance. This is effectively an extra inductor in series with the
primary coil of the transformer. Part of the voltage drop across the primary coil occurs at the leakage
inductance, which limits the voltage and current at the part of the primary coil that actually couples
field lines and therefore current to the secondary side of the coil. An ideal transformer without leakage
inductance would have no primary coil inductance when its secondary is shorted, however, in a real
transformer an inductance is still present if the secondary is shorted, which can be measured for
example by using a LCR meter, as opposed to the total of leakage plus coupling inductance, which is
measured with the secondary windings open circuit. Some leakage inductance can be helpful to limit
the current, however, excessive leakage inductance prevents the desired current from being coupled to
the secondary winding. The total inductance that limits the current when the secondary is shorted
circuited is the leakage inductance of the transformer plus any additional inductance added in series
with the transformer primary winding. More secondary windings need to be added to compensate for
the flux loss due to the leakage inductance as well.
The two halves of the transformer core are typically U-shaped as shown below. The gap is in between
the two core halves. The gap width may be adjusted by inserting strips of electrical tape or other
insulating, dimensionally stable tape between them. Increasing the gap width decreases the inductance
of the transformer primary and therefore increases the current in the load. The halves should be
clamped together tightly, but not so tightly that the brittle ferrite material cracks. Placing soft pads
between the clamping surface and the ferrite helps avoid cracking. An estimate of the optimal gap size
that prevents core saturation but minimizes the number of secondary turns needed is given by:
with the magnetomotive force in ampere turns given by:
and the effective permeability
of the core adding the gap is
Because of the skin effect, heavy duty wire should be used for the primary. Litz wire is ideal as it
minimizes the skin effect, however, large gauge wire, multiple wound smaller strands of wire, or ribbon
wire can suffice.
Core halves
Gap in core (not to scale)
usually less than 1.0 mm thick.
Primary wound on one side of core
The dotted line is the length of the High voltage secondary wound on bobbin
on the secondary side of the core
magnetic circuit Le.
Diagram of the two halves of the core and how they fit together.
Winding the secondary, high-voltage side of the core must be done carefully as the voltage can arc
between the wires. Typical windings for high voltage are designed in sections so that the stress of the
high voltage is distributed between the sections, either laterally and/or radially, as shown in the
following diagrams.
First a plastic insulating tube is chosen for the bobbin, for example of PVC, ABS, or PMMA (plexiglas
or lucite) materials. This tube should conform as closely as possible to the cross-section of the core, so
that there is minimum flux leakage. Tubes can be purchased from hobby stores or Small Parts, Inc., or
they can be designed and 3-D printed in ABS for the geometry needed. If water is used to wash
support material away from the 3-D printed part, make sure that the part has been allowed to dry
thoroughly before winding! The bobbin wall should be thick enough to prevent arcing of the voltage to
the ferrite core, which may require wrapping extra dielectric layers around the bobbin.
There are two common methods of winding high voltage transformers: side-by-side coils, and layered
coils. Side-by-side coils can be used if there is more space available laterally then radially in the
transformer core window. However, it uses space laterally for insulation between the sections.
Layered coils can be used if there more space available radially than laterally. However, thick layers of
insulation are required between the winding layers to prevent arcing between them. The two diagrams
below show the two methods. Kinks in wires and points on solder joints should be avoided as these
tend to be source of corona discharge which breaks down the dielectric. Depending on the voltage and
frequency, the bobbin may need to be potted in epoxy or silicone and/or immersed in mineral oil or
transformer oil when operating to prevent corona discharge from destroying the transformer. The
benefit of mineral oil or transformer oil is that trapped air can be more easily excluded from the
transformer, and that the dielectric is “self healing” to a degree. This is probably the easiest and best
alternative available to the hobbyist. Beware of buying or recycling old transformer oil, as this may
contain polychlorinated biphenyls (PCBs) which are toxic and an environmental hazard.
An alternative is solid encapsulation. Paraffin has been used for solid encapsulation instead of oils,
however, it melts at a rather low temperature which is a significant disadvantage. Clear two-part epoxy
encapsulant may be purchased as wood varnish in quart/liter quantities, for example as Envirotex Lite
which is available at craft stores, or in larger quantities online. Sand, fumed silica, quartz, or mica filler
can be mixed with the epoxy when potting the transformer to help reduce the amount of epoxy needed.
White or clear silicone caulk that is not acetic acid based (acetic acid based silicone smells like
vinegar) can also be used as a dielectric, especially to seal around wires. Vacuum impregnation of the
windings is needed for best results but often not available to the hobbyist. Vacuum impregnation of the
windings removes trapped air that causes an effect called “partial discharges” where dielectric
breakdown occurs at the dielectric-air interface. If vacuum impregnation is not available, agitation of
the part and/or smoothing the polymer into the windings to push out the air can help avoid trapped air
bubbles. Lower viscosity epoxy or silicone is often used to facilitiate better penetration of the windings
with dielectric. To prevent the epoxy or silicone from curing inside the bobbin tube, an insert can be
fashioned from expanded polystyrene (which can be obtained from packing materials) and placed into
the tube while potting, which is easy to crumble away after the epoxy sets.
If turns are layered on top of each other, the number of volts per turn should be calculated. Depending
on the insulation used, there should be no more than approximately 100 to 150 V between any two
turns in contact. If the core cross-section is large or the frequency is high, and therefore the number of
volts per turn is large, only a few turns across can be used. For example, if 30 AWG wire is used,
which is approximately 0.25 mm in diameter, and the transformer is 10 V/turn, then the permissible
width of the section is (100V/10V)*0.25 mm=2.5 mm or approximately 1/10 inch.
Corona is a discharge effect which slowly deteriorates the dielectric, and produces a purple glow in air.
The amount of corona is determined by the ratio between the spacing between the wires and their
diameter. For long, parallel wires, corona does not form if the ratio between the wire separation and
the wire diameter is kept below 2.92. Therefore maintaining a close spacing of the wires prevents
corona from forming. However, dielectric breakdown is still possible for a sufficiently high voltage.
Current handling capability and diameters of copper wire for typical secondary wire gauges are:
Gauge
24 AWG
26 AWG
28 AWG
30 AWG
32 AWG
34 AWG
36 AWG
38 AWG
40 AWG
Current Capacity
0.577 A
0.361 A
0.226 A
0.142 A
0.091 A
0.056 A
0.035 A
0.023 A
0.014 A
Diameter
0.51 mm
0.40 mm
0.32 mm
0.25 mm
0.20 mm
0.16 mm
0.13 mm
0.10 mm
0.078 mm
Resistance
0.084 Ohms/m
0.133 Ohms/m
0.213 Ohms/m
0.338 Ohms/m
0.538 Ohms/m
0.856 Ohms/m
1.360 Ohms/m
2.163 Ohms/m
3.440 Ohms/m
Skin Depth Frequency
68 kHz
107 kHz
170 kHz
270 kHz
430 kHz
690 kHz
1100 kHz
1750 kHz
2900 kHz
Strong dielectric materials include polyethylene, polyester (Mylar), and PTFE sheets. PVC has a
somewhat lower dielectric strength, but some grades are also useful. The high frequency AC
breakdown field is typically far less than the DC breakdown field. The DC breakdown field
measurement does not account for the stresses of the dielectric under the strain of repeated charging
and discharging, especially with the direction of the electric field reversing each cycle. The DC
breakdown field is as high as 500 V/mil or 20 kV/mm for polyethylene or PTFE, however, it is much
reduced at tens to hundreds of kHz. A safety factor also must be included as sparks or other sudden
discharges outside the transformer can cause temporary voltage spikes inside the secondary windings
that can exceed the dielectric strength of the layer. Six mil, or 0.15 mm thick low density polyethylene
sheets may be obtained in large rolls as undyed, translucent painters drop-cloth or tarp. However, these
typically have small holes or defects so many layers should be used together. Thick sheets of PVC are
available cheaply as shower pan liner, shower curtains, or mattress covers or bags. The ideal material
is nonmetallized, undyed Mylar sheet, which is often used in transformers. A good safety factor should
be to use an amount of insulation that produces three times the dielectric strength needed for DC
breakdown.
The sections must be narrow so that turns with different voltages greater
than 100 volts in difference do not come in contact. This limits sections to
2 to 6 mm in width.
Bobbin tube
Insulating
separators
High voltage wires soldered
onto magnet wire. Avoid
sharp solder points!
Notch cut in
separator to allow
wire to pass
through separator
on the surface of
the tube
Side-by-side coils high voltage bobbin. The bobbin is divided into sections by using insulating
separators, and the wire wound onto each section before proceeding to the next, so that the voltage
across any section is limited. A notch is cut into the section to allow the wire to cross to the next
section. The position of the notch alternates between sides of the bobbin. A nylon or polyester thread
can be tied over the wire in each section as it is completed, and at the bottom of each section as it is
started, to ensure the wire stays in place and does not unravel or contact turns unintentionally.
Alternatively, cyanoacrylate glue (super glue) may be used to glue the winds together which sets
quickly, holds them in place during winding and prevents unraveling. The width of each section should
be sufficient narrow so that turns with very different voltages do not come in contact with each other.
It is very important not to run the wire from one section down the windings from the next section,
otherwise arcing will occur between the wires of different potentials. The separator should be wide
enough to allow the wire to descend from the top of the last winding to the bottom of the next without
touching the windings of either the last or next section.
High voltage wires soldered
onto magnet wire on
opposite sides of the tube
Extra thick dielectric
separator at outside
Wound layer
Thick dielectric separator
Wire crosses from the end
of the last layer to the
beginning of the next layer
through a short slit cut into
the end of the separator
Bobbin tube
Thick dielectric separator
Wound layer
Layered high voltage bobbin. A coil of wire is wrapped around the bobbin tube for each layer. At the
end of each layer, a thick layer of dielectric is placed, enough to stand at least three times the potential
difference expected between the layers. The wire is then passed back along the bobbin tube to the
beginning of the next layer, and another layer of dielectric is placed on the wire, and the pattern repeats
for as many layers as needed. The layers may need to be several millimeters thick depending on the
voltages between the layers.
Two bobbins wound and
wired in series to achieve a
higher voltage.
Primary coil
Because the primary and
secondary are in closer
proximity, use sheets of
insulation around the
primary coil as well the
secondary.
Two bobbin or multibobbin separation. Multiple secondary bobbins can be placed on the core and used
separately, or wired in series to achieve a higher voltage. The windings on the two bobbins must be in
the same direction or the voltages will cancel, or the polarity of the connection on one of the bobbins
should be reversed. Multiple bobbins can also reduce leakage inductance.
Saturation of the transformer core
When a constant voltage V is applied to the inductor for a time T, the current in the inductor is I=LVT.
Because an inductor is really just a solenoid, we can calculate the magnetic field in the inductor from
, where N is the number of turns of wire, I is the current in the wire, l is the length of the
solenoid, and m is the magnetic permeability of the material inside the solenoid. Therefore put
together:
Therefore if a constant voltage is applied to the inductor, the current and the magnetic field rises in the
inductor rises linearly in time. Eventually, the magnetic field reaches a point known as the saturation
field. The permeability of soft magnetic materials depends on both the temperature and the applied
magnetic field. When the magnetic field in the material becomes large enough, the magnetic
permeability drops precipitously because the material can not be magnetized any further; it has reached
the limit of the magnetic field strength it can produce. Because of this, the inductance of the solenoid,
which depends on its permeability, drops as well until the magnetic field in the core falls below the
saturation field. For most soft ferrite materials used in transformers, this saturation field is 0.3 T to 0.4
T, so it is important that this magnetic field strength is not exceeded.
The effect of the sudden drop in the inductance of solenoid when saturation occurs is to allow a large
surge of current to flow through the inductor, as the inductance which previously restricted the flow of
current has been removed. This further drives the material into saturation, so that the current is likely
to be ultimately limited by some other undesirable mechanism, e.g. blowing a fuse, burning a transistor,
etc.
As the magnetic field is concentrated when the cross-sectional area of the core is small, the cross
sectional area of the transformer should be large enough to ensure that for the current required, the
magnetic field does not approach or exceed the saturation magnetization. This is why larger
transformer cores are needed for a higher power SMPS.
Another way that transformers may be driven into saturation is called “volt second” imbalance. The
polarity of the applied voltage is reversed periodically across the primary of the transformer. For one
half of the cycle, if a voltage of V1 is applied for a time T1, this means a current of I=LT1V1 flows
through the primary coil. If for the other half of the cycle, a voltage of V2 is applied for a time T2, the
opposite current is forced through the coil, so that the net change in current for an entire cycle is
I=LT1V1 -LT2V2. If the product T1V1 =T2V2 does not match, each cycle the current increases in the
primary by the imbalance LT1V1 -LT2V2. Eventually, the current builds up to a level that produces a
sufficiently strong field in the inductor to cause it to saturate. Therefore it is important to ensure that
no direct current can build up in the primary coil. This is typically achieved by placing a DC blocking
capacitor in series with the primary coil. It must be small enough so that the current does not
accumulate for the transformer to enter saturation, but large enough so that the voltage drop is mostly
across the inductor. In a half bridge, the series capacitor, which is on the order of 0.1 to 10 mF, serves
to block the DC component of the current, preventing volt-section imbalance. However, the series
capacitor does not completely eliminate the possibility of saturation, as detailed in the “staircase
saturation” section 6.1 and 6.2 of Switchmode Power Supply Handbook by K. Billings and T. Morey,
but their solution requires an active flux balancing technique which is beyond the scope of this design
or document.
Designing the current limiting inductor
The current limiting inductor is needed if the leakage inductance of the transformer is insufficient to
limit the current. Some leakage inductance in the transformer can be helpful to limit current, however,
excessive leakage inductance limits current and does not allow sufficient power to be transmitted to the
secondary.
If the power supply is used to charge capacitors, for example in a Tesla coil primary circuit or a Marx
generator, the capacitor appears as a short circuit when its begins to charge. Also, if there is arcing of
the high voltage (whether intentionally or unintentionally), or a short circuit occurs, without a current
limiting inductor there is the risk that the overcurrent circuit does not trip and the current rating of the
MOSFETs/IGBTs are exceeded, destroying them. Therefore it is wise to include a current limiting
inductor.
If a ferrite core is used for the inductor, the minimum number of turns required to not saturate the core
of the inductor again is given by
where A is the cross-sectional area of the inductor core and Bsat is the saturation field. Generally, more
turns than this are needed. The desired inductance needed to limit the current is given by
where IL is the desired current limit, which normally would equal IP. If you do not know the
inductance, in general it is wise to add many more winds than necessary to increase the inductance, and
remove winds from the current limiting inductor as needed. Because the inductance is proportional to
the square of the number of winds, as the number of winds becomes small, the inductance decreases
sharply. Therefore when the number of winds decreases below 12 or so, remove them one at a time.
Often inductor cores specify a quantity called AL, which is the inductance produced by 100 turns on the
core. To figure out the inductance of the core in with an arbitrary number of turns N:
Some iron powder cores may be unsuitable depending on the switching frequency used, so ferrites are
generally preferred. Too high a density of winds may produce undesired interwinding capacitance.
Another alternative is to use a large air-core solenoid. The benefit is that air does not saturate. For a
flat, multiturn circular coil
where r is the radius of the coil, N is the number of turns, and a is the radius of the wire. For a long,
thin air solenoid
where l is the length of the solenoid. An air solenoid can be formed by winding heavy gauge wire
around a PVC pipe.
Maximum current and power
Given that the peak current is
, make sure that your MOSFET/IGBTs are rated for this
current. In particular the lower the mains voltage, the higher the peak current is and therefore the larger
the power dissipation is in the transistors to achieve a particular power output. Furthermore, the power
rating of the MOSFETs/IGBTs should not be exceeded, as even with a heat sink the package of the
device has a limited thermal conductivity. MOSFETs are rated by an on resistance
, usually from
0.050 W to 0.500 W so that each of the MOSFETs dissipate an amount of power neglecting switching
power of
. IGBTs have a collector-emitter forward voltage drop
2.5 V, so that the power dissipated is
on the order of 1.5 V to
. For larger amounts of current, IGBTs dissipate
significantly less power than MOSFETs as the power scales linearly rather than as the square of current
with MOSFETs. However, IGBTs switch more slowly than MOSFETs, especially when these are
turning off.
There is an additional power dissipiated during the switching from off to on and vice-versa, as the
switch transitions from a low current, high voltage drop during its off state which dissipates little
power, and a high current, low voltage drop state which dissipates relatively little power. During the
transition, there is a significant voltage drop across the transistor as well as current, which is increasing
or decreasing during the transition. This transition time takes on the order of nanoseconds to
microseconds and depends on the inductance of the load connected to the transistor. A higher
switching frequency results in a larger amount of the power being dissipated during switching which
also must be factored into the heat dissipated by the transistors. A gate drive signal that switches
between high and low voltages as quickly as possible is needed to ensure that the amount of time the
transistor spends in its transition is minimized. As the capacitance of MOSFETs and IGBTs gates is
significant, the gate driver may have to deliver hundreds of mA to many amperes briefly during a
transition. A zero-voltage-switching (ZVS) gate driver times the switch transition so that the transistors
are switched when the voltage across the transistor is zero, minimizing the switching power, but is
more complicated to implement than the hard switching half bridge in this document.
The open and short circuit reactive average power and peak current are given by
Resonance and safe frequency operation
At the resonance frequency, the tank circuit formed by the inductors and capacitors presents a zero ohm
reactive load, shorting out the half bridge and destroying the transistors. While some designs operate
near resonance and change the frequency to regulate the delivery of power to the load, for a simple half
bridge it is safer to operate at a frequency far above the resonant frequency so that the current limiting
inductor limits the current. The capacitance in series with the load should be high enough to lower the
resonance frequency well below the operating frequency. The duty cycle of the half bridge may be
varied and the frequency held constant to regulate the load, with the maximum power delivered at 50%
duty cycle.
If the total capacitance in series with the inductors is C, then the resonance frequencies to avoid are
given by
Half-bridge driver circuits
There are two circuits included as half-bridge drivers. The first is based on the ubiquitous and
inexpensive IR2153D half-bridge driver IC. This provides low and high side drive for the
MOSFETs/IGBTs, and also oscillates at 50% duty cycle. Conveniently, it also has a built in zener
diode to regulate the gate drive voltage, which can also be used (with low current < 10 mA) for other
circuits. With the appropriate series resistor, the zener diode allows the IR2153D to be operated
directly from mains, obviating the need for a separate power supply. The primary disadvantage it is
that is not galvanically isolated from the half-bridge power supply. If the MOSFET/IGBT does short
out or blow up, it takes the IR2153D and any other silicon (and sometimes other passive components)
with it, often violently. This happens within milliseconds, so fuses or other circuit breakers do not react
fast enough to prevent this failure.
The second circuit is based on the UCC37321/UCC37322 gate driver ICs. These are used to drive a
gate drive transformer, which galvanically isolates the half bridge from the driver circuit. The gate
drive transformer is 1:1:1 and wound as a trifilar for best flux coupling, so that the drive pulse is
transmitted undistorted. One winding is connected to the UCC37321 and UCC37322 gate drive
outputs to alternate the voltage difference, and the other two windings are connected to each of the
gates of the transistors. The two windings must be connected with opposite polarities to each of the
gates of the transistors, so that they turn on alternately (rather than at the same time and produce shoot
through!). A LM339 configured as an oscillator produces the timing signal, however, this is not 50%
duty cycle. The output is then divided in half in frequency by a J-K flip flop to generate the 50% duty
cycle signal.
Each of these circuits is equipped with an overcurrent detector. The overcurrent detector uses a current
sense transformer. This transformer consists of a primary winding which is simply the load wire
wrapped one turn around the toroid, and a 50 to 100 turn secondary winding. A current sense load
placed on the secondary winding is effectively reduced in impedance by the square of the turn ratio on
the primary winding side. Therefore it allows a small sense load to be placed in series with the actual
load, while maintaining galvanic isolation using the transformer. For example, a 10 W load on the
secondary side with a 1:50 turn ratio appears to be 10 W/(50×50) or 0.004 W on the primary side. A full
wave bridge rectifier is used to turn either a positive or negative current into a positive voltage on the
secondary side. The voltage on the secondary is equal to the current in the primary times the secondary
load resistance divided by the turns ratio, minus the diode drop losses. Therefore, minus the diode drop
losses, for example 10 A of primary current with a 1:50 turn ratio and a 10 W sense load would produce
2 V on the sense load, which including the diode drop is near 1 V. There is a potentiometer to adjust
the threshold at which the current limiting occurs. Note: the overcurrent detector does not detect
shoot-through, only excessive load current. The overcurrent detector also doubles as a supply
undervoltage detector, which shuts off oscillation if the low voltage DC supply voltage falls below 12
volts.
The current limiting circuit consists of two comparators, the first of which compares the current sense
voltage to a fixed level. The open collector comparator output suddenly discharges a capacitor when
the current sense voltage threshold is exceeded. The voltage on this capacitor is sensed on a second
comparator, which if the voltage on the capacitor falls below half the supply voltage, pulls the output of
the second comparator low. This second comparator output is connected to an enable line on the gate
drives of the IR2153D or UCC37321/UCC37322 to shut off the gates and therefore both of the
transistors. A 1 MW resistor slowly recharges the capacitor and allows the oscillation to resume after
approximately 1/10 s, so that the circuit does not need to be power cycled to resume operation.
However, if the cause of the overcurrent condition is not remedied, the overcurrent condition will
reoccur and the cycle of current shutdown and restart will continue.
It is crucial to keep the wires short and fat between the collector and emitter of the IGBTs, the power
buses, and the load. An increased inductance of these wires increase the transient voltage spikes that
occur at the emitters and collectors. The snubber capacitor and resistor combination connected to the
collector and emitter of each IGBT should also be as close as possible as well. There are additional
capacitors between the positive and negative supply at the IGBTs to stabilize the voltage at the IGBT.
Snubber capacitors should be polypropylene, MKP preferred. The capacitor and resistor may be
replaced with several in parallel to reduce the inductance of the snubber. The IGBTs were inserted into
screw terminal sockets to make them easier to replace. The two IGBTs were mounted on a 3-by-3 inch
aluminum plate heatsink with a silicone pad between the IGBTs and the plate to isolate the collector
from the heatsink. A fan is blown on the plate to cool it.
Schottky diodes are shown in parallel with the gate drive resistors in these circuits. This is to reduce
the time required to turn the transistor off. These diodes may not be necessary, but adding the diodes
reduces the possibility of shoot through. The diodes should be rated for at least 40 V and 1 A.
The circuits are shown with an optional voltage doubler. Two capacitors are needed if a voltage
doubler is used, each rated for the mains voltage (plus a 2 times safety margin) of the supply. Without
voltage doubling a single capacitor may be used, rated for the mains voltage (plus a 2 times safety
margin) and the needed capacitance. A voltage doubler should not be used with a 230 V AC mains
supply, unless the IGBTs are rated for 1200 V operation. The IR2153D is not designed to operate with
a 1200 V half-bridge voltage, so a gate drive transformer should be used. It is often advantageous to
use a voltage doubler with 120 V mains as it reduces the current required to be switched by the IGBTs.
These circuits were designed to use a single +15 V DC, 2 A voltage supply as to minimize part count.
The IR2153D circuit may be operated using its internal Zener voltage source as well. A switching
power supply wall adapter may be used as long as its isolated.
Other possibilities
To mention some of the other potential refinements:
·
Using a pulse width modulated SMPS controller such as the TL494 to control the output
voltage, or alternatively, design the circuit as a frequency-modulated LLC resonant converter.
·
Adding output overvoltage protection or supply undervoltage protection.
·
Compare the design to a two-switch forward converter.
Example design spreadsheet: 120 VAC, no voltage doubling, 30 A maximum IGBT current, 1200 W
Half Bridge Design Spreadsheet
Units
Mains Supply Voltage
Mains Frequency
Design Power
Design Frequency
120 V RMS
60 Hz
1200 W
25000 Hz
Transformer Core Sat. Field
Cross-Sectional Area of Core
Desired Secondary Voltage
0.2 Tesla
0.0008165306 m2
20000 VAC
2000 Gauss
8.16530625 cm2
Current Limit Core Sat. Field
Cross-Sectional Area of Core
0.2 Tesla
0.0008165306 m2
2000 Gauss
8.16530625 cm2
Tank capacitor
Number of Tank Capacitors
0.0000041 F
4
Half Bridge +/- Voltage
Peak Current
Supply Capacitance
Number of Primary Turns
Volts per Turn
Primary Inductance
Primary Inductance Check
Primary Reactance
Number of Secondary Turns
84.8528137424 V peak
28.2842712475 A peak
0.0008841941 F
5.1959357766 turns
16.3306125 V/turn
0.00003 H
0.00003 H
4.7123889804 Ohms
1224.6938074123
Number of Inductor Turns Minimum
Inductance
Inductance Check
Reactance
5.1959357766 turns
0.00003 H
0.00003 H
4.7123889804 Ohms
Total Tank Capacitance
0.0000164 F
Resonance Frequency of Tank
5073.6742742157 Hz
Resonance Frequency No Load 7175.2589696593 Hz
Reactive Power No Load
600 VA
Reactive Peak Current No Load
14.1421356237 A
Reactive Power Short Circuit
1200 VA
Reactive Peak Current Short Circui 28.2842712475 A
4.1 uF
884.1941282883 uF
30 uH
30 uH
30 uH
30 uH
16.4 uF
Example design spreadsheet: 120 VAC, voltage doubling, 25 A maximum IGBT current, 1800 W
Cross-Sectional Area of Core
Mains Supply Voltage
Mains Frequency
Design Power
Design Frequency
Units
120 V RMS
60 Hz
1800 W
25000 Hz
Transformer Core Sat. Field
Cross-Sectional Area of Core
Desired Secondary Voltage
0.2 Tesla
0.0008165306 m2
20000 VAC
2000 Gauss
8.16530625 cm2
Current Limit Core Sat. Field
Cross-Sectional Area of Core
0.2 Tesla
0.0008165306 m2
2000 Gauss
8.16530625 cm2
Tank capacitor
Number of Tank Capacitors
Half Bridge +/- Voltage
Peak Current
Supply Capacitance
Number of Primary Turns
Volts per Turn
Primary Inductance
Primary Inductance Check
Primary Reactance
Number of Secondary Turns
0.0000041 F
4
169.7056274848 V peak
21.2132034356 A peak
0.0003315728 F
4.1 uF
331.5727981081 uF
10.3918715532 turns
16.3306125 V/turn
0.00008 H
0.00008 H
12.5663706144 Ohms
1224.6938074123
80 uH
80 uH
Number of Inductor Turns Minimum 10.3918715532 turns
Inductance
0.00008 H
Inductance Check
0.00008 H
Reactance
12.5663706144 Ohms
80 uH
80 uH
Total Tank Capacitance
0.0000164 F
Resonance Frequency of Tank
3106.9782732285 Hz
Resonance Frequency No Load 4393.9308119983 Hz
Reactive Power No Load
900 VA
Reactive Peak Current No Load
10.6066017178 A
Reactive Power Short Circuit
1800 VA
Reactive Peak Current Short Circui 21.2132034356 A
16.4 uF
Example design spreadsheet: 230 V AC, 50 Hz, no voltage doubling,
30 A maximum IGBT current, 2400 W
Half Bridge Design Spreadsheet
Mains Supply Voltage
Mains Frequency
Design Power
Design Frequency
Units
230 V RMS
50 Hz
2400 W
25000 Hz
Transformer Core Sat. Field
Cross-Sectional Area of Core
Desired Secondary Voltage
0.2 Tesla
0.000816531 m2
20000 VAC
2000 Gauss
8.16530625 cm2
Current Limit Core Sat. Field
Cross-Sectional Area of Core
0.2 Tesla
0.000816531 m2
2000 Gauss
8.16530625 cm2
Tank capacitor
Number of Tank Capacitors
0.0000041 F
4
4.1 uF
Half Bridge +/- Voltage
Peak Current
Supply Capacitance
162.6345597 V peak
29.51402217 A peak
0.000577651 F
Number of Primary Turns
Volts per Turn
Primary Inductance
Primary Inductance Check
Primary Reactance
Number of Secondary Turns
9.958876905 turns
16.3306125 V/turn
5.51042E-05 H
5.51042E-05 H
8.655742259 Ohms
1224.693807
55.10416667 uH
55.10416667 uH
Number of Inductor Turns
Inductance
Inductance Check
Reactance
9.958876905 turns
5.51042E-05 H
5.51042E-05 H
8.655742259 Ohms
55.10416667 uH
55.10416667 uH
Total Tank Capacitance
Resonance Frequency of Tank
Resonance Frequency No Load
Reactive Power No Load
Reactive Peak Current No Load
Reactive Power Short Circuit
Reactive Peak Current Short Circuit
0.0000164 F
3743.613375 Hz
5294.268808 Hz
1200 VA
14.75701109 A
2400 VA
29.51402217 A
577.6512112 uF
16.4 uF
IR2153D Half Bridge Driver
+
mains
250V, 25 A
slow blow
fuse
Double
pole,
double
throw
switch
Inrush current limiting
resistor 5 W, 20 A
e.g.
EPCOS/TDK
B57127P509M301
+
Switch closed
for voltage
doubling
Optional 20 A
AC ammeter
measures kW
-
IRV
15.6 V
+
-
100 kW
450 mF
25 V
1
100 kW
1 nF
Optional 1 mA DC
ammeter measures KVA
100 mF
25 V
+
-
14
IRV
CST
15 W
2 W Metal Film
10 W
4W
3.3 nF
12
11
10
9
8
LM339 quad comparator
1 MW
1
2
3
4
5
100 W
1 mF
25 V
6
7
+
-
IRV
100 kW
1 mF
25 V
Current limit set
+
-
50 mH
nominal
3W
10 W
CST
The wires between RM/RMG, the load, and the
collector and emitter of the IGBTs should be
short (less than 8 cm) and fat (14 AWG or larger)
to minimize inductance, The snubber
capacitors/resistors should be located as close
to the IGBT emitter and collector as possible as
shown.
IRV
10 kW
IRV
0.1 mF
5× UF4007 1 A
Fast Switching
Diodes
13
RM
50 turns
26 AWG
3W
3W
3.3 nF/600
V PolyProp
snubber
100 W
1k W
Current limit inductor. Wind on toroidal ferrite core,
3 to 7 cm2 cross-sectional area. Number of turns
depends on permeability of material and crosssection. Normally 7 to 20 turns. Start with a larger
number of turns and decrease if unknown
inductance.
3.3 nF/600
V PolyProp
snubber
`
3 W 10 nF/1000
3 W V PolyProp
snubber 1 turn
15 W
2 W Metal Film
IRV
10 kW
Current sense transformer. Wind on ferrite core,
0.25 cm2 cross-sectional area. Primary is one turn
(wrap wire around toroid once). Secondary is 50
turns of 26 AWG
+
-
4 × 470 nF
PolyProp
600 V
8
IR
2 2153D 7
Half
3 Bridge 6
driver
4
5
10 kW
22 kW
10 W
2× IGBT IKW40N65F5FKSA1
650 V, 40 A collector current, snubber diode integrated.
Aluminum Heatsink (100 to 200 cm 2 area) and fan
cooled. Drain is connected to case, so a thermally
conductive, electrically insulating pad must be placed
between transistor and heatsink if both transistors
placed on the same heatsink!
1 or 2 × 3300 mF 200 to 400 V (depending if
600 V 30 A full- voltage doubling is used and if the mains is
120 VAC or 230 VAC)
wave bridge
rectifier
RM
RM
30 kW
4 W for 170 V,
60 kW
4 W for 330 V
170 to
330 V RM
(rectified
mains)
AC
input
12 V Zener
e.g.
1N4742A
50 mH
nominal
primary
1000
turns 30
AWG
Capacitors 4×
4.1 mF 600 V
metallized
polypropylene
MKP
Transformer. Wind on toroidal ferrite
core, 3 to 7 cm 2 cross-sectional area.
Must be gapped for stable operation
(typical gap is 0.5 mm, can use plastic
shim). Widening the gap decreases
inductance and increases current.
IR2153D half-bridge driver.
Warning: entire circuit at
mains voltage! Do not
touch while energized!
Electrocution may result!
UCC37321/UCC37322 isolated gate drive half-bridge driver
+
mains
250V, 25 A
slow blow
fuse
Double
pole,
double
throw
switch
Inrush current limiting
resistor 5 W, 20 A
e.g.
EPCOS/TDK
B57127P509M301
+
Switch closed
for voltage
doubling
Optional 20 A
AC ammeter
measures kW
-
1
STATE
DISABLE
+
-
1 mF
25 V
2
7
UCC37321
6
3
`
4
`
5
1
8
1 mF
25 V
Current limit inductor. Wind on toroidal ferrite core,
3 to 7 cm2 cross-sectional area. Number of turns
depends on permeability of material and crosssection. Normally 7 to 20 turns. Start with a larger
number of turns and decrease if unknown
inductance.
RMG
4 × 470 nF
PolyProp
RM
600 V
3.3 nF/600
V PolyProp
snubber
1 mF
50 V
ceramic
15 W
2 W Metal Film
3.3 W
4W
RM
10 nF/1000
3 W V PolyProp
3 W snubber
1 turn
15 W
2 W Metal Film
50 turns
26 AWG
3W
3W
CST
3W
10 W
3.3 nF/600
V PolyProp
RMG snubber
+15 V
`
4× UF4007 1 A
Fast Switching
Diodes
`
5
The wires between RM/RMG, the load, and the
collector and emitter of the IGBTs should be
short (less than 8 cm) and fat (14 AWG or larger)
to minimize inductance, The snubber
capacitors/resistors should be located as close
to the IGBT emitter and collector as possible as
shown.
Gate drive transformer. Use a toroidal ferrite core minimum 0.5 cm 2 cross-sectional
area. Wind using a trifliar pattern wiith 12 turns for the primary and the two
secondaries. The gate-emitter connections of the two secondaries must be connected
with opposite polarities to each IGBT as shown, so that the IGBTs turn on alternately.
50 mH
nominal
50 mH
nominal
primary
1000
turns 30
AWG
RMG
RMG
Capacitors 4×
4.1 mF 600 V
metallized
polypropylene
MKP
RMG
2
7
UCC37322
6
3
4
2× IGBT IKW40N65F5FKSA1
650 V, 40 A collector current, snubber diode integrated.
Aluminum Heatsink (100 to 200 cm2 area) and fan
cooled. Drain is connected to case, so a thermally
conductive, electrically insulating pad must be placed
between transistor and heatsink if both transistors
placed on the same heatsink!
Current sense transformer. Wind on ferrite core,
0.25 cm2 cross-sectional area. Primary is one turn
(wrap wire around toroid once). Secondary is 50
turns of 26 AWG
+15 V
8
+
-
+
-
22 kW
10 W
+
-
1 or 2 × 3300 mF 200 to 400 V (depending if
voltage doubling is used and if the mains is
120 VAC or 230 VAC)
600 V 30 A fullwave bridge
rectifier
+15 V
100 mF
25 V
170 to
330 V RM
(rectified
mains)
AC
input
Transformer. Wind on toroidal ferrite core, 3 to 7 cm2 crosssectional area. Must be gapped for stable operation (typical
gap is 0.5 mm, can use plastic shim). Widening the gap
decreases inductance and increases current.
UCC37321/UCC37322 half-bridge
driver switch section Warning: parts
of the circuit are at mains voltage!
Risk of electrocution!
NOTE: the drive circuit and the mains are galvanically isolated by the gate drive and
current sense transformers. DO NOT tie the grounds together of these circuits!
+15 V
16
15
14
13
12
11
10
9
Frequency divider
to balance on/off
time
CD4027 JK flip flop
1
2
3
4
5
6
7
8
STATE
+15 V
+15 V
+15 V
10 kW
Schmitt trigger
relaxation oscillator
50 kW
100 kW
100 kW
100 kW
100 kW
10 kW
Optional 1 mA DC
ammeter measures KVA
+15 V
+15 V
470 pF
10 kW
10 kW
DISABLE
1k W
14
13
12
1
2
3
+15 V
CST
10 W
4W
3.3 nF
1 MW
10
9
8
5
6
7
12 V Zener
e.g.
1N4742A
LM339
4
+15 V
+15 V
100 W
0.1 mF
5× UF4007 1 A
Fast Switching
Diodes
11
1 mF
25 V
+
-
100 kW
+
1 mF
25 V
Current limit set
Oscillator and
overcurrent detector.
Warning: parts of the
circuit are at mains
voltage! Risk of
electrocution!
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R. Hartwell, “A KW Switch Mode Regulated High Voltage Power Supply,” retrieved from
http://w5jgv.com/hv-ps1/ .
Data Sheets:
“LM139/LM239/LM339/LM2901/LM3302 Low Power Low Offset Voltage Quad Comparators,”
National Semiconductor March 2004.
“L6565 Quasi-Resonant SMPS Controller,” ST Microelectronics January 2003.
“IR2153(D)(S)&(PbF) self-oscillating half-bridge driver,” International Rectifier, data sheet no
PD60062.
“UCC2732x/UCC3732x Single 9-A High-Speed Low-Side Mosfet Driver With Enable,” retrieved from
http://www.ti.com/lit/ds/symlink/ucc37321.pdf .
“TL494 Pulse-Width-Modulation Control Circuits,” retrieved from
http://www.ti.com/lit/ds/symlink/tl494.pdf .