A Low-Noise Integrated Bioamplifier with
Active DC Offset Suppression
Wei Zhao, Hongge Li*, Youguang Zhang
School of Electronic and Information Engineering
Beihang University
Beijing, China
*honggeli@buaa.edu.cn
A large amount of neural amplifiers have been presented
in the literature. There are few DC offset suppression schemes
adopted for on-chip integration in neural recording system.
First method that uses a capacitor feedback network with ac
coupling of input devices to reject the offset is very popular in
designs [1], [3], [5]. The MOS-bipolar pseudo-resistor
component in parallel with a capacitor feedback provides the
very small low-cutoff frequency. The midband gain is the ratio
of feedback capacitors. The major drawback is this
configuration requires very large capacitors which occupy
great percents area (>60%) of chip. The second method
employs a closed-loop resistive feedback and electrode
capacitance to form a highpass filter [4], [6]. The
performances of this approach depend on the electrode
parameters which change from one site to another. The third
method adopts active low-frequency suppression that an active
integrator in the feedback path of low noise amplifier forms a
highpass cutoff frequency response [2]. This technology
achieves small size without large passive devices. However,
the amplifier works with the open-loop mode in the midband,
and the gain suffered from the variation of the open-loop gain
of low noise amplifier.
Abstract—This paper describes a novel low-power, low-noise
amplifier for neural recording applications. The bioamplifier
achieves the best power-size tradeoff compared to the previous
design. By means of a new active feedback configuration, the DC
offset is rejected without the large capacitors. An active
differentiator with an amplifier in the feedback path places a
high-pass cutting frequency in the transfer function. The
midband gain consists of the passive components, and is
insensitive to the mismatch of process. The bioamplifier has
been implemented in the Chartered 0.35-μm 2P4M CMOS
process and occupies 0.022mm2 of chip area. The current
consumption of amplifier is 2μA at ±1.5V supply. The
bioamplifier achieves a midband gain of 46dB and a -3dB
bandwidth from 13Hz to 8.9kHz. The input-referred noise is
5.7μVrms corresponding to an NEF of 3.1.
I.
INTRODUCTION
The clinicians demand a system to record the simultaneous
activities of large numbers of neurons in the brain for
understanding how the brain works. Neural recording from the
large-scale chronic multi-electrodes has been widely used in
many biomedical applications [1], [2], [7]. The bioamplifier is
one of the most important parts in this system.
This paper presents a bioamplifier by means of a novel
active DC offset suppression which takes the merits of
capacitors feedback network configuration [1] and active lowfrequency suppression configuration [2]. The midband gain is
determined by the ratio of the capacitors, and size of
capacitors is similar to the active cutoff-frequency suppression
configuration without increasing the power and noise.
Neural signals from extracellular recording are very weak.
The amplitude is in 50−500μV range with the noise level in
5−10μV. Such signals cannot be processed further without
amplification. To increase the SNR of the system, the input
referred noise of the amplifier need to minimize. However,
the tradeoff between low-noise and low-power designs puts
the challenge to the designers. The multi-electrodes recording
system incorporates a large number of neural amplifiers. Too
much power dissipated by the amplifiers may damage the
surrounding issues by heating. The electrode shape restricts
the size of the each amplifier as low as 0.16mm2 [1]. Due to
the electrode potential mismatch, the DC offset up to 1V is
cross the recording electrodes. The offset which is orders of
magnitude larger than the biopotential signal, must be rejected
to prevent the saturation of the amplifier. The power, noise,
size and DC offset are the major problems faced by designers.
II.
This work was supported by National Natural Science Foundation of
China (60971084).
978-1-4244-4918-7/09/$25.00 ©2009 IEEE
ARCHITECTURE OF BIOAMPLIFIER
Fig. 1(a) shows a schematic of our bioamplifier design
with the proposed DC offset suppression configuration. This
design consists of a differentiator and a close-loop amplifier
within its feedback path. The differentiator is made from an
amplifier (A1), capacitors (C3, C4), and a high-value resistor
(Req). A close-loop amplifier is composed of an amplifier (A2),
and feedback capacitors (C1, C2). The differentiator is
equivalent to a highpass filter to reject the DC offset. The
5
MR1 MR2
Vin
Req
Vin(s)
C3
Vout+
Vout -
A1
C2
F2(s)
-1
G(s)
C4
A2
Vout(s)
F1(s)
C1
(a)
(b)
Figure 1. (a)Schematic of the bioamplifier and (b)equivalent feedback
control system for the bioamplifier transfer function.
Figure 2. Equivalent model for evaluating the effect of non-ideal factors
to bioamplifier.
close-loop amplifier in feedback path corrects the highpass
function of the differentiator to form the stable midband gain.
Fig. 1(b) shows a functional block diagram of the
bioamplifier which consists of differentiator F1, F2 and
amplifier G with capacitors feedback network. In the case of
all the amplifiers are ideal with no parasitic capacitance, the
transfer function of the ideal bioamplifier is
sReq C 2 C 3
Vout ( s )
F1
=
=−
Vin ( s ) 1 − F2 ⋅ G
sReq C1C 4 + C 2
AV 1 (4)
If Av1 fails to satisfy (4), |Vout(s)/Vin(s)| will be smaller
than C2C3/C1C4. In order to characterize the effect of finite
gain on the midband gain, a parameter α is defined as
(1)
V out ( s )
= α AM
V in ( s )
The midband gain AM=−C2C3/C1C4 is comprised of two
capacitive ratios. The gain of the new bioamplifier architecture
equals two previous amplifiers working in series. For a given
gain, the bioamplifier demands smaller capacitors.
(5)
The actual midband gain falling into α times of ideal
value requires the gain Av1 to be
The transfer function reveals the −3dB cutoff frequency to
1
C2
1 AM
=
2π Req C1C 4 2π Req C 3
C 1C 4
the midband gain of bioamplifier is the same as ideal
condition.
where F1(s)=−SReqC3, F2(s)=−SReqC4, G(s)=−C1/C2.
be
( C 3 + C 4 + C in1 ) C 2
AV 1 ≥
1
1 α −1
AM
(6)
(2)
The gain Av2 of amplifier A2 has no effect on the midband
gain in the case of Av2≫ (C1+C2)/C2 which can achieve easily.
To achieve −3dB highpass cutoff frequency around 10Hz
would requires the resistor Req in the order of 1012Ω.
Transistors MR1−MR2 are MOS-bipolar devices which act as
pseudo resistors [1]. For small input amplitude, the equivalent
resistor is more than 1013Ω.
Considering the A1 as a simple first-order model for the
transfer function of a dominant-pole compensated amplifier,
Av1(s), is given by
fH =
AV 1 ( s ) =
In the previous section, the transfer function is deduced
under ideal conditions. The amplifier finite DC gain, parasitic
capacitances and noise are devoted to errors that alter the
transfer function of bioamplifier. Equivalent model of these
non-ideal factors in the circuits is shown in Fig. 2.
AV 1,open
1 + s ω0
(7)
Where Av1,open=GmRout is the DC gain of amplifier A1, and
ω0 =1/(RoutCL) is the dominant pole. The −3dB lowpass cutoff
frequency can be evaluated from (3)
fL =
A. Effect of amplifier finite gain
The finite gain of the amplifier (A1 and A2) donated by Av1
and Av2, respectively. Cin1 and Cin2 are the input parasitic
capacitors of the amplifier. For the case where Av2C2≫
C1+C2+Cin2 and Av1≫1, the transfer function is
1 g m C1 C 4
1 gm
=
AM
2π C L C 2 C 3 2π C L
(8)
The -3dB lowpass cutoff frequency depends on the
transconductance gm of A1, load capacitor CL and midband
gain AM of bioamplifier.
B. Effect of noise
Noise sources associated with the amplifier A1 and A2 are
represented as Vn1 and Vn2, respectively. Thermal noise
source of the equivalent resistor Req is VR, as shown in Fig. 2.
sReq AV 1C 2 C 3
Vout ( s )
(3)
≈−
Vin ( s )
sR ⎡⎣ AV 1C1C 4 + ( C 3 + C 4 + C in1 ) C 2 ⎤⎦ + AV 1C 2
When Av1 satisfies the following relationship
The input referred noise power of the bioamplifier can be
obtained, as follows
6
⎛ 1
V ≈⎜
⎜ sReq C 3
⎝
2
n
2
gain (−C1/C2). In the case of the amplifier is the simple pole
configuration, the open-loop gain is Av2,open and the main pole
is ω1. The block G has the corner frequency that is equal to the
gain-bandwidth product (Av2,open∙ω1) and the phase is 90°. The
neural signal bandwidth must be within this corner frequency.
2
⎞
⎛ C + C 4 + C in1 ⎞
2
2
⎟⎟ ⋅ Vn , R + ⎜ 3
⎟ ⋅ Vn ,1
C3
⎝
⎠
⎠
(9)
2
⎡ ( C + C 2 + C in 2 ) C 4 ⎤
2
+⎢ 1
⎥ ⋅ Vn ,2
C
C
2 3
⎣
⎦
The proposed bioamplifier configuration presents a phase
shift of 180° between node Vout and −Vout. The differentiator
F2 (Fig. 1(b)) generates a negative phase shift of −90° in loop
gain. The block G introduces a positive phase shift φG. The
φG equals 90° under the assumption that the amplifier A2 has
only the dominant pole ω1. If the A2 has the other pole high
than ω1, the phase shift φG will be large than 90°. Therefore,
the total phase shift introduced in the loop gain is
φLG(jω)=180°−90°+φG. At unity-gain frequency, the phase
margin (PM) of the whole bioamplifier is
The flick noise limits the performance of the low power,
low frequency amplifier. In order to minimize the flick noise,
the input devices of amplifier are PMOS transistors with large
area. The large input transistors increase the input capacitance,
and decrease the noise performance as show in (9). The design
of amplifier A1 needs to take care of the tradeoff between the
flick noise and input capacitance. The noise of amplifier A2
has less contribution to the input referred noise of bioamplifier.
For the case where C2=C3=10C1=10C4, Cin1=Cin2, the input
referred noise related noise source Vn2 is
PM = ϕ LG ( jω t ) − 180 ° = ϕ G − 90 °
2
V n2,in 2 =
1 ⎛ C 3 + C 4 + C in1 ⎞
1
2
Vn2,in1
⎜
⎟ ⋅ Vn ,1 =
100 ⎝
100
C3
⎠
(10)
Hence at ωt, a phase shift of φG=90° yields a PM of zero.
Where a phase shift of φG>90° yields the PM>0.
The gain and noise of A2 have less effect on bioamplifier
that allows us to relax the requirement of design for saving
power. Integrated the input referred noise given by Req is
V
=
∫
fH
fL
CIRCUIT IMPLEMENTATION
III.
The midband gain of the bioamplifier is the product of
C2/C1 and C3/C4. If the gain is constant, decrease the total
capacitance demands minimum the C1 and C4. However the
parasite capacitance restricts the minimum value of C1 and C4.
The total size of capacitor is minimum under the relation of
C2/C1=C3/C4. This scheme is not the optimization of noise.
Increase C3/C4 can attenuate the noise with Vn1 as given in (9).
The optimal value of C3/C4 is select to achieve the minimum
noise-capacitor product for the bioamplifier. With the
midband gain of 200, the normalization of noise-capacitor
product varies with C3/C4 is show in Fig. 4.
2
2
n , inR
⎛
⎞ 2
V n2, R
1
f H − f L (11)
⎜⎜
⎟⎟ Vn , R df =
π
fR
C
π
R
C
fH fL
2
2
eq 3 ⎠
eq 3
⎝
The noise of equivalent resistor Req undergoes attenuation
in the bioamplifier. Therefore, the input referred noise of
bioamplifier is donated by A1 only.
Normalization of
noise-capacitor product
C. Analysis of Stability
As depicted in Fig. 3(a), the magnitude of differentiator
increases with 20dB/decade and its phase is −90°. The nonideal differentiator presents a finite gain at the high frequency
that equal to the gain Av1 of amplifier A1. The differentiator
low corner frequency is at ω=Av1/(Req∙C). Notice that the Av1
must be as large as possible to fade non-ideal effects in
differentiator. Assume the τp is the dominant pole of the
amplifier A1. The differentiator has the high corner frequency
of τp. As the frequency is high than τp, the gain of
differentiator decreases and the phase is -180°. By this reason,
the τp should be large than the neural signal bandwidth.
80
60
40
20
5
10
M18
VCM
M19 M13
+Vout -
M20
A1 RC
τp
ω ( rad s )
25
30
Ibias1
ω ( rad s )
M14
M15
Vb1
M21
C2
Aω 0 ≈ Aω0
C1 + C 2
20
Figure 4. For the midband gain is 200, the normalization of noisecapacitor product varies with C3/C4.
C1
C2
ABW
15
C3/C4
The block G(s) which consisted of the amplifier A2 and
capacitors feedback network as show in Fig. 1 (b) has the DC
A1
(12)
1.6uA
Ibias2
Vin+
M16
M1
M2
Vb3
M3
M7
Vin - M9
M4
ω ( rad s )
M5
M22
ω ( rad s )
R
Figure 3. Magnitude and phase responses of (a) the differentiator and
(b)the amplifier with capacitors feedback network.
R
M11
M8
+Vout -
M17
Vb2
M10
Vb4
M12
Ibias3
M18
50nA
M6
R
Figure 5. Circuit schematic of the low power, low noise amplifier A1.
7
50
The amplifier A1 has a folded-cascode topology with
source-degenerated current mirrors [3] showed in Fig. 5. The
full differential configuration which has the high CMRR to
reject interference is adopted. The common mode feedback
circuit is formed by the transistor M13-M14 in triode region to
sense the CM level. The DC gain of A1 should satisfy the
relationship given in (6). For α=0.95, the gain cannot be low
than 72dB. The −3dB bandwidth of A1 is determined by
neural signal bandwidth. Simulations show that the amplifier
A1 has a DC gain of 138dB, a phase margin of 65°, and a
bandwidth of 1.9MHz.
-200
30
20
0
10
1
2
10
3
4
10
10
Frequency (Hz)
10
5
10
Figure 7. Transfer function of bioamplifier.
2
(V/(Hz)1/2)
Input referred noise
-7
As stated above, the finite DC gain, noise and bandwidth
of amplifier A2 have less impact on the bioamplifier. The
specifications can be relaxed for A2 to save the power. A2 has
the same configuration as A1 except using the common current
mirrors instead of the source-degenerated current mirrors. The
DC gain of A2 is 136dB. The bandwidth is 892 kHz and the
phase margin is 57°, while consuming 300nA.
IV.
-100
40
Phase (deg)
Magnitude (dB)
The noise-capacitor product has the minimum value of
C3/C4=16 by setting the value of C3 to 1.6pF, C2 to 1.25pF, C1
and C4 to 100fF using the poly-poly capacitors. The total
capacitance of the full differential scheme is only 6.1pF.
x 10
1.5
1
0.5
1
10
2
3
10
10
4
10
Frequency(Hz)
Figure 8. Input referred voltage noise density of bioamplifier.
The thermal noise level is about 56nV/√Hz. Integrated noise
from 10Hz to 10kHz yields an RMS noise of 5.7μVrms. Table I
shows the state of the art in bioamplifier. The noise efficiency
factor (NEF) is a measure of the noise performance of a given
bioamplifier as a function of its power consumption and the
bandwidth. Base on the simulation results, the NEF of this
bioamplifier is found to be 3.1.
RESULTS
The bioamplifier with the proposed active dc offset
suppression architecture was designed using the Chartered
0.35μm 2P4M CMOS process. The midband gain of 46dB and
the bandwidth from 13Hz to 8.9 kHz have been shown in Fig.
7. The bioamplifier with full differential scheme fits in
120μm×184μm area.
V.
CONCLUSION
A novel active dc offset suppression structure has been
proposed which can be applied to low power, low noise, and
area efficient bioamplifier. This structure features the small
feedback capacitors and less gain variation. Simulation results
of the bioamplifier with a midband gain of 46dB have the
input-referred noise of 5.7μVrms over 10Hz-10kHz. The full
differential bioamplifier with the small chip area as singleend output amplifier has high immunity to interference.
120μm
Fig. 8 shows a plot of the simulated input-referred noise.
REFERENCES
[1]
184μm
Figure 6. Layout of the bioamplifier.
[2]
TABLE I
PERFORMANCE CHARACTERISTIC OF BIOAMPLIFIER
[1]
Gain
(dB)
39.5
[2]
49.5
[3]
40.8
5
39.3
39.3
45.6
43.5
46
[4]
[5]
[6]
This
work
4
Noise
(μVrms)
2.2
4.9
5.6
45-5.32k
2.67
3.06
0.001-5k
19.4
7.8
0.015700
0.001-5k
13-8.9k
4.9
3.6
10
3.1
3.66
5.7
BW
(Hz)
0.0257.2k
98.4-9.1k
NEF
Power
16μA
@±2.5V
4.67μA
@1.8V
2.7μA
@2.8V
38.3μA
@±1.5V
8μA
@±1.7V
22μA
2μA
@±1.5V
Area
(mm2)
0.160
[3]
[4]
0.050
[5]
0.160
0.107
[6]
0.201
0.05
0.022
[7]
8
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