APPLICATION NOTE
Mailing Address: PO Box 11400 • Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd. • Tucson, AZ 85706
Tel: (602) 746-1111 • Twx: 910-952-1111 • Telex: 066-6491 • FAX: (602) 889-1510 • Immediate Product Info: (800) 548-6132
CURRENT OR VOLTAGE FEEDBACK:
THE CHOICE IS YOURS WITH THE NEW, FLEXIBLE,
WIDE-BAND OPERATIONAL AMPLIFIER OPA622.
By Christian Henn and Andreas Sibrai, Burr-Brown International GmbH
With the recently introduced wide-band op amp OPA622,
Burr-Brown has reached a new height in op amp design. In
the past, engineers designing a circuit with feedback had to
choose between voltage and current feedback according to
the requirements of their particular applications. Both feedback types involve a trade-off. The current-feedback structures available up to now use their symmetrical circuit
design and short feedback loop to process wide-band analog
signals, while the traditional voltage-feedback amplifiers
provide more optimized DC performance but slow down the
signal processing rate. But with the OPA622, an IC is
available that can be configured for both modes. The first
voltage-feedback op amp manufactured using a complementary circuit technique, the OPA622 achieves bandwidths and
slew rates previously attainable only with current-feedback
amplifiers, while also offering two identical high-impedance
inputs, improved common-mode rejection, and external adjustment of the open-loop gain and quiescent current. The
OPA622’s extremely flexible pin configuration lets the user
assemble it as a voltage-feedback op amp, a fast comparator,
an AGC amplifier, an open-loop or direct-feedback amplifier, and even a 350MHz current-feedback amplifier. Its
powerful output stage can easily drive 50Ω and 75Ω transmission systems and operates stably on capacitive load
resistors. This application note will present the internal
circuit configuration, specifications, and frequency response
alignment of the OPA622 and will also describe its diverse
applications.
impedance output, followed by a buffer amp as impedance
converter. Between these two components are a resistor and
a capacitor to determine the open-loop gain, slew rate, and
bandwidth. The differential amplifier charges the capacitor,
C, with quiescent current for rising and falling signals so that
the slew rate can be determined as follows:
CLASSICAL CIRCUIT TECHNIQUES
CURRENT-FEEDBACK CONFIGURATION:
THE ALTERNATIVE OF THE 80s
About ten years ago, current-feedback amplifiers were developed as an alternative to conventional op amps. They
consist of a transconductance amplifier in Diamond structure and an output stage made up of complementary emitter
followers as shown in Figure 2. The feedback loop connects
the output of the amplifier to the low-impedance input, thus
transforming the usual voltage feedback into current feedback. The current-feedback method not only allows optimal
frequency response adjustment using the parallel impedance
of the feedback network (which also influences the openloop gain) but also eliminates the need for an internal
compensation capacitor. The design does have one parasitic
capacitor at the high-impedance OTA output, but its capacitance is much smaller than that of compensation capacitors
in classical configurations, and the improvement in capacitor charging (10 to 20 times IQ) produces slew rates of up to
SRMAX =
R1
f–3dB =
R2
C
B
R3
FIGURE 1. Operational Amplifier Consisting of a
Transconductance Amplifier (TA) and Buffer
(B).

1993 Burr-Brown Corporation
SBOA073
2
2π
• SR/Vp0
Internally compensated amplifiers, which include most classical amplifiers, use an integrated capacitor for the worst
case or smallest closed-loop gain. This compensation capacitor reduces the maximum open-loop gain to
–6dB per octave starting at very low frequencies but ensures
sufficient phase margin for stable operation even at gain +1.
This method of frequency response adjustment is not at all
suitable for wide-band amplifiers, since the compensation
capacitor allows neither slew rates over 1000V/µs nor largesignal bandwidths over 100MHz.
VOUT
TA ∞
V
S
Usually, a sine-wave signal is applied to an op amp to
determine its –3dB bandwidth. Since sine-shaped signals
have the largest signal variation at the zero crossing point,
the –3dB bandwidth of the op amp can be calculated by the
following equation:
As shown in Figure 1, a classical op amp consists of a
differential or transconductance amplifier (TA) with high-
VIN
∆V
I
=
∆t
C
AN-186
Printed in U.S.A. October, 1993
2000V/µs and large-signal bandwidths of up to 250MHz.
The current-feedback configuration does, however, have
drawbacks such as asymmetrical inputs, reduced commonmode rejection, and relatively high input voltage offset
compared to state-of-the-art conventional op amps.
the OPA622 comprises an OTA, a feedback buffer, and a
±70mA output stage. The differential input stage with highimpedance inputs is made up of two identical complementary buffers so that the only input offset voltage is the
difference between their offset voltages. Both buffer outputs
are connected to package pins and to each other via the
resistor ROG. This resistor functions like the emitter degeneration resistor of a classical 2-transistor differential stage
and allows external open-loop gain adjustment. Figure 4
shows the frequency response of the open-loop gain at
various ROG values. When the input voltage is differential, a
current flows through the R OG. The current mirror in the
OTA reflects this current to its high-impedance output,
which is decoupled by the output stage and functions practically in open-loop mode. The feedback loop connects the
output to the input of the feedback buffer (FB), which when
inserted switches the circuit from current-feedback to voltage-feedback mode.
THE WHOLE WORKS:
CURRENT AND VOLTAGE FEEDBACK IN ONE
The OPA622 combines the speed of a current-feedback
design with the precision of a voltage-feedback design using
two identical high-impedance inputs. As shown in Figure 3,
+VCC
SR+MAX ≈ 10 I/C
SR–MAX ≈ 10 I/C
I
FLEXIBILITY
When defining a product, engineers always face a conflict
between ensuring pinout compatibility, providing the smallest possible package, and guaranteeing circuit flexibility.
Only with a flexible configuration is it possible for engineers
to adapt an IC to their particular applications or to implement functions that are impossible or extremely difficult to
achieve with standard op amps. The OPA622 makes it easy
for users to customize the configuration to their particular
needs and allows multiple access points to the inputs and
outputs of the individual circuit parts. The compromise for
the OPA622’s flexibility is its slightly reduced AC performance, as its multiple pin-to-pin connections generate a
longer delay time in the feedback loop. For optimum AC
B
C
R3
VIN
VOUT
R2
I
–VCC
FIGURE 2. Current-Feedback Amplifier in Diamond Structure.
–VCC
12
+VCC OUT
+In
4
OTA
13
ROG
8
FB
3
11
10 COTA
–In
9
OB
6
–VCC OUT
R2
5
–VCC
R1
CCOTA : Sets the first open-loop pole.
ROG : Sets the open-loop gain.
R2/R1 : Sets the closed-loop gain.
FIGURE 3. Voltage-Feedback Amplifier in Diamond Structure.
2
VOUT
performance and for standard circuit functions, the OPA623
and the planned 8-pin standard pinout voltage-feedback
amplifier series OPA655/6/7/8 are the better choice.
PIN NO.
DESCRIPTION
FUNCTION
2
3
4
5
6
IQ Adjust
–In
+In
–VCC
–VCC OUT
8
9
10
11
BUF–
VOUT
OTA
+VCC OUT
12
13
+VCC
BUF+
Quiescent Current Adjustment: typ 3-8mA
Inverting Analog Input
Noninverting Analog Input
Negative Supply Voltage: typ –5VDC
Negative Supply Voltage Output Buffer:
typ –5VDC
Analog Output Feedback Buffer
Analog Output
Analog Output OTA
Positive Supply Voltage Output Buffer:
typ +5VDC
Positive Supply Voltage: typ +5VDC
Analog Output/Input
60
ROG = 0Ω
50
Gain (dB)
40
ROG = 27Ω
30
20
ROG = 150Ω
10
TABLE I. Functional Description of the Pin Configuration.
0
ROG = 390Ω
–10
+VCC
–20
10k
100k
1M
10M
100M
12
1G
OTA
Frequency (Hz)
+ VCC OUT
FIGURE 4. Open-Loop Gain at Various R OG Values.
COTA
Figure 5 shows the pin configuration of the OPA622, and
Table I describes the function of the individual pins. The
OPA622 is available in a 14-pin DIL or SO package and is
specified over the industrial temperature range from
–40°C to +85°C. The internal power supply provides the
quiescent current, which rises with temperature to maintain
constant AC performance. In addition, the external resistor,
RQ, allows the user to vary the total quiescent current
consumption between ±3mA and ±8mA. The quiescent
current is specified at ±5mA using a quiescent current
resistance (RQ) of 430Ω. RQ is connected to Pin 5 and the
negative supply voltage (–5V). In normal operation, a negative current flows at Pin 2. If the user forces a positive
current by using an external current source, the OPA622 is
switched off and requires practically no current. Figure 6
shows the OPA622 as a current-feedback amplifier, which
offers 350MHz bandwidth at 2.8Vp-p and a slew rate of over
2000V/µs. The feedback buffer is not necessary for the
current-feedback configuration so it can be used for other
Top View
NC
13
1
4
IQ Adjust
2
–In
3
+In
4
–VCC
5
–VCC OUT
6
NC
7
OB
5
5
FIGURE 6. OPA622 as a Current-Feedback Amplifier.
applications. Since the output stage of the feedback buffer is
purposely small to ensure a short delay time, and since it
may not be overloaded, the load resistance at the maximum
output signal should not exceed 500Ω.
One special feature of the OPA622 is its separate supply pins
for the differential amp and the output buffer, which is
capable of driving the ±70mA output stage. The separate
supply decouples the differential amplifier from the output
stage, through which large charge currents must flow at
200MHz large-signal bandwidth, and also improves the
pulse response and allows various power supply sensing
techniques for higher output voltage swings. In addition, it
is possible to limit the current consumption to protect the
output stage from overload.
12 +VCC
Figure 7 shows inverting and noninverting versions of the
OPA622 in current- and voltage-feedback modes.
11 +VCC OUT
FB
8
Biasing
12
2
–VCC OUT R2
–VCC
13 BUF+
9
–VOUT
R1
14 NC
6
3
BUF+
OTA
11
10
13
SO/DIP
OPA622
11
9
OB
6
10
+In 4
OPTIMIZING THE FREQUENCY
RESPONSE ADJUSTMENT
10 OTA
9
VOUT
8
BUF–
Analyses of various op amp configurations have proven that
the frequency response is optimally flat when the following
equation holds:
1
C/gm = 2kO • TD, GCL =
kO
FIGURE 5. Pin Configuration.
3
20
Voltage-Feedback
Non-inverting
VOUT
OTA
15
Inverting
VOUT
OTA
OB
10
OB
R2
ROG
R2
ROG
FB
FB
R1
GCL = 1 +
Gain (5dB/Div)
+VIN
R2
R1
GCL = 1 –
R1
–VIN
R2
5
0
–5
GCL = +10
ROG = 10Ω
GCL = +2
ROG = 150Ω
GCL = +1
ROG = 390Ω
GCL = –1
ROG = 200Ω
GCL = –2
ROG = 120Ω
–10
–15
–20
–25
R1
OPA622AP
VO = 1.4Vp-p
–30
100k
Current-Feedback
Non-inverting
VOUT
OTA
1M
100M
1G
VOUT
OTA
OB
10M
Frequency (Hz)
Inverting
OB
FIGURE 8. Frequency Response at Various Gains.
+VIN
R2
R2
10
GCL = 1 +
R2
R1
FB
GCL = 1 –
R1
5
–VIN
0
Amplitude (dB)
R1
FB
R2
R1
FIGURE 7. Overview of the Various Op Amp Configurations Possible with the OPA622.
–5
ROG = 50Ω
ROG = 150Ω
–10
ROG = 300Ω
–15
–20
GCL = +2V/V
–25
100k
1M
Thus the ratio of the capacitance, C, at the high-impedance
OTA output to the internal OTA transconductance is equal
to the closed-loop gain times the delay time (T D) times 2.
Adjusting the open-loop gain externally to produce optimal
frequency response is the best way to guarantee that wideband amplifiers will function stably in feedback mode. As
already shown in Figure 4, raising R OG lowers the open-loop
gain while simultaneously maintaining sufficient phase
margin. What’s important for wide-band amplifiers is that
the bandwidth does not decrease with open-loop gain—thus
the gain-bandwidth product rule, stating that the gain bandwidth of internally compensated op amps decreases with
increasing closed-loop gain while the product of the gain
times the bandwidth remains constant, no longer applies.
Figure 8 shows that the OPA622 is quite successful in
putting these theoretical analyses into practice. At a gain
range of –2 to +10 and output voltage of 1.4Vp-p, the –3dB
bandwidth varies from 110MHz to 230MHz while R OG
varies from 10Ω to 390Ω. The differences between the
theoretical and real measurements are due to parasitic capacitances and other minor influences. Figure 9 shows the
effect of varying R OG at constant closed-loop gain. The
frequency response is optimally aligned at a gain of +2 and
ROG of 150Ω. Increasing the ROG flattens the frequency
response curve, while decreasing R OG produces a rise in
frequency response at the end of the bandwidth. In some
applications, slight peaking is useful; while increasing overshooting, it also expands the –3dB bandwidth and lowers the
pulse slew rates.
10M
100M
1G
Frequency (Hz)
FIGURE 9. Effect of ROG on the Frequency Response Curve.
The ability to adjust the open-loop gain externally lets users
adapt the frequency response to capacitive load resistances
to a certain extent. Figure 10 shows the flat frequency
response attained at gain +2 for three different load capacitances.
To put the finishing touches on the general presentation of
the OPA622, Figures 11, 12, and 13 show the bandwidth at
different output signals, the pulse response at 5Vp-p, and the
group delay time, respectively.
The broad range of applications for the OPA622 impressively demonstrates the flexibility of its individual circuit
parts and pin configuration. The OPA622’s specifications,
which are shown in Table II, fulfill requirements for applications in high-speed analog and digital communications,
broadcasting and video, test and instrumentation, fiber optic
transmission, and data acquisition equipment. Typical applications include an input differential amplifier for test equipment and monitors, a line driver for analog and digital
systems, an ADC input and DAC output amp, a multiplier
output amp, and a magnetic head driver when combined
with a discrete current source. The following discussion
presents several applications in more detail.
4
LARGE SIGNAL PULSE RESPONSE
GCL = +2V/V, VO = 2.8Vp-p
2.5
Input/Output Voltage (V)
Gain (5dB/Div)
47pF
22pF
10pF
CLOAD
ROG
COTA
10p
22p
47p
180Ω
200Ω
150Ω
0.5p
0.5p
0.5p
300k
VIN
VOUT
0
–2.5
1M
10M
100M
G = +1V/V, VO = 5Vp-p, tRISE = tFALL = 1ns (Generator)
1G
Frequency (Hz)
0
10
20
30
40
50
60
70
80
90 100
Time (ns)
FIGURE 10. OPA622 Bandwidth at Various Capacitive
Load Resistances.
FIGURE 12. Pulse Response at 5Vp-p.
20
5Vp-p
10
2.8Vp-p
5
1.4Vp-p
0
GROUP DELAY TIME
0.25
0.20
Delay Time (ns)
15
0.6Vp-p
–5
–10
0.2Vp-p
–15
–20
–25
dB
300k
Group Delay Time
0.10
150Ω
50Ω
VIN
GCL = +2V/V
1M
0.15
VOUT
DUT
0.05
10M
100M
DEM-OPA622-IGC GCL = +2V/V
1G
0
100k
Frequency (Hz)
1M
10M
100M
1G
Frequency (Hz)
FIGURE 11. OPA622 Bandwidth at Various Output
Signals.
Large Signal Bandwidth (5Vp-p)
Slew Rate
Gain Deviation: DC to 30MHz
DC to 100MHz
Quiescent Current Consumption
Input Bias Current
Output Current
Offset Voltage
Common-Mode Rejection
FIGURE 13. Group Delay Time.
corresponding winding ratio to achieve the required output
impedance matching. In the application shown in Figure 14,
the OPA622 with its high-impedance inputs converts the
complementary output currents of the multiplier into an
asymmetrical output voltage. The low-impedance output can
be adapted to 50Ω or 75Ω systems by inserting a series
resistor. In this way, the OPA622 can be used as a true
differential amplifier.
1700V/µs (SOIC)
200MHz (SOIC)
0.12dB
0.3dB
±5mA
–1.2µA
±70mA
100µV
78dB
TABLE II. Typical Parameters of the OPA622 as a
Voltage-Feedback Amplifier.
For the multiplier to function perfectly, both open collector
outputs have to have a potential of more than the positive
supply voltage to prevent saturation of the output transistors.
Additional resistors (R3 and R4) are located in series to the
multiplier supply pins so that no extra power supply is
necessary. These resistors reduce the supply voltage of the
multiplier to less than that required by the output amplifier.
We recommend raising the supply voltage from typically 5V
to 6V to expand the common-mode output range of the
OPA622 and to improve the multiplier’s dynamic performance.
A TRUE DIFFERENTIAL AMPLIFIER
FOR WIDE-BAND MULTIPLIERS
Four-quadrant, wide-band multipliers produce differential
output currents to achieve the high bandwidth required by
their applications. A multiplier’s differential open collector
outputs are undesirable, however, in applications in which
the multiplication should produce a voltage referred to
ground. Since current-feedback amplifiers have asymmetrical inputs, they require complex discrete circuitry to be used
as output amplifiers for wide-band multipliers. Another
possibility is to use an RF transmitter with a sender tap and
Using the multiplier, it is possible to multiply wide-band
signals of up to 100MHz with little distortion. Other typical
5
+5V
C1
470pF
+ C
3
2.2µF
C2
10nF
R3
100Ω
ST 1
R7
100Ω
R8
100Ω
R5
10Ω
C7
1µF
X1
OPA622
U2
14
1
NC
NC
R1
13
2
X2
8
X2
7
6
5
X1
+VS
W1
150Ω
–ROG
–In
+VCC
+In
+VCC OUT
12
3
150Ω
Wideband
Multiplier
IQ Adjust
11
4
U1
10
5
Y1
1
Y2
–VS
W2
2
3
4
–VCC
C8
1µF
R2
–VCC OUT
VOUT
NC
+ROG
ST 2
R11
330Ω
VOUT
8
7
R9
100Ω
Y2
R13
50Ω
9
6
Y1
C9 1µF
OTA
R12
100Ω
R6
10Ω
R10
330Ω
R4
100Ω
–5V
C4
470pF
C5
10nF +
C6
2.2µF
FIGURE 14. Output Amplifier for Wide-Band Multipliers.
multiplier applications include modulation, demodulation,
signal control, mixing, phase detection, and fast video switching and keying.
resistors R1 and R2. Signal excitation at Pin 4 produces
noninverting mode, while excitation at Pin 3 produces inverting mode.
EXPANDING THE OPEN-LOOP AMPLIFIER
Amplifiers with no external feedback loop deliver bandwidths and slew rates that are unattainable using currentand voltage-feedback op amps. They can not, however,
provide the benefits of a feedback loop, such as improvement in linearity, reduction in output impedance, performance adjustment using the feedback network, and
reduced sensitivity to parameter and temperature variations
and aging effects. The OPA622, however, provides a solution combining the advantages of both methods. As shown
in Figure 15, the OPA622 functions like a discrete common
emitter circuit. Its working points are internally fixed and the
transconductance of the OTA can be adjusted by varying R Q.
The transconductance remains constant over the input voltage range, which is the basic prerequisite for distortion-free
signal transmission in nonfeedback mode. Inserting the
feedback buffer lowers the input offset voltage to 100µV,
and the output stage then decouples the amplifier stage.
Unlike that of feedback op amps, the output impedance is
not in the mΩs, but rather between 4Ω and 10Ω depending
upon the quiescent current. It remains, however, extremely
constant over frequency. The gain can be adjusted using the
VIN
RIN
150Ω
4
13
OTA ∞
10
OB
9
ROUT
50Ω
VOUT
R1
200Ω
R2
200Ω
13
R2
200Ω
3
RIN
100Ω
FB
8
Offset Compensation
FIGURE 15. Open-Loop Amplifier Common Emitter
Circuit.
6
AGC AMPLIFIER OR AMPLITUDE MODULATOR
Near the zero point (V DS ~ mV), a FET functions like a linear
resistor controlled by V GS (rDS = dVDS/dID). As shown in
Figure 16, this property of a FET can be used to control the
amplitude of an RF signal by the DC voltage, V GS, according
to the equation in the figure. In this configuration, the
OPA622 functions as an open-loop amplifier with offset
compensation. The signal excitation can take place either at
the inverting or at the noninverting input. The only components lacking for a complete AGC circuit are a peak detector
to measure the output voltage and a control amplifier, which
can also be used to linearize the FET.
DIRECT-FEEDBACK AMPLIFIER
It has already been mentioned that a short delay time in the
feedback loop is important to provide large bandwidth. As
indicated by its name, the direct-feedback amplifier, shown
in Figure 17, uses a direct, short feedback loop. The output
signal at the OTA is transferred via R 2 to the inverting input.
Since the currents at the OTA output and the inverting input
have the same polarity, they combine to counteract the
voltage at the noninverting input. As already shown using
the OPA660 direct-feedback amplifiers can deliver bandwidths of up to 500MHz and excellent pulse behavior at
slew rates of up to 2ns.
It’s easy to convert an AGC amplifier into an amplitude
modulator. The carrier frequency is applied to the input of
the OPA622, and the FET modulates the carrier frequency
amplitude in tact with the LF signal voltage.
RIN
150Ω
VIN
RIN
150Ω
3
FB
8
COTA
1pF
4
13
OTA ∞
10
9
OB
FET
VCONTROL
RIN
150Ω
RIN
150Ω
3
FB
8
VOUT
R1
22Ω
ROG
150Ω
RF
ROUT
50Ω
Gain =
(R1 + RFET)
ROG
COTA
1pF
4
13
OTA ∞
10
OB
9
ROUT
50Ω
VOUT
R1
22Ω
ROG
150Ω
VSIGNAL
FET
FIGURE 16. AGC Amplifier and Amplitude Modulator.
RIN
150Ω
VIN
RIN
150Ω
3
FB
8
COTA
1pF
4
13
R1
82Ω
OTA ∞
10
OB
R2
200Ω
C1
5.6pF
FIGURE 17. Direct-Feedback Amplifier with Offset Compensation.
7
9
ROUT
50Ω
VOUT
VOLTAGE-FEEDBACK AMPLIFIER
WITH OUTPUT VOLTAGE LIMITATION
the comparator. The positive feedback with R 1 increases the
open-loop gain of the comparator at low frequencies, while
C1 increases the overall open-loop gain over frequency.
Some integrated circuits such as very fast ADCs react
extremely sensitively to overdrive at the input. In such cases,
overdrive protection is vital to prevent damage to the converters. Unlike circuits offering external protection for sensitive inputs, the circuit shown in Figure 18 limits the
maximum output voltage from a TV camera or production
mixer that can go into a broadcast distribution system. The
high-impedance OTA output functions linearly only up to
the positive or negative limits determined by the limitation
circuitry.
DESIGN TOOLS
To enable users to test the parameters and applications of the
OPA622 for themselves, Burr-Brown offers a completely
assembled demo board and two PSpice  models. The demo
board gives important tips for layout design, as well as
facilitating the test phase. Since almost all pulse and bandwidth parameters specified in the PDS were determined
using this demo board, it is guaranteed to deliver comprehensible results. As can be seen in Figure 20, the noninverting
variation can be tested using voltage feedback and a gain of
+2. The board uses SMA RF connectors as an interface to
the test devices. For users who wish to reassemble the pins,
Table III lists recommended component values for various
gains.
LOW JITTER COMPARATOR
Integrated circuits with good pulse behavior and a short
recovery time after overdrive make excellent comparators.
Figure 19 shows a comparator using the OPA622. This
configuration might appear unusual at first glance, but it
offers a comparator with jitter of approximately 20ps at a
signal frequency of 10MHz. The OPA622 compares the
voltages at its two inputs and changes the output according
to the difference between these input voltages. The two
antiparallel GaAs diodes limit the output voltage and keep
the OPA622 in the linear range. A configuration using R 1
and C1 produces positive feedback from the OTA output to
the positive input and works like a Schmitt trigger, accelerating the output voltage change during the reversal phase of
VIN
RIN
150Ω
13
Limiter
with
Knee
Function
T1
4
OTA ∞
Another way to sample the OPA622 is by simulation using
the PSpice circuit simulator program. The two PSpice 
macromodels for the OPA622 feature differing accuracy and
simulation times. Both models are part of a shell, which
makes it easier for the user to run the simulations and
compare simulation results. The disk also contains completely dimensioned application circuits for the most common applications, which can be displayed immediately after
installation.
10
Pos
9
OB
ROUT
50Ω
T2
R1
820Ω
Neg
ROG
10Ω
8
FB
3
R2
91Ω
FIGURE 18. Operational Amplifier with Output Voltage Limitation.
8
VOUT
C1
0.5pF to 2.5pF
R1
27kΩ
VIN
RIN
100Ω
ROUT
50Ω
4
13
OTA ∞
10
9
OB
VOUT
R1 Hysteresis
ROG
C1 Slew Rate
Optimization
FB
2 X HP2811
FIGURE 19. Comparator with Voltage Feedback.
–VCC +VCC
1pF
COTA
12
5
RQC
390Ω
–VCC
2
+VCC OUT
10
11
R5
NC
Biasing
RLR
150Ω
ZO = 50Ω In
POS
RSOURCE
= 50Ω
RL2
100Ω
RL1
100Ω
Out
9
OB
RL
50Ω
4
RING
NC
OTA
OPA622
R1
330Ω
3
FB
R3
0
ZO = 50Ω InNEG
NC
RSOURCE
= 50Ω
8
ROG
150Ω
13
–VCC OUT
R4
NC
R6
NC
6
R2
330Ω
12
R9
10Ω
+5V
11
C1
C2
470pF
10nF
C6
C5
470pF
10nF
C3
2.2µF
Gnd
2.2µF
–5V
6
5
C4
R8
10Ω
FIGURE 20. Circuit Schematic Demo and Board DEM-OPA622-1GC.
9
ZO = 50Ω
RIN
50Ω
OPA622AP
IQ = 5mA
RQC = 430Ω
OPA622AU
GCL
+1
+2
+5
+10
–1
–2
UNITS
R1
R2
R3
ROG
COTA
RLR
R4
R5
Ring
Bandwidth:
VOUT = 0.2Vp-p
VOUT = 2.8Vp-p
0
—
220
330
2.2
150
—
—
—
330
330
0
150
1
150
—
—
—
620
160
0
56
—
150
—
—
—
1,600
180
0
10
—
150
—
—
—
390
—
0
200
1
150
390
62
150
470
—
0
150
1
150
240
62
150
Ω
Ω
Ω
Ω
pF
Ω
Ω
Ω
Ω
170
220
160
200
140
170
110
110
135
150
125
150
MHz
MHz
IQ = 5mA
+1
+2
+5
+10
–1
–2
UNITS
R1
R2
R3
ROG
COTA
RLR
R4
R5
Ring
Bandwidth:
VOUT = 0.2Vp-p
VOUT = 2.8Vp-p
150
—
0
270
2.2
200
—
—
—
240
240
0
150
1
150
—
—
—
470
120
0
47
—
200
—
—
—
820
91
0
10
—
200
—
—
—
240
—
0
160
1
150
240
68
150
300
—
0
100
1
150
150
68
150
Ω
Ω
Ω
Ω
pF
Ω
Ω
Ω
Ω
200
250
170
240
160
230
100
100
180
250
175
240
MHz
MHz
TABLE III. Typical Parameters of the OPA622 as a Voltage-Feedback Amplifier.
FIGURE 21. Layout and Silkscreen of the Demo Board.
10
RQC = 430Ω
GCL
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