Load Commutated Current Source Inverter fed
Induction Motor Drive
Debmalya Banerjee, V.T.Ranganathan†
Department of Electrical Engineering
Indian Institute of Science, Bangalore, India
† Email: vtran@ee.iisc.ernet.in
Abstract— Current Source Inverter (CSI) is an attractive
solution in large power drives. The conventional GTO
based CSI fed Induction Motor drives suffer from drawbacks such as low frequency torque pulsation, harmonic
heating and unstable operation at low speed ranges. These
drawbacks can be overcome by connecting a current
controlled Voltage Source Inverter (VSI) across the motor
terminal replacing the bulky ac capacitors. The VSI
provides the harmonic currents, which results in sinusoidal
motor voltage and current even with the CSI switching at
fundamental frequency. This paper proposes a CSI fed
Induction Motor drive scheme where, GTOs are replaced
by thyristors in the CSI without any external circuit to
assist the turning OFF of the thyristors. Here, the current
controlled VSI, connected in shunt, is designed to supply
the VAR requirement of the induction motor and the
CSI is made to operate in leading power factor mode
such that the thyristors in the CSI are autosequentially
turned OFF. The resulting drive will be able to feed
even 11KV induction motors directly. A sensorless vector
controlled CSI drive based on proposed configuration is
developed. The experimental results from a 5 hp prototype
are presented. Experimental results show that the proposed
drive has stable operation through out the operating range
of speeds.
Index Terms— Three level VSI, Pulse Width Modulation,
Load Commutated CSI, Silicon Controlled Rectifier(SCR),
Induction Motors
I. I NTRODUCTION
Voltage Source Inverter (VSI) fed Induction Motor
drive is an attractive solution for low power applications
because of the availability of fast switching devices like
IGBT and MOSFET. But for medium and high power
applications, VSI can not switch as fast as in the case
of low power counterpart due to increased switching
loss. Furthermore, for medium voltage applications, the
devices of required voltage rating for two level VSI are
not readily available. Multilevel VSI [1] are used in
medium voltage applications as the device voltage stress
decreases with the increase in the number of levels of
multilevel VSI. VSI fed Induction Motor drives have
certain drawbacks as follows:
1. The voltage overshoots caused by the long cable
connected between the converter and the motor.
2. Additional losses in the motor caused by the high
frequency current ripple.
3. Electromagnetic emission caused by the high dV /dt
in the cable.
4. Common mode voltage and bearing current in the
motor.
Current Source Inverter (CSI) fed Induction Motor
drive is an attractive solution for high power applications
due to its inbuilt shoot through protection and
regenerative capability. Due to large dc link inductor
connected at the output of the phase controlled rectifier,
the dynamic response of the torque of the motor is poor
compared to that of VSI fed induction motor drive.
This drive is suitable for fan and pump type of load
where dynamic response of the torque is not of major
concern. Fig. 1 shows the schematic diagram of the
conventional CSI fed induction motor drive with output
capacitor. With self commutating devices like GTO
in the CSI, Pulse Width Modulation (PWM) of input
current can be done to get better quality output current
and voltage [2]. But for large induction motor drives
the switching frequency is low to minimize switching
losses. Bulky ac capacitors have to be employed at
the motor terminal to reduce the voltage spikes due to
switching of the currents. These capacitors form two
modes of resonance with motor magnetizing inductance
and leakage inductance. Reference [3] clearly explains
the resonance modes and proposes selective harmonic
elimination techniques to avoid them. But it is not
possible to completely avoid resonance at low values of
fundamental frequency.
Reference [4] proposes a novel CSI fed induction motor
CHOKE
CHOKE
IM
GRID
IM
GRID
GTO−INVERTER
GTO−INVERTER
CAPACITOR BANK
CAPACITOR BANK
AC CHOKE
RECTIFIER
RECTIFIER
Fig. 1. Functional block diagram of conventional CSI fed Induction
Motor drive
ACTIVE FILTER
This paper proposes a Load Commutated Current
Source Inverter (LCI) fed Induction Motor drive with
sinusoidal voltage and current. For successful commutation of the thyristors of the LCI, it has to be operated in
leading power factor mode through out its operating frequency. Conventionally, LCI is used to feed synchronous
motors, as it can be operated in leading power factor
mode by controlling the excitation of the synchronous
motor [5]. But, if LCI is employed to feed the induction
motor, the current controlled VSI, connected across the
motor terminal, has to inject reactive current to operate
the LCI in leading power factor mode. Additionally, the
VSI, filters all the harmonic frequency currents produced
by the LCI, which results in sinusoidal motor current and
Fig. 2. Functional block diagram of CSI fed Induction Motor drive
with an Active Filter connected across the motor terminal
CAPACITOR BANK
CHOKE
T1
T3
T5
T4
T6
T2
IM
GRID
RECTIFIER
SCR−INVERTER
AC CHOKE
drive in which the VSI is used as an active filter across
the motor terminals. The proposed configuration not
only overcomes most of the drawbacks of conventional
CSI drive but also ensures sinusoidal motor voltage and
current waveforms throughout the operating range of
the drive. Fig. 2 shows the functional block diagram of
the drive. The CSI is switched at fundamental frequency
with reduced switching losses, producing quasi square
wave output current. The VSI, at the motor terminal
ideally filters all the harmonic frequency currents
produced by the CSI, which results in sinusoidal motor
current and voltage. The filtering action is limited by the
bandwidth of the VSI current controller. As the VSI has
to carry only harmonic current, its VA rating is around
30% of the CSI VA rating. So, it can be switched at
multiples of kHz to get reasonable bandwidth of the
current controller. Due to limited bandwidth of the
VSI, output capacitors can not be totally eliminated. A
small capacitor is connected across the CSI terminal to
filter out high switching frequency harmonic component
of the CSI current which is not compensated by the
VSI. Additionally, it absorbs switching frequency
component of VSI current. It results in good quality
of motor voltage. Although GTO with complex gate
drive, is used in the CSI, PWM is avoided to reduce the
switching losses in this scheme.
ACTIVE FILTER
Fig. 3. Functional block diagram of the proposed Load-Commutated
CSI fed Induction Motor drive
voltage, as in reference [4]. Fig. 3 shows the functional
block diagram of the proposed drive scheme.
The proposed LCI drive configuration is explained in
Section II. A theoretical background of the control
system of the proposed drive configuration is presented
in Section III. The proposed drive is implemented on
an experimental prototype. The experimental results are
presented in Section IV. From these results it can be
concluded that thyristors can be used in the LCI feeding
an Induction Motor and the proposed drive has superior
performance over conventional CSI drives in terms of
improved motor current and voltage waveforms.
II. D ESCRIPTION OF L OAD C OMMUTATED CSI
I NDUCTION M OTOR DRIVE
FED
RECTIFIER in the Fig. 3 is a three phase, line
commutated, phase controlled rectifier using thyristors.
It is used in conjunction with a DC link inductor to
function as a constant current source. A current loop
can be designed to produce a regulated current out of
the converter by adjusting the output voltage with firing
angle control.
SCR-INVERTER in the Fig. 3 is a current fed,
load commutated inverter using thyristor. It is a simple
three phase thyristor bridge. The capacitors and the
diodes present in Auto Sequentially Commutated
Inverter (ASCI), are not required in this CSI. It acts
as the main converter and supplies the active power
required by the induction motor and associated load.
ACTIVE FILTER in the Fig. 3 is a current controlled,
voltage fed, three level, neutral point clamped inverter
based on IGBT. It acts as an active filter whose function
is to inject the required reactive and harmonic currents,
to the motor terminal. The dc voltage source is available
from a diode bridge and capacitor filter arrangement,
connected to the grid.
The following two constraints decide the reactive and
harmonic current requirement from the active filter
respectively.
1. The thyristor based CSI, should see the combined
load (Induction motor and active filter) as a leading
power factor load. The fundamental component of
CSI output current should lead the motor terminal
phase-neutral voltage, through out the whole range of
operating frequency, by a specific time, tc which must
be greater than the turn off time, tq of the thyristors,
for successful commutation of the thyristors.
2. The induction motor must be fed with sinusoidal
current to reduce the low frequency torque pulsation
and harmonic heating. As a result, the motor voltage
will be sinusoidal in nature and the problems associated
with the operation from PWM VSI will be eliminated.
As the thyristor based LCI is working in six step mode
(1200 conduction), the residual harmonic currents must
be compensated by the active filter.
A small capacitor bank, connected at the output
of the CSI, absorbs the rapid change in CSI output
current, which occurs on switching, and also provides
a low impedance path for high current harmonics.
It helps in the commutation process as follows: the
motor terminal voltage reverse biases the outgoing
thyristor, instantaneously turning it off, unlike the
conventional LCI fed synchronous motor drive where
the motor back emf and subtransient reactance decide
the commutation time. In the present case, current
transfer from outgoing thyristor to incoming thyristor is
decided by the stray inductance between thyristor and
capacitor terminal. It should be noted that, harmonic
and reactive compensation is not the objective of the
small capacitor bank. Hence, the capacitance value
(0.05 to 0.1 p.u.) of the bank is designed accordingly. It
absorbs the switching frequency component of the VSI
current also.
The three level VSI (active filter) is connected to the
motor terminal with inductors in series. These inductors
are necessary for the current control of the active filter.
The value of the inductors decides the switching frequency ripple in the currents injected to the main system.
A. Advantages of the proposed system
The proposed system eliminates the major
disadvantages of the thyristor based conventional
Auto Sequentially Commutated Inverter (ASCI) fed
induction motor drive and GTO-CSI fed induction
motor drive.
1. Low frequency torque pulsation is totally eliminated,
although the CSI is working in six step mode.
2. The commutation process of the thyristors in the CSI
does not depend on the load.
3. The motor experiences sinusoidal voltage, which
solves all of the problems associated with PWM
voltage fed adjustable speed motor drive like insulation
failure, bearing currents, common mode voltages and
overvoltages due to long cable.
4. The system does not require bulky capacitors at the
output of CSI, as was required in GTO based CSI fed
induction motor drive.
III. T HEORETICAL BACKGROUND OF THE CONTROL
SYSTEM OF THE PROPOSED DRIVE SCHEME
Broadly, the control system of the total drive, has
two main objectives.
1. Speed and torque control of the induction motor,
keeping the motor flux constant i.e. independent control
of motor flux and torque.
2. Control of harmonic current and reactive current
injection by the active filter.
The control of motor torque and flux is incorporated in
a synchronously rotating reference frame (d-q frame),
whose direct axis (d-axis) is aligned with rotor flux
~r and the quadrature axis (q-axis)
linkage space vector, Ψ
is leading the d-axis by 90o .
The current control of the active filter is incorporated
in another synchronously rotating reference frame (γ -δ
frame), whose quadrature axis (δ -axis) is aligned to the
motor terminal voltage space vector, V~s and the direct
axis (γ -axis) lags the δ -axis by 90o .
δ − axis
q − axis
q − axis
δ − axis
~s
V
~s
V
ω1
~s
E
~s E
~r
E
ω1
I~s
I~r
~m
Ψ
I~m
~m
Ψ
I~m
~r
Ψ
I~r
d − axis
~r
Ψ
γ − axis
d − axis
I~s
γ − axis
~r
E
Fig. 4. Phasor Diagram of Induction Machine in motoring operation
From the steady-state phasor diagrams of the induction
machine (see Fig. 4 and Fig. 5), it can be appreciated
that, (d-q) and (γ -δ ) axes are rotating at synchronous
speed, ω1 . They make a constant angle, µ between
them, as shown in Fig. 6 and Fig. 7.
It can be seen that, in the case of motoring operation
of the induction machine, the (γ -δ ) axes lead the (d-q)
axes (see Fig. 6) with a positive angle, µ between them,
but in the case of generating operation, the (d-q) axes
lead the (γ -δ ) axes (see Fig. 7) with a negative angle,
µ between them.
The component of the stator current space vector,
I~s on the d-axis, isd is responsible for rotor flux in
the machine, and the component along q-axis, isq is
responsible for torque production in the machine. If the
same stator current space vector, I~s is projected in the
(γ -δ ) reference frame, the component of I~s along δ -axis,
isδ is responsible for active power consumption by the
machine and the associated load, while the component
along the γ -axis, isγ is the reactive current responsible
for the flux production of the machine. This reactive
component is actually responsible for the lagging power
factor operation of the machine.
The active filter has to draw a leading fundamental
frequency current, I~c such that, the fundamental
component of the CSI output current, I~ac leads the
motor terminal voltage, V~s .
I~c = I~ac − I~s
(1)
Fig. 5.
operation
Phasor Diagram of Induction Machine in generating
b − axis
q − axis
δ − axis
γ − axis
ω1
V~s
µ(t)
θ(t)
Ψ~r
d − axis
ρ(t)
ω1
a − axis
Fig. 6. Vector Diagram showing two rotating reference frames in
motoring mode
b − axis
δ − axis
q − axis
d − axis
V~s
ω1
~r
Ψ
µ(t)
γ − axis
ρ(t)
θ(t)
ω1
a − axis
Fig. 7. Vector Diagram showing two rotating reference frames in
generating mode
I~ac
vs1
iac1
is1
ωt
0
φ
β
I~c
vs12
I~c
β
Fig. 8.
I~s
I~ac
~s
V
φ
Commutation of R-phase top device in motoring mode
δ − axis
I~s
This condition should be met, through out the whole
range of operating frequency of the drive, for successful
commutation of the thyristors in the CSI.
During speed reversal and speed reduction process, the
drive will go to the generating mode transiently. The
control system should be able to inject proper reactive
current such that, the thyristors successfully commutate,
while the drive is working in generating mode.
To illustrate the requirement of leading power factor
operation of the CSI, for successful commutation of
the thyristors, R-phase motor terminal voltage (phaseneutral), vs1 , R-phase CSI output current, iac1 , line-line
voltage, vs12 , and R-phase motor current, is1 have been
shown for both motoring and generating mode.
Fig. 8 shows the 50Hz steady state waveforms in
motoring mode of operation. Motor terminal voltage
(phase-neutral), vs1 is assumed to be sinusoidal. The
R-phase motor current, is1 is lagging vs1 by the power
factor angle, φ. The R-phase CSI output current, iac1
is drawn such that, its fundamental component leads
vs1 by an angle β . At the instant, when Y-phase top
group thyristor, is turned ON, the line voltage vs12 is
positive or the voltage across the outgoing thyristor, T1
is negative for the time interval, (β/ω1 ), where ω1 is
the operating frequency of the drive and β is chosen
such that (β/ω1 ) should be greater than the device turn
OFF time, tq for any ω1 .
Fig. 9 shows the relative positions of motor current, I~s ,
CSI output current, I~ac , and active filter current, I~c in
γ -δ coordinate system for motoring mode of operation.
I~s lags the δ -axis by the power factor angle, φ and I~ac
leads the δ -axis by the angle of commutation margin, β .
From Eqn 1,
(2)
Equating real and imaginary part of the above equation,
icγ = iacγ − isγ
Fig. 9.
mode
Phasor Diagram showing load commutation in motoring
icδ = iacδ − isδ
(3)
(4)
As, the active power demanded by the motor, has to be
supplied by the CSI, the component of I~s on δ -axis, isδ
must be equal to the component of I~ac , on δ -axis, iacδ .
iacδ = isδ
(5)
From Eqn. 4 and Eqn. 5 it can be seen that,
icδ = 0
(6)
which implies, there is no active power demand from the
active filter.
Eqn. 3 quantifies the reactive current requirement from
the active filter, where iacγ is the reactive component of
I~ac . As, I~ac leads V~s , iacγ is expected to be negative.
iacγ = |I~ac |.sin(−β)
(7)
where, |I~ac | is the magnitude of the CSI output current
space vector, I~ac and it can be found out from the
following equation.
|I~ac | = |iacδ |/cosβ
(8)
From Eqn. 5 and Eqn. 8,
|I~ac | = |isδ |/cosβ
(icγ + jicδ ) = (iacγ + jiacδ ) − (isγ + jisδ )
(icγ + jicδ ) = (iacγ − isγ ) + j(iacδ − isδ )
γ − axis
(9)
From Eqn 7 and Eqn 9 iacγ can be expressed as,
iacγ = −|isδ |.tanβ
(10)
So, the active filter has to provide the reactive current
requirement of the motor, isγ along with the reactive
I~ac
vs1
I~c
ωt
0
iac1
φ
is1
I~s
current required by the CSI for successful commutation,
iacγ .
From Eqn. 3 and Eqn. 9
(11)
From Eqn. 6 and Eqn. 11, the active filter current, I~c
can be found out, in the folowing way,
I~c = −(|isδ |.tanβ + isγ )
~s
V
φ
Commutation of R-phase top device in generating mode
icγ = −(|isδ |.tanβ + isγ )
I~c
I~ac
β
vs12
Fig. 10.
I~s
(12)
From Eqn. 12 it is clear that, to determine the active filter
current, the stator current expressed in γ -δ axis and the
lead angle, β are essential.
Fig. 10 shows the 50Hz steady state waveforms in
generating mode of operation. Machine terminal voltage
(phase-neutral), vs1 is assumed to be sinusoidal. The Rphase machine current, is1 is lagging vs1 by the power
factor angle, (π − φ). The R-phase CSI output current,
iac1 is drawn such that, its fundamental component lags
vs1 by an angle (π + β). At the instant, when Y-phase
top group thyristor, is turned ON, the line voltage vs12
is positive or the voltage across the outgoing thyristor is
negative for the time interval, (β/ω1 ), where ω1 is the
operating frequency of the drive and β is chosen such
that (β/ω1 ) should be greater than the turn OFF time,
tq for any ω1 .
Fig. 11 shows the relative position of motor current, I~s ,
CSI output current, I~ac , and active filter current, I~c in γ -δ
coordinate system for generating mode of operation. I~s
lags the δ -axis by an angle of, (π −φ) and I~ac lags the δ axis by the angle of, (π + β). φ is the power factor angle
and β is the angle of commutation margin. The active
component of the stator current, isδ is out of phase with
terminal voltage, V~s , whereas, the reactive component,
isγ lags V~s by π/2 radian. From the phasor diagram
(Fig. 11), the reactive current requirement by the active
filter can be found out in the following way,
δ − axis
(π + β)
γ − axis
Fig. 11.
mode
Phasor Diagram showing load commutation in generating
Recalling Eqn. 1,
(icγ + jicδ ) = (iacγ + jiacδ ) − (isγ + jisδ )
(icγ + jicδ ) = (iacγ − isγ ) + j(iacδ − isδ )
(13)
From the above equation, Eqn. 3 and Eqn. 4 can be
established. Following the same way, as for the motoring
mode, Eqn. 12 can also be established for generating
mode of operation.
I~c = −(|isδ |.tanβ + isγ )
(14)
It can be noticed that the above control equation
is unique irrespective of the operating mode of the
machine (motoring or generating).
IV. E XPERIMENTAL V ERIFICATION
The proposed drive is implemented on an experimental prototype.A six-pulse three-phase fully controlled
SCR ac to dc converter is used on the line side. An
IGBT based three-phase three-level diode clamp inverter, designed and fabricated in the laboratory is used
as active filter inverter. A three-phase thyristor bridge
similar to the input phase controlled rectifier has been
used as SCR based CSI. A digital controller based on
TMS320LF2407A DSP, designed and developed in the
laboratory, has been used to implement the total control
system (as shown in Fig. 12). It is employed to generate
the PWM for the active filter, firing pulses for the SCR
based CSI, control angle for the input phase controlled
rectifier. The schematic diagram of the experimental setup is shown in Fig. 13. A 400 Volts, delta connected,
8A, 3.78kW, 1425 rpm squirrel cage induction motor
Machine
Converter
Line
Converter
Grid
Motor
Filter
Capacitor
DC−Link
Inductor
CSI
Output
Current
Motor
Voltage
IM
(iac1 , iac2 )
DC
Link
Current
Active
Filter
[G1 − G6 ]
θ
CSI
Gating
Estimator ρ
(is1 , is2 )
ωr
imr
∗
∗
(i∗
ac1 ,iac2 ,iac3 )
vdci
DC
Link
Current
Control
(vs12 , vs23 )
(ic1 , ic2 ) Current
DC
Link
Voltage
of CSI
αcont
Motor
Current
i∗
dc
idc
[SR1 − SB2 ]
Active
Filter
IP−SPWM
of
3−level
VSI
i∗
sq
∗
∗
∗
(vc1
,vc2
,vc3
)
Active
Filter
Current
Control
Current
Speed
Control
Reference
∗
(i∗
cγ ,icδ )
Generation
i∗
sd
ρ
β
∗
ωr∗
Flux
Control
i∗
mr
θ
Fig. 12.
Block diagram of the control structure of the drive scheme
RECTIFIER
SCR−CSI
iac1
idc
vs12
iac2
vdci
GRID
is1
vs23
G1
ic1
G6
ic2
vref
Digital Controller
SR1
SB2
ACTIVE FILTER
Fig. 13.
Block diagram of the Experimental Setup
is2
IM
has been used as the test motor. A DC machine coupled
to the induction motor, is used as a load machine. The
rating of the DC machine is 230 Volts, 13A, 3KW, 1475
rpm.
A. Experimental Results
An experiment has been carried out to run the motor
at different speeds from 10 Hz to 50 Hz of fundamental frequency to test the steady state performance of
the drive. The objective of the experiment was to test
the commutation of the thyristors in the CSI and the
waveform quality of motor current and voltage. Fig. 14
shows the steady state waveforms of R-phase CSI current
(iac1 ), R-phase VSI current (ic1 ), R-phase motor current
(is1 ) and motor line-line voltage between R and Y phase
(vs12 ) at 50Hz in motoring mode. Fig. 15 shows the
waveforms of Y-phase motor current (is2 ), Y-phase VSI
current (ic2 ), Y-phase CSI current (iac2 ) and motor lineline voltage between Y and B phase (vs23 )at 25Hz steady
state in generating mode. Fig. 16 shows the waveforms
of line-line grid voltage between R and Y phase (vi12 ),
R-phase phase controlled rectifier input current (iR ), Yphase CSI current (iac2 ) and motor line-line voltage
between Y and B phase (vs23 )at 25Hz steady state in
generating mode. The phase relation between vi12 and iR
clearly shows that the drive is operating in regenerating
mode.
Fig. 14. Steady state performance of the drive at 50Hz in motoring
mode Ch:1 iac1 (10A/div.); Ch:2 ic1 (10A/div.); Ch:3 is1 (10A/div.);
Ch:4 vs12 (660V/div.)
Fig. 15. Steady state performance of the drive at 25Hz in regenerating mode Ch:1 is2 (10A/div.); Ch:2 ic2 (10A/div.); Ch:3 iac2
(5A/div.); Ch:4 vs23 (330V/div.)
Fig. 16. Steady state performance of the drive at 25Hz in regenerating mode showing line side current and voltage Ch:1 vi12 (500V/div.);
Ch:2 iR (5A/div.); Ch:3 iac2 (5A/div.); Ch:4 vs23 (330V/div.)
analyzer is used to measure the total RMS of motor current Irms and motor voltage Vrms at different fundamental frequency and load condition. For each fundamental
frequency, the motor voltage and current waveforms are
captured in a digital CRO. 1000 data points are stored
for 50 ms time window. From the stored datapoints,
total RMS (Irmscal , Vrmscal ) and RMS of fundamental
component (I1rmscal , V1rmscal ) are calculated for one
fundamental cycle. From Table I and Table II, total RMS
measured from power analyzer, matches closely to that
calculated from stored data from CRO. Total Harmonic
Distortion (THD) is calculated as,
q
B. Experimental measurement of harmonic distortion
As the active filter has a limited bandwidth depending
on the switching frequency, it is not possible to get
purely sinusoidal voltage and current waveform of the
induction motor. From the experimental setup, motor
voltage and currents have been measured to get a quantitative understanding of their waveform quality. A power
T HDis1 =
2
2
Irmscal
− I1rmscal
I1rmscal
q
T HDvs12 =
2
2
Vrmscal
− V1rmscal
V1rmscal
(15)
(16)
TABLE I
M EASUREMENT OF THD OF R- PHASE MOTOR CURRENT, is1
Irms
(Amps)
5.4
6.8
7.6
7.1
7.8
F
(Hz)
26.92
33.2
37.7
42.09
48.193
Irmscal
(Amps)
5.4019
6.8814
7.66
7.1091
7.6677
I1rmscal
(Amps)
5.3829
6.8645
7.6534
7.0705
7.6508
TABLE IV
T HDis1
M EASUREMENT OF MOTOR EFFICIENCY, ηmot
0.0839
0.0723
0.0442
0.1047
0.0665
Pmot
(KW)
3.81
3.56
TABLE II
M EASUREMENT OF THD OF MOTOR LINE - LINE VOLTAGE , vs12
Vrms
(Volts)
216.5
256.1
297.9
326.2
368.5
F
(Hz)
26.9
31.467
36.864
42.467
46.85
Vrmscal
(Volts)
214.3538
256.568
300.4013
328.8813
368.948
V1rmscal
(Volts)
212.8848
254.993
298.8040
326.812
366.245
T HDvs12
0.1177
0.1113
0.1035
0.1108
0.1217
Vload Iload
Pin
(17)
From the measurement shown in the Table III, ηsystem
TABLE III
M EASUREMENT OF SYSTEM EFFICIENCY, ηsystem
Qin
(KVAR)
3.52
3.4
Sin
(KVA)
5.14
4.85
Vload
(Volts)
180
170
Iload
(Amps)
13.5
13.5
comes out as 0.65. Similarly, the efficiency ηmot of
the induction motor coupled with dc generator can be
determined as follows,
ηmot =
Vload Iload
Pmot
Smot
(KVA)
4.56
4.29
Vload
(Volts)
180
187
Iload
(Amps)
13.5
14
drawn by the induction motor is measured at 40Hz of
fundamental frequency of operation. Output power is
measured by measuring armature voltage Vload and load
current Iload of the dc generator. From the measurement
shown in the Table IV, ηmot comes out as 0.7. System
efficiency ηsystem is the product of converter efficiency
ηconv and motor efficiency ηmot .
(19)
From the above equation, the converter efficiency ηconv
can be determined to be around 0.9.
The total system efficiency is measured in order to
understand the loss distribution in converters and motor.
The motor is electrically loaded by connecting a variable
resistor across the armature of the separately excited
DC generator coupled to the induction motor. Active
power Pin , reactive power Qin and VA Sin drawn from
the grid is measured at 40Hz of fundamental frequency
of operation. Output power is measured by measuring
armature voltage Vload and load current Iload of the dc
generator. The total system efficiency ηsystem can be
calculated as follows,
Pin
(KW)
3.75
3.45
Qmot
(KVAR)
2.52
2.4
ηsystem = ηconv ∗ ηmot
C. Measurement of system efficiency
ηsystem =
Active power Pmot , reactive power Qmot and VA Smot
(18)
V. C ONCLUSION
The proposed drive scheme is implemented in an
experimental prototype. The experimental results show
the feasibility of the control strategy. This type of drive
scheme is suitable for large power (multiples of MW)
applications. For high power induction motors, the full
load power factor generally is above 0.9. The reactive
VAR requirement falls below 30%. As the VSI has to
inject reactive and harmonic component current, its KVA
rating is around 40% of the CSI KVA rating. So, for
large power applications also, IGBT based VSI can be
used switching at multiples of kHz. There are harmonics
generated on the line side due to the use of three
phase thyristor controlled bridge as shown in Fig. 16.
This is not acceptable for systems of medium and high
power rating. Standard solutions like passive and shunt
active filter can be used in the future systems to ensure
sinusoidal source current drawn at unity power factor
irrespective of the speed of the drive. The experimental
results show that the proposed drive can achieve almost
sinusoidal voltage and current of the induction motor.
R EFERENCES
[1] A.R. Beig, “Application of three level voltage source inverters to
voltage fed and current fed high power induction motor drives”
-Ph.D. Thesis of Electrical Engineering, IISc, Bangalore,April
2004.
[2] P.M. Espelage, L.M. Nowak, and L.H. Walker, “Symmetrical
GTO current source inverter for wide speed range control of
2300 to 4160 Volt, 350 to 7000 Hp, induction motors,”-Proc.
IEEE Ind. Appl. Soc. Annu. Meeting, Pittsburgh, Oct. 27, 1988,
vol. 1, pp. 302307..
[3] B. Wu, S. B. Dewan, and G. R. Slemon, “PWM CSI inverter
for induction motor drives, IEEE Trans. Ind. Appl., vol. 28, no.
1, pp. 6471, Jan./Feb. 1992.
[4] A.R. Beig, and V.T. Ranganathan, “A novel CSI-fed Induction
Motor Drive,” IEEE Trans. on Power Electronics, vol. 21, no. 4,
July 2006
[5] H.Stemmler, “High-power industrial drives,” Proc. IEEE, Vol.
82(8), 1994, pp. 1266-1286.