3. SIMULATION
To prove the above analysis, the conventional and proposed power
dividers were designed at 2.0 GHz with Z1 ⫽ 50#⍀ and
Z2 ⫽ 25#⍀. Table 1 indicates that new structure has lower inductance, reducing the effect of self resonance.
Figure 4 compares the simulated S-parameters of the conventional and proposed power divider. On the basis of ⫺15 dB, the
new power divider provides wider bandwidth by 25% in input
return loss (S11) and by 70% in the isolation (S23) than the
conventional one. The insertion loss (S21) of the proposed divider
rolls off more softly at low frequency end as shown in Figure 4(c).
4. MEASUREMENT
The proposed power divider was fabricated on Teflon substrate
using 1.6 ⫻ 0.8 mm2 chip elements. Figure 5 shows the fabricated
power divider. Since this structure has a termination impedance of
25 ⍀ at ports 2 and 3, quarter-wave long lines with characteristic
impedance of Z0 ⫽ 冑Z1 䡠 Z2 ⫽ 35.4⍀ are placed at the output
ports 2 and 3 for a 50 ⍀-based measurement. To minimize the
parasitic inductance of interconnection and via-hole, the shunt
inductor L1 was inserted into the via-hole and directly grounded.
The measurement results are given in Figure 6. The center
frequency shifts to 1.9 GHz from the targeted 2.0 GHz, probably
due to the manufacturing errors and parasitic effect of lumped chip
elements and interconnections. On the basis of ⫺15 dB, the input
return loss (S11), output return loss (S22) and isolation performance
(S23) shows the fractional bandwidth of 13.8, 19.9, and 36.5%,
respectively. The insertion loss (S21) is less than ⫺3.3 dB at the
center frequency with broadband performance.
5. CONCLUSIONS
In this article, a new impedance-transforming power divider was
proposed using lumped elements only. It was designed to have a
low Q-factor during the impedance transformation to achieve wide
bandwidth. The comparative study proved that the new power
divider can achieve wider bandwidth than the conventional one in
terms of the return loss, insertion loss and isolation performance.
Therefore, the proposed power divider can be applied for advanced
microwave systems with compact size requiring the impedancetransforming characteristics such as balanced power amplifiers.
Figure 6
Measured S-parameters of the fabricated power divider
ACKNOWLEDGMENTS
The present research has been conducted by the research grant of
Kwangwoon University in 2008.
REFERENCES
1. D.M. Pozar, Microwave engineering, 3rd ed., Wiley, NJ, 2005.
2. R.K. Gupta, S.E. Anderson, and W.J. Getsinger, Impedance-transforming 3-dB 90° hybrids, IEEE Trans Microwave Theory Tech MTT-35
(1987), 1303–1307.
3. M. Dydyk, Impedance transforming three-port power divider/combiner
using lumped elements, U.S. Patent 5,469,129, November 21, 1995.
4. H.S. Nagi, Miniature lumped element 180° Wilkinson divider, IEEE
MTT-S Dig, Philadelphia, PA (2003), 55–58.
5. L.-H. Lu, X-band and K-band lumped Wilkinson power dividers with a
micromachined technology, IEEE MTT-S Dig (2000), 287–290.
6. I.D. Robertson and S. Lucyszyn, RFIC and MMIC design and technology, The Institution of Electrical Engineers, United Kingdom, 2001.
7. P. Vizmuller and R.F. Design guide systems, circuits, and equations,
Artech House, Norwood, MA, 1995.
© 2009 Wiley Periodicals, Inc.
PRECISE FREQUENCY AND
BANDWIDTH CONTROL OF
MICROSTRIP SWITCHABLE BANDSTOP
FILTERS
Zabdiel Brito-Brito, Ignacio Llamas-Garro, and Lluis Pradell
Department of Signal Theory and Communications, Technical
University of Catalonia, 08034 Barcelona, Spain; Corresponding
author: llamasi@ieee.org
Received 16 February 2009
Figure 5 Fabricated impedance-transforming power divider. Two quarter-wave long microstrip lines are placed at ports 2 and 3 to transform the
impedance from 25 back to 50 ⍀ for a 50-based measurement. [Color
figure can be viewed in the online issue, which is available at www.
interscience.wiley.com]
DOI 10.1002/mop
ABSTRACT: In this article two switchable bandstop filters able to
switch precisely between two different center frequency and bandwidth
states are presented. The filter topologies allow precise control over the
design parameters frequency and bandwidth, achieved by choosing adequate resonator sections switched by PIN diodes. Center frequency control was obtained by modifying resonator length. Bandwidth control was
achieved by choosing a resonator width and controlling the normalized
reactance slope parameter of the decoupling resonators by means of
switchable resonator extensions. Both filters were designed to have a
32.5% center frequency tuning range at L band between a low and high
filter center frequency. One topology focuses on producing an 8% fractional stopband bandwidth for both filter states. The second filter topol-
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 11, November 2009
2573
ogy focuses on producing 7 and 9% fractional stopband bandwidths for
the low and high filter center frequency states, respectively. Measured
results show a very good agreement with the simulations, for the first
topology a 32.2% measured center frequency tuning range was obtained
with 8.1 and 8.75% fractional stopband bandwidths for the low and
high filter center frequency states, respectively. The second filter topology showed a 31.2% measured center frequency tuning range with 6.94
and 9.66% fractional stopband bandwidths for the low and high filter
center frequency states, respectively. © 2009 Wiley Periodicals, Inc.
Microwave Opt Technol Lett 51: 2573–2578, 2009; Published online in
Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.
24679
Key words: bandstop filter; reconfigurable filter; stopband bandwidth;
PIN diode
1. INTRODUCTION
Switchable filters are an important part in communication systems
that allow selecting different bands of operation. Reconfigurable
bandstop filters can be used in adaptable image rejection receivers
or to tune out interfering signals in a microwave instrument. Filter
parameters like center frequency or bandwidth can be controlled
continuously, discretely, or a combination of both. Continuously
tuned filters have been implemented using varactor diodes [1–11],
MEMS varactors [12–18], or ferroelectric materials [19 –22]. On
the other hand, discretely tuned filters have been implemented
using PIN diodes [23–29] or MEMS switches [30 –34]. Considering only reconfigurable bandstop filters, most of the designs reconfigure filter center frequency and do not include fractional
stopband bandwidth control for each filter state in the design [1, 2,
10 –13, 31, 32]. In [35], a bandstop filter combining continuous
and discrete tuning using varactor and PIN diodes is presented.
In this article, we demonstrate two filter topologies able to tune
filter center frequency with an independently defined fractional
stopband bandwidth for each filter state. Although the first topology was destined to produce a fixed fractional stopband bandwidth
for both filter center frequency states [36], the second, new filter
topology is able to produce a different fractional stopband bandwidth for each filter state. In addition to the design procedure of
[36], a loss analysis has been done for both topologies to show the
effect of PIN diode contact resistance on filter insertion loss.
Both filters in this article were designed to have a 32.5% center
frequency tuning range; in the first topology, the fractional stopband bandwidth for both filter states is fixed to 8%, while the
second filter has a 7 and 9% fractional stopband bandwidth for the
filter low and high center frequency states, respectively. It can be
noted that by varying the filter design parameters exposed in this
article, other fractional stopband bandwidths and center frequencies can be obtained for switchable microstrip bandstop filters.
Figure 1 Switchable bandstop filter with center frequency control and
fixed stopband bandwidth
2574
Figure 2 Normalized reactance slope parameter for different resonator
spacing s from the main transmission line. [Color figure can be viewed in
the online issue, which is available at www.interscience.wiley.com]
2. SWITCHABLE BANDSTOP FILTER WITH CENTER
FREQUENCY CONTROL AND FIXED STOPBAND
BANDWIDTH
The design of narrow bandstop filters can be based on the reactance slope parameter of individual resonators normalized to the
characteristic impedance of a main transmission line x/Z0, where
the resonators are connected by immittance inverters as discussed
in [37].
Figure 1 shows a two pole switchable bandstop filter topology
using four PIN diodes. Large fractional stopband bandwidths require small x/Z0 values, which results in small distances between
the main line and the resonators. The normalized reactance slope
parameter x/Z0 can also be adjusted by varying resonator width W;
small values of resonator width result in small x/Z0 values for a
given resonator spacing s from the main line (see Fig. 1). When
reducing resonator width to adjust x/Z0 values, the designer must
ensure appropriate resonator unloaded quality factors for a given
filter design, especially when dealing with narrow stopband filters.
For the filter topology in Figure 1, all PIN diodes are forward
biased to produce the filter low center frequency state, and all PIN
diodes when reverse biased produce the filter high center fre-
Figure 3 Photograph of the switchable bandstop filter with center frequency control and fixed stopband bandwidth. [Color figure can be viewed
in the online issue, which is available at www.interscience.wiley.com]
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 11, November 2009
DOI 10.1002/mop
Figure 4 Simulated and measured results for the switchable bandstop
filter with center frequency control and fixed stopband bandwidth. [Color
figure can be viewed in the online issue, which is available at www.
interscience.wiley.com]
quency state. The filter provides a 32.5% center frequency tuning
range.
The relation between the normalized reactance slope parameter
that defines filter fractional stopband bandwidth and the spacing s
between the main transmission line and both resonators is shown
in Figure 2. It is apparent from Figure 2 that the length of the
switchable resonator extensions A and B in Figure 1 have been
chosen to produce two center frequency states with a fixed fractional stopband bandwidth.
The resonator extensions A in Figure 1 are carefully chosen to
produce the same normalized reactance slope parameter x/Z0 for
both, high frequency and low frequency resonators, to assure a
fixed fractional stopband bandwidth. The resonator extensions B
are added up to the resonators to produce a center frequency tuning
range of 32.5% between the filter low and high center frequency
states. The fractional stopband bandwidth for the design is fixed to
8% for both filter center frequency states. HPND-4028 Avago
Technologies beam lead PIN diodes were used for the filter. The
RF chokes used to supply DC bias to the diodes were 82 nF
inductors from Tyco Electronics with a self resonance at 1.7 GHz.
1 K⍀ limiting resistors were used to fix a 10 mA current through
the diodes. The filter was fabricated using conventional photolithographic techniques on a Rogers RO4003C substrate. A photograph
of the fabricated filter is shown in Figure 3.
The measured and simulated [38] results for the filter low and
high center frequency states are shown in Figure 4. Table 1
contains a comparison between the simulated [38] and measured
results, where a good agreement between theory and experiment in
terms of filter center frequency and stopband bandwidth was
obtained for both the filter states.
Figure 5 Simulated results for different values of PIN diode contact
resistance for the switchable bandstop filter with center frequency control
and fixed stopband bandwidth; r1 ⫽ 1 ⍀, r2 ⫽ 2 ⍀, r3 ⫽ 3 ⍀, and r4 ⫽ 4
⍀. [Color figure can be viewed in the online issue, which is available at
www.interscience.wiley.com]
Figure 5 shows the simulated [38] results considering the effect
of different PIN diode contact resistances r on the switchable
bandstop filter with center frequency control and fixed stopband
bandwidth. The simulation includes choke inductors to supply DC
bias to the diodes [39]. The lumped element models for the PIN
diodes used in simulations are similar to the ones exposed in [40].
The simulations in Figure 5 were done using diode contact resistances r1 ⫽ 1 ⍀, r2 ⫽ 2 ⍀, r3 ⫽ 3 ⍀, and r4 ⫽ 4 ⍀. It is apparent
from Figure 5 that these values of PIN diode contact resistance
have negligible effect on filter performance.
3. SWITCHABLE BANDSTOP FILTER WITH CENTER
FREQUENCY CONTROL AND VARIABLE STOPBAND
BANDWIDTH
Figure 6 shows a two pole switchable bandstop filter using six PIN
diodes able to produce two filter center frequencies with a 32.5%
center frequency tuning range. The filter provides two different
fractional stopband bandwidths for each filter center frequency
state. In this filter topology, D1 and D2 are reverse biased, whereas
D3, D4, D5, and D6 are forward biased to produce the filter low
center frequency state with a 7% fractional stopband bandwidth.
For the filter, high center frequency state D1 and D2 are forward
biased, whereas D3, D4, D5, and D6 are reverse biased resulting in
a 9% fractional stopband bandwidth.
The length of the switchable resonator extensions C, D, and E
in Figure 6 are chosen to produce two center frequency states with
a different stopband bandwidth for each filter state. First the length
of the high frequency resonator is set to produce a resonance at the
TABLE 1 Simulated and Measured Results for the
Switchable Bandstop Filter with Center Frequency Control
and Fixed Stopband Bandwidth
Simulated
Measured
Stopband Bandwidth
Center Frequency
Tuning Range
(%)
Low Frequency
State (%)
High Frequency
State (%)
32.6
32.2
8.5
8.1
8.08
8.75
DOI 10.1002/mop
Figure 6 Switchable bandstop filter with center frequency control and
variable stopband bandwidth
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 11, November 2009
2575
Figure 7 Low frequency resonator coupled to the main transmission line
filter high center frequency state. Then the high frequency resonator in Figure 6 is bent to assure an adequate normalized reactance slope parameter x/Z0 between the main line and the resonator
to produce the 7% fractional stopband bandwidth. Diodes D1 and
D2 disconnect the switchable resonator extensions C of the high
frequency resonators from the coupled central sections to avoid
unwanted resonances near the band of interest when the filter
operates in the low center frequency state. To set the filter low
center frequency response, the length of the low frequency resonator in Figure 7 is chosen to produce a resonance at the desired
filter low center frequency state. The length, width, and angle ⍜ of
the resonator extension D in Figure 7 is chosen to produce the
required x/Z0 value to produce a 9% fractional stopband bandwidth.
The calculated effect [38] when varying the angle ⍜ in Figure
7 to fine adjust x/Z0 to produce the 9% stopband bandwidth is
shown in Figure 8, where x/Z0 values increase as ⍜ increases, thus
reducing the filter stopband bandwith, and x/Z0 values decrease as
⍜ decreases, increasing the filter stopband bandwidth.
Plots of the normalized reactance slope parameter versus the
spacing s between the main transmission line and both resonators
for the filter with center frequency control and variable stopband
bandwidth of Figure 6 are shown in Figure 9.
The surface mount components, substrate, and fabrication techniques used to produce this filter were the same as the ones used
for the filter discussed in Section 2. A photograph of the fabricated
filter is shown in Figure 10.
Figure 8 Normalized reactance slope parameter for different ⍜ values.
[Color figure can be viewed in the online issue, which is available at
www.interscience.wiley.com]
2576
Figure 9 Normalized reactance slope parameter for different resonator
spacing s from the main transmission line. [Color figure can be viewed in
the online issue, which is available at www.interscience.wiley.com]
Figure 11 shows the simulated [38] and measured results for
the filter with center frequency control and variable stopband
bandwidth. Table 2 contains a comparison between the simulated
[38] and measured results, where a good agreement between
theory and experiment in terms of filter center frequency and
stopband bandwidth was obtained for both the filter states. A slight
and uniform frequency shift of about 5.5% between simulations
and measurements was obtained, and is believed to be attributed to
small resonator extensions added to the final layout before fabrication, to ease the placement of diodes D1, D2, D3, and D4, this
resulted in an enlargement the overall electrical length of the
resonators when compared with the simulated response.
Figure 12 shows the simulated [38] results considering the
effect of different PIN diode contact resistances r on the switchable bandstop filter with center frequency control and variable
stopband bandwidth. The simulations include the RF choke discussed in Section 2.
For this filter topology, PIN diode contact resistance determines
the degree of stopband rejection, it is apparent from Figure 12 that
a PIN diode with a low contact resistance will produce a deeper
Figure 10 Photograph of the switchable bandstop filter with center
frequency control and variable stopband bandwidth. [Color figure can be
viewed in the online issue, which is available at www.interscience.
wiley.com]
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 11, November 2009
DOI 10.1002/mop
Figure 11 Simulated and measured results for the switchable bandstop
filter with center frequency control and variable stopband bandwidth.
[Color figure can be viewed in the online issue, which is available at
www.interscience.wiley.com]
notch for the filter stopband due to lower resonator loss. The losses
in the filter low frequency state are also higher than the filter high
frequency state due to the use of thinner resonators, combined with
the effect of having two PIN diodes along the resonator for the low
filter center frequency state. The filter discussed in this section is
capable of producing the 7 and 9% fractional stopband bandwidths
for the low and high filter center frequency states with good
accuracy. Loss improvement can be obtained by using PIN diodes
with lower contact resistances, or by having a filter topology where
the diodes are not placed near the maximum current distribution on
the resonators and also by using wider resonators for the filter low
frequency state.
4. CONCLUSION
Two switchable bandstop filters with two center frequency states and
full control of their stopband bandwidth were demonstrated. One filter
topology is adequate to produce the same fractional stopband bandwidth for both filter center frequency states. The second topology
exposed is able to produce two different stopband bandwidths for
each filter center frequency state. The filters discussed in this article
use switchable resonator extensions to fix center frequency and stopband bandwidth parameters. The filter topologies in this paper can be
modified to produce other frequencies and bandwidths which can be
controlled by choosing adequate resonator extensions.
ACKNOWLEDGMENTS
This work has been financed by research project TEC2007– 65705/
TCM from the Spanish Ministry of Education and Culture, and
research project 2006ITT-10005 from AGAUR-Generalitat de
Catalunya.
TABLE 2 Simulated and Measured Results for the
Switchable Bandstop Filter with Center Frequency Control
and Variable Stopband Bandwidth
Simulated
Measured
Stopband Bandwidth
Center Frequency
Tuning Range
(%)
Low Frequency
State (%)
High Frequency
State (%)
32.7
31.2
7.19
6.94
9.14
9.66
DOI 10.1002/mop
Figure 12 Simulated results for different PIN diode contact resistance
for the switchable bandstop filter with center frequency control and variable stopband bandwidth; r1 ⫽ 1 ⍀, r2 ⫽ 2 ⍀, r3 ⫽ 3 ⍀, and r4 ⫽ 4 ⍀.
[Color figure can be viewed in the online issue, which is available at
www.interscience.wiley.com]
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DESIGN OF SQUARE MICROSTRIP
ANTENNA FOR DUAL WIDEBAND
OPERATION
Kishan Singh, R. B. Konda, N. M. Sameena, and S. N. Mulgi
Department of PG Studies and Research in Applied Electronics,
Gulbarga University, Gulbarga-585106, Karnataka, India;
Corresponding author: kishanskrish@gmail.com
Received 15 March 2009
ABSTRACT: A novel design of square microstrip antenna is designed for
dual band operation. By embedding unequal length inverted right angle slot
at optimum place on the square patch, the upper operating band enhanced
from 27.53 to 60.01% with minimum change in the lower operating band
retaining the nature of broad side radiation characteristics. The proposed
antennas may find application in radar communication. © 2009 Wiley Periodicals, Inc. Microwave Opt Technol Lett 51: 2578 –2582, 2009;
Published online in Wiley InterScience (www.interscience.wiley.com).
DOI 10.1002/mop.24678
Figure 1
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 11, November 2009
Geometry of SQMA
DOI 10.1002/mop