RF Transceiver Architectures for W-CDMA
Systems Like UMTS:
State of the Art and Future Trends
R. Weigel1;2 , L. Maurer1 , D. Pimingsdorfer2 , A. Springer1
1
Institute for Communications and Information Engineering, University of Linz, Austria
2
DICE-Danube Integrated Circuit Engineering, Linz, Austria
{weigel, maurer, pimingsdorfer, springer}@mechatronik.uni-linz.ac.at
Invited Paper
Abstract— The standardization phase for wideband
CDMA systems like UMTS is running towards its finalization. As is typical for mobile communication systems
standardizing, sufficient RF performance has been assumed and most efforts have been put to baseband issues.
This is especially true for the pocket phone transceivers
the RF part of which is, although its baseband part is
much more complex in terms of number of devices, still
the bottleneck of the entire system. In the RF concept
engineering of today’s commercial products with their
short time-to-market requirements, a prediction of the
needed RF performance by using RF system simulation
is meanwhile indispensable. This is in particular the case
with third generation (3G) wireless systems which, from
the RF design point of view, are quite different from 2G
TDMA/FDMA systems due to the fact that the users are
now separated in the power domain (using codes) rather
than being separated in the time and/or frequency domain. The present work gives an insight how to derive
the transceiver requirements of 3G mobiles in terms recognizable by RF designers.
Keywords— Transceivers, CDMA, Radio Communication, Spread Sprectrum Communications.
I. I NTRODUCTION
This work gives an introduction to transceiver design for third generation (3G) wireless communications
systems. We start with a review of the transition from
second generation (2G) cellular systems to 3G systems.
The following section introduces basic terms and equations of the spread spectrum technique. The relation
between specific transceiver characteristics like noise
figure or linearity and testcases from the third generation partnership project (3GPP) specifications are covered in the succeeding section. A review of different receiver and transmitter architectures and their suitability for wideband-code division multiple access (W-
CDMA) systems follows. The last section gives a perspective of future trends in transceiver front-end design
for 3G systems.
In 1985 the ITU (International Telecommunications
Union) started work on 3G systems unter the acronym
FPLMTS (Future Public Land Mobile Telephone System) which was later renamed to IMT-2000 (International Mobile Telecommunications) [1].
The key factors and main objectives for 3G systems
include worldwide coverage and roaming incorporating
a satellite component, capacity and capability to serve
more than 50% of the population [2], multimedia service capability, high-speed access, low-cost operation,
and integration of residential, office, and cellular services into a single system based on one piece of user
equipment. Further issues are:
Æ Packet access.
This item is closely linked to the
above. Most of the traffic in 3G networks will originate
from data communications. Therefore, packet switched
communication must be provided in addition to a circuit
switched mode to ensure efficient resource usage. This
feature is or will be introduced already in 2.5G systems
like GPRS [3].
Æ Evolution. The transition from 2G to 3G will be an
evolutionary path. In the beginning 3G systems and
services must coexist with todays 2G and 2.5G systems, since no one (neither users nor network operators) would be able or willing to afford a hard transition
from 2G or 2.5G to 3G.
From the above mentioned items the basic demands
for data throughput over the air interface were identified as 144 kbps (preferably 384 kbps) with full coverage and high mobility of the user equipment (UE) and
up to 2 Mbps for low mobility and coverage limited to
high traffic areas. These bit rates were harmonized to
the ISDN (Integrated Services Digital Network) 2B+D
(144 kbps), H0 (384 kbps), and H12 (1920 kbps) channels [1]. The general vision of 3G systems, is that
they should basically ensure communications from anywhere to anybody at any time.
Upon a request from the ITU for radio transmission
technology (RTT) proposals, different regional standardization bodies submitted their proposals for IMT2000 in 1998 [4]. Details of these proposals are available at [5]. The vast majority of the submitted proposals were based on W-CDMA or at least contained a WCDMA component. During the evaluation of the different proposals by the ITU it turned out that the vision of
a global standard with a single radio interface was not
realizable for 3G systems. This was due to the different 2G technologies used in the different regions in the
world. It would have been impossible to find one technology as evolutionary path for all existing 2G systems.
IMT-2000 Terrestrial
Radio Interfaces
IMT-DS
Direct
Spread
IMT-MC
Multi
Carrier
CDMA
IMT-TC
Time
Code
IMT-SC
Single
Carrier
TDMA
II. E VOLUTION
FROM
2G
TO
3G S YSTEMS
As pointed out before, there is already existing demand for data rates higher than the few kbps nowadays
possible with 2G systems. With the exception of the
PDC system all 2G standards have provided add-on features supporting higher data rates to account for this
traffic demand. A review of these modes, commonly
referred to as 2.5G systems, can be found in, e.g., [10].
If the 2.5G systems are fully deployed they will allow
for data rates up to 384 kbps. With this the mobile
units will evolve from mobile phones to so-called smart
phones including, e.g., PDA’s (Personal Digital Assistants) to better support data applications like email, Internet or location based services. Also mobile Web panels are under development. If the maximum data rate of
1 Mbps in UMTS is available such demanding applications like video conferencing could be supported. This
will introduce again a new generation of mobile terminals capable of dealing with multimedia contents.
III. S PREAD S PECTRUM BASICS
IMT-FT
Frequency
Time
FDMA
Fig. 1. The set of IMT-2000 Terrestrial Radio Interfaces
Therefore, a family concept was adopted and agreed
upon at the end of 1999 [6]. The five standards included
in IMT-2000 are shown in Fig. 1. As IMT-DS (Direct
Spread) the UTRA FDD (UMTS Terrestrial Radio Access Frequency Division Duplex) mode was adopted in
Europe and Japan, IMT-TC (Time-Code) is a combination of the UTRA TDD (Time Division Duplex) (Europe and Japan) and the TD-SCDMA (China) proposals, cdma2000 (USA) is found in IMT-MC (Multi Carrier), IMT-SC (Single Carrier) corresponds to UWC136 (USA), and IMT-FT (Frequency Time) is the European DECT proposal. These five standards are now
further developed in the regional standardization bodies. For the W-CDMA based technologies (IMT-DS
and IMT-TC) the third generation Partnership Project
3GPP was created [7]. A similar group was founded
for the development of the cdma2000 based systems,
termed 3GPP2 [8]. This activity is running in parallel
to 3GPP and is coordinated with 3GPP [9].
The basic of spread spectrum (SS) technology is
given by Claude Shannon’s well known formula for the
channel capacity [11]:
C = W log2
1 + NS :
(1)
C is the channel capacity in bps, W the bandwidth in
Hz, N the noise power and S the signal power.
A spread spectrum system must meet two criteria:
Æ The transmitted bandwidth is much greater than the
bandwidth or rate of the information being sent.
Æ The spreading signal must be independent of information bearing signal (i.e. FM is not SS!).
An important reason for using SS is the linear dependency of the channel capacity C from the bandwidth W
in equation (1), whereas C increases only with the logarithm of the signal to noise ration S/N. Furthermore,
SS techniques have an inherent resistance against interference and jamming. Figure 2 gives an explanation of this ability. Suppose a narrowband interferer is
present in your received signal. Due to the high correlation between the wanted signal (which was spread in
the transmitter by the same sequence) and the locally
generated code for despreading in the receiver, the signal level increases. At the same time the interference
Fig. 2. An interfering signal is spread by a code sequence resulting in a lower power spectral density. The wanted signal level
is increased due to the high correlation of the spreading code and the signal.
signal is spread to a larger bandwidth and therefore the
interference power in the receiver bandwidth decreases.
To gain this increase of the wanted signal level, the locally generated code has to be exactly synchronized to
the incoming wanted signal.
SS-systems can be classified in three main groups:
Æ Direct Sequence SS (DSSS): Spreading is done by a
multiplication of the data carrying signal with a code
sequence of much larger bandwidth.
Æ Frequency Hopping SS (FHSS): Spreading is accomplished by periodicly changing the carrier frequency.
Æ Chirp SS: Carrier frequency varies continuously
(usually linear) during a time interval.
Most SS-systems are based on Direct Sequence-SS. For
that reason we will deal in the following exclusively
with DSSS (also the term SS will refer to DSSS).
The best known advantages of SS systems for cellular system design include the possibility of selective
addressing (Code Division Multiple Access (CDMA))
and the ability to eliminate the effect of multipath propagation by using a RAKE receiver in the mobile station.
Disadvantages incorporate the relatively complex structure of the RAKE-receiver and the need of an accurate
output power control in order to deal with the near-far
problem [12]. The wideband nature of the signal also
leads to the necessity of a wideband modem and wideband baseband amplifier stages. Furthermore, a fast and
accurately working automatic gain control (AGC) circuitry is a prerequisite for an efficient handling of the
multipath signal [13].
SS systems can be described by a few equations
and terms. Important terms when talking about spread
spectrum are the so-called spreading factor SF and the
spreading gain SG. SF describes the ratio of the infor-
mation data rate (represented by the bit duration Tbit )
to the rate of the spreading code (represented by the
chip duration Tchip ). This ratio ranges for, e.g., the
3GPP system from 4 to 512.
SF = TTbit
(2)
SG[dB] = 10 log SF
(3)
chip
Let us denote the signal level before despreading the
chip energy to interference ratio ( EIc /dB) and the signal
level after despreading the bit energy to interference ratio ( EIb /dB). Than EIc /dB, EIb /dB and SG are related by
Eb = Ec + SG + OF:
I
I
(4)
The factor OF/[dB] describes the degree of orthogonality between wanted user signal and interference signal. The orthogonality factor (OF) for e.g. Gaussian
noise equals 0 dB. Therefore, in a Gaussian noise environment the wanted user signal level is increased by an
amount of SG dB. For perfectly orthogonal signals OF
results to 1 dB. Thus the choice of codes employed for
the spreading of the user signals greatly influences the
overall performance of a CDMA system. The orthogonality among the spreading codes should be as large as
possible.
At this point we emphasize once again that these results are only valid for perfect synchronization of the
received signal and the locally generated code for despreading. It can be shown, that a timing error of, e.g.,
one half of the chip time results in an SNR loss of 6 dB.
Therefore obtaining initial synchronization and keeping
the code synchronized by a code tracking loop can be
considered as key problems in SS system design.
IV. T RANSCEIVER R EQUIREMENTS AND
3GPP F RONT-E ND T ESTCASES
The short explanation of some of the 3GPP testcases
below should give the reader an introduction of how RF
key parameters can be derived from specifications given
by 3GPP. The complete set of RF specific testcases for
the 3GPP FDD mode can be found in [14]. Further
comments to these testcases are made in [15].
A. Reference Sensitivity Level Testcase
;
(5)
Inserting values in equation (5) results in:
PN =-117 dBm+21 dB+4 dB-5 dB-2 dB=-99 dBm.
This leaves a margin for the front end noise figure
(NF) of
NF = PN 10 log(kTB=1 mW):
(6)
Inserting values for the Boltzmann constant k, temperature T (300 K) and bandwidth B (3.84 MHz), equation (6) results in a tolerable noise figure NF of
NF = 99 dBm + 108 dBm = 9 dB:
Power Spectral
Density
-52 dBm/3.84 MHz
-92,7 dBm/3,84 MHz
The reference sensitivity is the minimum receiver input power measured at the antenna port at which the bit
error rate (BER) does not exceed a value of 10 3 . This
testcase determines the tolerable noise figure of the receiver front end. The cumulative value of the incoming
signal power is -106,7 dBm. The wanted user signal
level before despreading EIc is -117 dBm. The reference
channel is a 30 ksps channel which yields an SF of 128.
According to equation (3) this result in a value for SG
of approximately 21 dB. Let us assume that the required
E
bit energy to interference ratio bI;req is 5 dB, the insertion loss (IL) for the baseband implementation is 2 dB
and the coding gain (CG) is 4 dB. Then the acceptable
interference signal level after despreading (PN ) result
in:
PN = EIc + SG + CG EbIreq IL:
the center frequency of the assigned channel. Adjacent
channel selectivity is the ratio of the receive filter attenuation on the assigned channel frequency to the receive
filter attenuation on the adjacent channel(s).
(7)
B. Adjacent Channel Selectivity Testcase
Adjacent channel selectivity (ACS) is a measure of
a receiver’s ability to receive a W-CDMA signal at its
assigned channel frequency in the presence of an adjacent channel signal at a given frequency offset from
10,3 dB
DPCH_Ec=-103 dBm
5 MHz
Adjacent
Wanted
Channel
Channel
Fig. 3. Signal levels for the Adjacent Channel Selectivity
testcase.
The ACS shall be better than 33 dB. Simultaneously,
the bit error rate shall not exceed 10 3 for the following test parameters (see Fig. 3). Received power spectral density of the wanted signal at the terminal antenna
connector is -92,7 dBm/3.84 MHz. The wanted user
signal level before despreading EIc is -103 dBm. The
symbol rate of the physical channel is 30 ksps. This results in an SF of 128 which yields a processing gain
of approximately 21 dB. The power spectral density of
the band limited white noise 5 MHz away from the
wanted channel PI is -52 dBm/3.84 MHz.
We assume again that the required bit energy to interE
ference ratio b;Ireq is 5 dB, the insertion loss (IL) for the
baseband implementation is 2 dB and the coding gain
(CG) is 4 dB. This leads us to an acceptable interference level PA of
PA = EIc + SG + CG EbIreq IL;
;
(8)
resulting in a value of
PA = 103 dBm + 25 dB 7 dB = 85 dBm:
(9)
If the adjacent channel interference signal is treated as
Gaussian noise like interference, the required adjacent
channel selectivity (ACS) can be derived:
Image Reject
Filter
ACS = PI PA = 52 dBm + 85 dBm = 33 dB:
(10)
The equations used in this section exemplify of how
the signal levels are influenced by the despreading operation and by interference sources. However, one should
always keep in mind that these results can only serve
as estimates. Further estimations such like the above
mentioned ones can be found in [16].
Wanted Band
11
00
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
Interferer
Image
11
00
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
ωwanted
ωimage
Channel Select
Filter
ω
ωIF
ωIF
ωLO
0
11
00
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
00
11
ωIF
ω
ω
Fig. 5. Image rejection and channel selection for the heterodyne receiver structure.
V. T RANSCEIVER D ESIGN
A. General Considerations
Complexity, cost, power dissipation, and the number
of external components have been the primary criteria
in selecting transceiver architectures. As IC technologies evolve, however, the relative importance of each
of these criteria changes, allowing approaches that once
seemed impractical to return as plausible solutions [17].
B. Receiver Architectures
B.1 Heterodyne Receiver
Figure 4 shows the heterodyne receiver structure.
This architecture first translates the signal band down
to some intermediate frequency (IF), which is usually
much lower than the initially received frequency band.
Channel select filtering is usually done at this IF, which
relaxes the requirements of the channel select filter. The
choice of the IF is a principal consideration in heterodyne receiver design (see Fig. 5).
As the first mixer downconverts frequency bands
symmetrically located above and below the local oscillator (LO) to the same center frequency, an image reject
filter in front of the mixer is needed. As depicted in the
left part of Fig. 5, the filter is designed to have a relatively small loss in the desired band and a large attenuation in the image band, two requirements that can be
simultaneously met if 2!IF is sufficiently large. Thus,
a large IF relaxes the requirements for the image rejection filter, which is placed in front of the mixer (see Fig.
4). On the other hand it complicates the design of the
channel selection filter (right part of Fig. 5), because of
the higher IF. In today’s cellular systems the channel selection filtering is normally done with surface acoustic
wave (SAW) filters [18].
Another interesting situation arises with an interferer at (!wanted + !LO ) =2. If this interferer experiences second-order distortion and the LO contains
a significant second harmonic, then a component at
j(!wanted + !LO) 2!LOj = !IF arises. This phenomenon is called half-IF problem [19].
A major advantage of the heterodyne receiver structure is its adaptability to many different receiver requirements. That is why it has been the dominant
choice in RF systems for many decades. However, the
complexity of the structure and the need for a large
number of external components (e.g., the IF filter) make
problems if a high level of integration is necessary. This
is also the major drawback if costs are concerned. Furthermore, amplification at some high IF can cause high
power consumption.
The IMT-2000 proposal specifies an operation mode
using two times or four times the base chiprate resulting
in a bandwidth of 7,68 MHz or 15,36 MHz, respectively
(multiband operation). On the other hand the handsets should also be able to receive GSM signals with a
bandwidth of approximately 200 kHz (multimode operation). Due to the fixed receive bandwidth of the heterodyne receiver structure caused by the external IF-filter,
these multimode and multiband capability can only be
implemented by using a separate IF section for each
mode. This would result in high costs and a complex
receiver structure.
B.2 Homodyne Receiver
The homodyne receiver structure (also called zeroIF or direct-conversion architecture) entails vastly different issues from the heterodyne topology. Suppose
that the IF in a heterodyne receiver is reduced to
zero. The LO will then translate the center of the desired channel to 0 Hz, and the channel translated to
VGA
Band
Select
Filter
LNA
Image
Reject
Filter
Channel
Select
Filter
0
90
I
A
Q
A
D
VCO
VCO
D
Fig. 4. Heterodyne receiver structure.
Channel
Select
Filter
Preselect
filter
VGA
I
A
Q
A
D
LNA
0
90
VCO
D
Fig. 6. Homodyne receiver structure.
the negative frequency half-axis becomes the image to
the other half of the same channel translated to the
positive frequency half-axis. The downconverted signal must be reconstituted by quadrature downconversion (or some other phasing method), otherwise the
negative-frequency half-channel will fold over and superpose on to the positive-frequency half-channel [20].
The simplicity of this structure offers two important
advantages over a heterodyne counterpart. First the
problem of image is circumvented because !IF = 0. As
a result, no image filter is required. This may also simplify the LNA design, because there is no need for the
LNA to drive a 50 load, which is normally necessary
when dealing with image rejection filters. Second, the
IF SAW filter and subsequent downconversion stages
are replaced with low-pass filters and baseband amplifiers that are amenable to monolithic integration. The
possibility of changing the bandwidth of the integrated
low-pass filters (and thus changing the receiver bandwidth) is a major advantage if multimode and multiband applications are concerned.
On the other hand the zero-IF receiver topology entails a number of issues that do not exist or are not
as serious in a heterodyne receiver. Since in a homodyne topology the downconverted band extends to zero
frequency, offset voltages can corrupt the signal and,
more importantly, saturate the following stages. There
are three main possibilities how DC-offsets are generated. First, the isolation between the LO port and the
inputs of the mixer and the LNA is not infinite. Therefore, a finite amount of feedthrough exists from the LO
port to the mixer or the LNA input. This “LO leakage”
arises from capacitive and substrate coupling and, if the
LO signal is provided externally, bond wire couplings.
This leakage signal is now mixed with the LO signal,
thus producing a DC component at the mixer output.
This phenomenon is called “self-mixing”. A similar effect occurs if a large interferer leaks from the LNA or
mixer input to the LO port and is multiplied by itself. A
time varying DC offset is generated if the LO leaks to
the antenna and is radiated and subsequently reflected
from moving objects back to the receiver.
Large amplitude modulated signals that are converted to the baseband section via second order distortion of the IQ-mixers also lead to time varying DC offset. The spectral shape of this signal contains a signif-
icant component at DC accounting for approximately
50% of the energy. The rest of the spurious signal extends to two times of the signal bandwidth before downconverted by the second order nonlinearity of the mixers. The cause for the large signal content at DC is that
every spectral component of the incident interferer is
coherently downconverted with itself to DC. In order
to prevent this kind of DC offset, a large second order
intercept point (IP2) of the IQ-mixer is necessary.
3GPP compliant receivers need approximately 80 dB
gain. Most of this gain is contributed by the baseband
amplifiers. That means that even small DC offsets (in
the range of several mV) at the mixer outputs may lead
to DC levels sufficient to saturate the analog to digital
converters (ADC).
In time-division multiple access (TDMA) systems
idle time intervals can be used to carry out offset cancellation. This would be a practical solution for the
3GPP-TDD mode. It can not be used for offset cancellation in the FDD mode, because of the continuous
signal reception. Here, the natural solution for DC offset cancellation is high-pass filtering. Since the signal
band extends from DC to approximately 2 MHz, a highpass filter with a cut-off frequency of several kHz results in an acceptable degradation of the system performance [21]. This approach is only possible because of
the wideband nature of the signal. A system level DC
offset compensation approach is described in [22].
I/Q mismatches are another critical issue for the zeroIF receiver topology. Fortunately, pilot symbol assisted
channel estimation is done in W-CDMA systems. Irrespective of the pilot symbols used (either the time multiplexed pilot symbols or the common pilot signal), this
estimation leads also to a correction of the I/Q phase
and amplitude mismatch.
B.3 Digital-IF Receivers
In the heterodyne receiver architecture of Fig. 4 the
second downconversion and subsequent filtering can be
done digitally. The principal issue in this approach is
the performance required from the ADC. To limit the
requirement on the ADC, a sufficiently low IF has to
be chosen, which makes it impossible to employ bandpass filtering to suppress the image frequency. Thus,
an image suppression mixer has to be used. The image suppression feasible in today’s systems is limited
to a range of 30-55 dB. Due to the high demands on
the ADC and the image suppression mixer performance
this architecture has not been used for terminal applications. Nevertheless, it is utilized in base stations where
man channels must be received and processed simultaneously.
C. Transmitter Architectures
C.1 Direct Conversion Transmitter
It the transmitted carrier frequency is equal to the local oscillator frequency, the architecture is called “direct conversion”. In this case, modulation and upconversion occur in the same circuit. The architecture in
Fig. 7 suffers from an important drawback. Through
a mechanism called “injection pulling” or “injection
locking” the transmit LO spectrum is corrupted by the
power amplifier (PA). The problem worsens if the PA is
turned on and off periodically, as it is the case for the
3GPP-TDD mode.
Problems also arise if the system has to fulfil tight
requirements on output power range, which is usually
necessary in W-CDMA systems. Most of the gain has
to be done in the baseband section, leading to high
linearity requirements for the baseband filters and the
modulator. Furthermore, the LO lies always in the
transmit band, which causes high requirements on the
LO-RF isolation. I/Q phase mismatches are also an issue when using direct upconversion. Even a low error in
the phase shifting network may lead to a severe degradation of the error vector magnitude (EVM).
C.2 Heterodyne Transmitter
The second possibility of signal upconversion, which
circumvents the problem of LO pulling in transmitters,
is to upconvert the baseband signal in two steps so that
the PA output spectrum is far from the frequency of the
VCO’s. An advantage of two-step upconversion over
the direct conversion approach is that since quadrature
modulation is performed at lower frequencies, I and Q
matching is superior. On the other hand, an IF filter (in
most cellular applications again a SAW filter) is needed
which can rise costs considerably.
If high integration is an important feature, then both
heterodyne transmitters and receivers can cause problems. Trying to find intermediate frequencies for the
transmit and receive section, that do not lead to spurious frequencies falling, e.g., in the receive band, may
prove to be impossible. This is especially true, if single
chip transceivers are concerned.
I
D/A
AGC
RF SAW
LO
90°
Q
D/A
+
PA
AGC
Fig. 7. Direct upconversion transmitter structure.
I
D/A
IF SAW
LO IF
90°
Q
+
RF SAW
AGC
D/A
AGC
PA
LO RF
Fig. 8. Heterodyne transmitter.
VI. C URRENT D EVELOPMENTS
T RENDS
AND
F UTURE
The first operable UMTS IF transceiver front end was
published in Reference [23]. This paper describes a
fully integrated Si-bipolar IF receiver and transmitter
with on-chip synthesizer for use in third-generation WCDMA mobiles. Both devices in a small outline leadless package incorporate an on-chip IF synthesizer with
on-chip VCO tuning and tank as well as 6th /5th order
baseband filters and comply with ARIB W-CDMA and
UMTS standards. The IF-chips are fabricated with Infineon’s high frequency 0.4 m/25 GHz silicon bipolar process. IF receiver and IF transmitter die size is
2.33x2.9 mm2 . The devices are designed for low external component count and, together with the small package size, minimize the required board area of a complete IF transceiver. The chips operate at 2.7-3.3 V supply, an ambient temperature range of -30 to +85, and
incorporate several power-down modes for efficient use
in W-CDMA mobile stations. The W-CDMA IF receiver includes two complete IF paths for antenna diversity/service channel monitoring and a common LO
generation and distribution. Each path features a variable gain amplifier with >95 dB gain range at an IF
frequency of 318 MHz, a quadrature demodulator and
a 5th -order Chebyshev filter and 1st -order all-pass for
the differential I/Q outputs. The IF synthesizer includes
a completely integrated on-chip VCO with integrated
transformer and varactor diodes, tuning circuitry and
on-chip voltage regulator for the VCO/buffer. A fixed
PLL with reference divider, RF prescaler, lock detect
circuitry and three external elements for 3rd -order loop
filter complete the on-chip synthesizer. The W-CDMA
IF transmitter includes a 5th order active Butterworth
baseband pre-filter, a quadrature modulator, a variable
gain amplifier with >60 dB gain range at a fixed IF frequency of 285 MHz. The fully integrated VCO operates
at a frequency of 1520 MHz.
Most of the published work on receiver design is
based on the direct conversion topology. It seems that
especially the need for high integration restricts the
receiver architecture to the zero-IF structure. Examples can be found in [24][25][26][27][28]. All these
receivers are designed using standard BiCMOS processes. An interesting option is the use of Si/Ge bipolar technology for the receiver front end [29][30]. This
receiver incorporates an I/Q down conversion mixer,
broad-band I/Q-generation, fully- integrated VCO,
dual-modulus prescaler, low-noise baseband buffer and
a blocking filter. Integrated in a 75 GHz ft BiCMOS
technology with 35 m CMOS it draws 33 mA from
a 2.7 V supply. Remarkable are the extremely low
LO leakage of -95 dBm together with a high IIP2 of
55 dBm. This results in very low DC offset values of
less than 20 mV at the baseband output of the IC.
With the continuing development of complementarymetal-oxide-semiconductor (CMOS) components and
processing techniques, CMOS technology is also ex-
pected to become suitable for RF applications within
a few years. Advances with respect to improved devices, circuit topologies, and system-level architecture make CMOS a strong contender for implementing
transceivers for the IMT-2000 global wireless system
[31]. RF CMOS seems to be a promising approach,
even though the performance of some key components
like PAs, VCOs and filters is not sufficient yet. Considering the rapid development of RF CMOS technology,
however, some of these building blocks may become
available within a few years. An interesting project
aimed at RF-CMOS design for W-CDMA transceivers
is described in Reference [32].
VII. C ONCLUSION
A review of cellular phone transceiver concepts for
IMT 2000 system use was given. It was shown how
basic spread spectrum equations lead to an estimation
of the transceiver system requirements. Examples exhibit the influence of 3GPP testcases on key parameters
for the transceiver design. Furthermore, a review of the
most popular receiver and transmitter architectures was
given. Their suitability for the W-CDMA system was
evaluated and possible problems were addressed. The
last section reviews the state-of-the art and discusses
future trends of W-CDMA transceiver design. Possible advances with respect to improved devices, circuit
topologies, and system-level architecture can make RF
CMOS based transceivers a promising possibility.
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