EXTENDED ABSTRACT FOR OBTAINING THE MASTER’S DEGREE IN ELECTRICAL AND COMPUTER ENGINEERING, OCTOBER 2014
1
Microstrip antennas with cylindrical geometry
Pedro José da Silva Paraı́so (77107)
University of Lisbon (UL)
Instituto Superior Técnico (IST)
Lisbon, Portugal
pedro.paraiso@ist.utl.pt
Abstract—Microstrip patch antennas are probably the most
used antenna type of the present. Such success comes from
well known advantageous mechanical characteristics (low profile,
light weight, planar but conformal to non-planar structures, easy
to fabricate), flexibility in terms of electromagnetic parameters
(radiation pattern, gain, impedance, polarization) and low cost.
The objective of this thesis is to study microstrip patch
antennas with cylindrical geometry. Application in mobile communication systems’ base stations with one, two and three sectors
is envisaged.
For the one sector and three sectors applications only proof of
concept is addressed. For the two sectors antenna configuration a
complete approach is followed. Design, optimization, fabrication
and test of arrays of four microstrip patches for the LTE1 band
is carried out.
The experimental results obtained show a good agreement with
numerical simulation and therefore revalidate the approach used.
This concept can be extended to base stations with more sectors
with the necessary adjustment of the radius of the supporting
cylinder.
Index Terms—Microstrip antennas, conformal antennas, cylindrical antennas, base station antennas, sectorial antennas.
I. I NTRODUCTION
ICROSTRIP antennas became very popular in the
1970s primarily for space borne applications. Today
they are used for military and commercial applications. These
antennas consist of a metallic patch on a grounded substrate.
The metallic patch can take many different configurations, the
most common being rectangular and circular. The rectangular
and circular patches are the most popular because of easy of
analysis and fabrication, and their attractive radiation characteristics. The microstrip antennas are low profile, conformable
to planar and nonplanar surfaces, simple and cheap to fabricate using modern printed-circuit technology. They are also
mechanically robust when mounted on rigid surfaces and very
versatile in terms of resonant frequency, polarization, radiation
pattern and impedance. These antennas can be mounted on
the surface of high-performance aircraft, spacecraft, satellites,
missiles, cars, and even in cellphones [1].
Since many surfaces are not exactly planar, conformal
microstrip structures have received increasing attention in the
last decades, due to the need of antennas and devices that can
be mounted on curved surfaces.
With the continuous improvement in computing power and
numerical techniques, the analysis of electromagnetic phenomena in these curved surfaces is now easier, cheaper and faster,
since they allow reducing time and costs in comparison to a
purely experimental optimization.
M
A. Motivation
Conformal microstrip structures have received increasing
attention in the last decades but their applications are essentially on vehicles: aerial, nautical and terrestrial. However
other novel applications can be envisaged. This thesis focus
its application in a base station, to be used in cells with
different traffic requirements. Their robustness and flexibility
make conformal microstrip antennas a good approach for these
scenarios. This base station is adaptable to different types
of structures which permit more flexibility in covering areas
which would not be possible to cover with another antennas.
This kind of antennas are also a good low cost approach and
that is why they are so appealing.
There are many ways of building these antennas and its
implementation depends always on the type of connection that
wants to be established, namely, the traffic profile specifications. ”Wraparound” antenna is a type of cylindrical microstrip
antenna. It can replace a typical dipole since its characteristics
are really similar in terms of radiation pattern. As dipoles,
this type of antenna can be used in a rural area as base
station. If interference is a problem in the system, it may
be installed a cylindrical sectorial antenna, which consists on
printing two or three rectangular arrays in a substrate, the first
one, diametrically oppose or the second one, separated 120◦ ,
respectively. Once again, it depends on the traffic requested
per users.
B. Objectives
The main scope of this thesis is to study cylindrical microstrip antennas to be used in a mobile communication system
as base station. Three types of cylindrical antennas will be
studied: ”wraparound” antenna, 180◦ sectorial antenna and
120◦ sectorial antenna. The ”wraparound” antenna is based on
an omnidirectional printed ring. The 180◦ sectorial antenna
is based on two microstrip arrays whereas 120◦ sectors are
obtained with three microstrip arrays. The prototype application is a base station operating in LTE1 downlink band
[2.11, 2.17] GHz. First approach is to design and optimize
an element, an array and then the sectorial antenna in CST
Microwave Studio. After obtaining good simulation results,
a prototype which will be subject to measurements in the
anechoic chamber. This thesis provides a comparison between
simulation and measurement results in order to validate the
antenna’s behaviour.
EXTENDED ABSTRACT FOR OBTAINING THE MASTER’S DEGREE IN ELECTRICAL AND COMPUTER ENGINEERING, OCTOBER 2014
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II. F UNDAMENTAL C ONCEPTS
A. Antenna Modelling
There are two big classes of antennas analysis: analytical or
almost analytical (based on empirical methods) and numerical
methods. The main analytical methods are transmission line
and cavity models.
1) Transmission Line Model: The Transmission Line
Model is the easiest of all but the results are less accurate
and it lacks versatility. A rectangular microstrip antenna can
be represented as an array of two radiating narrow apertures
(slots), each of width W and height h, separated by a distance L, as shown in figure 1. Basically the transmission
line model represents the microstrip antenna separated by a
low-impedance Zc transmission line of length L. Thus the
microstrip radiator can be characterized by two slots separated
by a transmission line, where each slot is represented by a
parallel circuit of conductance (G) and susceptance (B) [2].
Fig. 1: Microstrip line.
The characteristic impedance (Zc ) to match the antenna
using a microstrip line with W
h > 1 is given by,
120π
Zc = √
ref f
W
+ 1.393 + 0.667ln
h
W
+ 1.444
h
(1)
Note that this W is the transmission line width and not the
patch antenna width.
Because of fringing effects, it must be considered a distance
∆L, which is the extension of the length, L. It can be seen
on figure 2. This ∆L is a function of the effective dielectric
constant and the width-to-height ratio (W/h). A popular and
empirical approximate relation for the normalized extension
of the length is
(ref f + 0.3) W
∆L
h + 0.264
= 0.412
(2)
h
(ref f − 0.258) W
h + 0.8
Fig. 2: Physical and effective length of a rectangular microstrip
patch.
The normalized fields, with respect to their maximum value,
within the dielectric substrate (between the patch and the
ground plane) can be found more accurately by treating that
region as a cavity bounded by electric conductors (above and
below it) and by magnetic walls (to simulate an open circuit)
along the perimeter of the patch. This model is limited to
electrically thin substrates, since it is based on the assumption
that the only electric field component existing inside the cavity
is the one perpendicular to the ground and to the patch.
To analyse the cavity lets understand the formation of the
fields within the cavity and radiation through its side walls.
When the microstrip patch is energized, a charge distribution
is established on the upper and lower surfaces of the patch, as
well as on the surface of the ground plane according to figure
3. The charge distribution is controlled by two mechanisms:
an attractive and a repulsive mechanism [3]. The attractive
mechanism is between the corresponding opposite charges on
the bottom side of the patch and the ground plane, which
tends to maintain the charge concentration on the bottom of
the patch. The repulsive mechanism is between like charges
on the bottom surface of the patch which tends to push
some charges from the bottom of the patch, around its edges,
to its top surface. The movement of theses charges creates
corresponding current densities Jb and Jt, at the bottom and
top surfaces of the patch, respectively, as shown in figure
3. These current flow decreases as the height-to-width ratio
decreases but it is never exactly zero creating some tangential
magnetic field components to the edges of the patch. However,
since they will be small, a good approximation to the cavity
model is to treat the side walls as a perfectly magnetic
conducting. This model produces good normalized electric and
magnetic field distributions (modes) beneath the patch.
For the dominant T M010 mode, the resonance frequency of
the microstrip antenna is a function of its length. Usually it is
given by
(fr )010 =
1
√ √
2L r µ0 0
(3)
2) Cavity Model: Any microstrip radiator can be thought
of as an open cavity bounded by the patch and its ground
plane. The open edges can also be represented by radiating
magnetic walls. Such a cavity will support multiple discrete
modes similar to that of a completely enclosed metallic cavity
[2].
Fig. 3: Charge distribution and current density creation on
microstrip patch.
For example for T M100 mode, field is radiated from the
EXTENDED ABSTRACT FOR OBTAINING THE MASTER’S DEGREE IN ELECTRICAL AND COMPUTER ENGINEERING, OCTOBER 2014
1) Cavity model: The assumption that the conducting patch
and the conducting cylinder (ground surface) act as electric
walls, and that the open cavity ends act as magnetic walls
is applied to the analysis for obtaining the fields and associated modal resonant frequencies. These assumption should be
particularly valid when using these fields for determining the
radiation pattern for the limiting case of thin cavities (h << a)
which are utilized for most microstrip antenna applications [4].
Note that h is the thickness of the substrate and a the surface
curvature. All the analysis for simplicity also assumes that the
permittivity and permeability µ are constant (homogeneous
medium filling cavity) and real (no dielectric losses).
The geometry of the cavity is shown in figure 5 where
figure (a) is a perspective drawing of a conducting patch on
a cylindrical surface, figure (b) is a cross section through
the patch and normal to the z-axis, and figure (c) shows the
cavity isolated by itself in cross section. The conducting patch
and cylindrical surface are treated as electric walls and the
magnetic walls of the cavity are defined by dropping perpendiculars from the patch edges to the cylindrical conducting
surface. The electric walls are located at ρ = a and ρ = a + h.
The magnetic walls are located at z = 0, −2b and φ = 0, 2θ.
The field solution is found through a lot of mathematical
steps described in detail in the dissertation, [5]:
(a)
(b)
Fig. 4: Resonant cavity model.
four side slots. The slots radiation (not radiating) y = ± W
2
cancels in main planes E(φ = 0, π) and H φ = π2 , 3π
2 , as
shown in figure 4 (b).
•
lπ
2b
mlπ 2
4θb
Hρ =
j
Amli kmi
ωµ
Hφ =
j
Amli
ωµρ
H-Plane (φ= π2 , 3π
2 )
Eθ = 0
lπ
z
2b
(9)
mπ lπ
φ cos
z ; Ez = 0
2θ
2b
(10)
Rυ (kmi ρ)sin
(5)
2W e−jkr
h
Le
Eθ = −jV0
cos k0 cosθ cos k0 sinθ
λ0 r
2
2
(6)
•
(4)
E-Plane (φ=0,π)
Eφ = 0
mπ mπ 1
φ cos
Rυ (kmi ρ)cos
Eρ = − Amli
ρ
2θ
2θ
Eφ = Amli kmi Rυ0 (kmi ρ)sin
According to figure 4 (a), the far-zone fields are
→
−
E = Eθ eˆθ + Eφ eˆφ
3
(7)
sin k0 ω2 sinθ
2W e−jkr
h
Eφ = −jV0
cos k0 cosθ cosθ
λ0 r
2
k0 ω2 sinθ
(8)
B. Cylindrical-rectangular microstrip antenna
In this section are presented two types of initial approaches:
cavity model and electric surface current model. These models
are a good initial approach, in fact, they can give a good insight
of the problem. According to [4], cavity model can be used
to find the radiation field solutions (curvature is taken into
account).
Hz =
mπ lπ
φ sin
z
2θ
2b
(11)
Rυ (kmi ρ) cos
mπ lπ
φ sin
z
2θ
2b
(12)
mπ −j
Amli (kmi )2 Rυ (kmi ρ)sin
φ cos
ωµ
2θ
lπ
z
2b
(13)
There is an alternative model which calculates fields through
the electric surface current but it is not explored in this
extended abstract.
III. D ESIGN OF A CYLINDRICAL MICROSTRIP ANTENNA
OPERATING AT LTE1 BAND (2.045 GH Z )
A. Introduction
This section discusses all steps adopted during the procedure
in order to obtain the prototype operating in LTE1 band and
respective simulations. Table I shows all the LTE bands and
bandwidths.
EXTENDED ABSTRACT FOR OBTAINING THE MASTER’S DEGREE IN ELECTRICAL AND COMPUTER ENGINEERING, OCTOBER 2014
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B. Design and Development
In this section it will be presented three different antennas
are presented: a ”wraparound antenna”, a two sectors antenna
(180◦ sectorial antenna) and a three sectors antenna (120◦
sectorial antenna). The first attempt is to design a cylindrical microstrip antenna that behaves like an omnidirectional
antenna. Wraparound antenna reveals to be a good approach
to obtain an omnidirectional antenna and at the same time
a cylindrical microstrip antenna. Then a two sectors antenna
simulations is presented. In a scenario with more users (urban
area) may be a problem, this is a good solution comparing
to the first one. Last, but not least, an antenna solution for
a 3 sectors base station is proposed. After the simulation
processes, one of the antennas will be chosen to evaluate in a
prototype fabrication and measurement.
1) ”Wraparound” Antenna: ”Wraparound” antenna is an
example of cylindrical microstrip antennas that can replace a
typical dipole because its radiating characteristics are almost
the same. As a base station, it can replace a common dipole
to be installed in a scenario with small traffic requirements,
such in a rural area.
The first goal is to find the best surface curvature radius of
the cylinder in order to obtain an omnidirectional antenna.
Since surface curvature radius is varying, the length of
the patch and feed line penetration are optimized in each
simulation in order to obtain the best matching possible. The
geometry of the structure in study is presented in figure 6. The
radiating element involves the cylindrical surface and it is fed
by a transmission line (or fed probe).
(a)
(b)
(c)
Fig. 5: Perspective drawing of cylindrical-rectangular cavity
and different cross sections.
Band
UL
(MHz)
DL
(MHz)
1
19201980
18501910
17101785
17101755
824-849
..
.
21102170
19301990
18051880
21102155
869-894
..
.
2
3
4
5
..
.
Simp.
BW
(MHz)
60
Total
BW
(MHz)
120
Mode
60
120
FDD
75
150
FDD
45
90
FDD
25
..
.
50
..
.
FDD
..
.
FDD
TABLE I: Global frequency band for LTE deployment.
Fig. 6: Patch antenna printed on a cylindrical surface.
Taking into account the CST simulation results presented
below, the best choice is clearly 10 mm. Note that are considered the following surface curvature radius: 5, 10, 15, 20, 25
and 30 mm. Bigger than this does not have any meaning to this
study because the antenna will not present an omnidirectional
radiation pattern. This surface curvature allows the antenna to
radiate with an almost omnidirectional radiation pattern, which
is clearly what is intended. Note that, (a) represents the farfield
gain for φ = π/2 and θ is varying between 0 and 2π in every
EXTENDED ABSTRACT FOR OBTAINING THE MASTER’S DEGREE IN ELECTRICAL AND COMPUTER ENGINEERING, OCTOBER 2014
(a)
5
H plane once again looses its omnidirectional property and it
is turning into a cardioid curve and E plane is not radiating
its maximum value of gain at 270◦ . H plane and E plane
maximum gain are 3.7 dBi. In the vertical plane, half-power
beamwidth is 55.2◦ and front-to-back ratio is 1.8 dB.
Hereupon, a surface curvature radius of 10 mm is considered. This case is studied in detail in this section.
The frequency range is reduced to [1.9, 2.2] GHz. Thus,
it is possible to study the antenna characteristics in detail.
Figure 8 shows a -23 dB of return loss for the average
frequency of the LTE1 band (note that this average results
from the minimum value of uplink and the maximum value of
downlink). On horizontal plane, figure 7, antenna is practically
omnidirectional. The maximum value is 4.4 dBi of farfield gain
absolute at 0◦ and minimum value is approximately 3.3 dBi
at 180◦ .
(b)
Fig. 7: Surface curvature radius: 10 mm.
Fig. 8: |S11 | with surface curvature radius of 10 mm.
figures (H plane). (b) represents the farfield gain for φ varying
between 0 and 2π and θ = π/2 (E plane).
For a surface curvature radius of 5 mm, the antenna behaves
as expected. The H plane is pretty good with the main lobe
reaching 3.9 dBi of gain. In the vertical plane, half-power
beamwidth is 45.1◦ and front-to-back ratio is 1.2 dB. For
a surface curvature radius of 10 mm, the antenna behaves
as expected. The H plane is even better than for a surface
curvature radius of 5 mm with the main lobe reaching 4.4 dBi
of gain. In the vertical plane, half-power beamwidth is 44.2◦
and front-to-back ratio is 1 dB.
For a surface curvature radius of 15 mm, the antenna
behaves in a strange way. The H plane looses quality in its
omnidirectional property and in E plane the main lobe is
radiating to 279◦ instead of being radiating on 270◦ . H plane
maximum gain is 3.7 dBi whereas E plane maximum gain is 4
dBi. In the vertical plane, half-power beamwidth is 45.8◦ and
front-to-back ratio is 1.3 dB. For a surface curvature radius
of 20 mm, the antenna behaves as expected. The H plane
presents a good behavior but not so good as for a surface
curvature radius of 10 mm. The same happens to E plane. H
plane maximum gain is 4.1 dBi whereas E plane maximum
gain is 3.9 dBi. In the vertical plane, half-power beamwidth is
54.1◦ and front-to-back ratio is 1 dB. For a surface curvature
radius of 25 mm, the antenna behaves in a strange way. The
H plane once again looses its omnidirectional property and
E plane is not radiating its maximum value of gain at 270◦ .
H plane maximum gain is 4 dBi whereas E plane maximum
gain is 3.9 dBi. In the vertical plane, half-power beamwidth is
53.4◦ and front-to-back ratio is 1.6 dB. For a surface curvature
radius of 30 mm, the antenna behaves in a strange way. The
It is seen that the wraparound antenna behaves like a dipole
so it is a good solution for an omnidirectional antenna. It is
crucial to verify if the wraparound antenna covers the LTE1
band. In order to occupy all the band it is necessary,
2.170 − 1.920
× 100 = 12.22%
(14)
2.045
The wraparound antenna does not cover all the LTE1 band,
through |S11 | parameter. If it is considered a threshold of -6 dB
for good matching, the bandwidth covered by this antenna is
close to 40 MHz which is not enough. One of the many ways
to fix it, is to use stacked microstrip antennas to enhance the
bandwidth.
2) 180◦ Sectorial Antenna: The first approach is to simulate
a single element printed in the cylindrical surface in order to
adjust its dimensions so that it resonates in the required central
frequency (2.14 GHz). The second approach is to simulate an
array of four elements printed in the cylindrical substrate, once
again to adjust its dimensions so that the array resonates in
the central frequency. Finally, it is necessary to replicate the
array and put it diametrically oppose to the first one to build a
sectorial antenna (180◦ sectorial antenna). Note that, only the
downlink band is taking into account during simulations.
Since it is an extended abstract only sectorial simulations
will be presented.
After adjusting and analyzing, the array is replicated and
placed diametrically oppose to the first one in order to obtain
the 180◦ sectorial antenna. The surface curvature of 55 mm
may be more than enough to avoid coupling between the two
arrays (perimeter is greater than 2λ). To prove it, the following
B% =
EXTENDED ABSTRACT FOR OBTAINING THE MASTER’S DEGREE IN ELECTRICAL AND COMPUTER ENGINEERING, OCTOBER 2014
6
figures shows three different simulation results: |S11 |, |S22 |
and |S12 |.
Fig. 11: |S11 | of array.
Fig. 9: |S11 | and |S22 | of sectorial antenna.
The black curve represents |S11| parameter and the red
curve represents |S22|. Ideally, curves on figure 9 should
overlap. Every bandwidth needed is covered if it is considered
a threshold of adaptation below -6 dB. Figure 10 shows the
coupling between the two arrays. In practise, -30 dB would
be a reasonable value. According to figure 10, in LTE1 band
the values of |S12 | are below -50 dB.
Fig. 10: |S12 | of sectorial antenna.
3) 120◦ Sectorial Antenna: As for the 180◦ sectorial antenna, first approach is to simulate a single element printed in
the cylindrical surface in order to adjust its dimensions so that
it resonates in the central frequency (2.14 GHz). The second
approach is to simulate an array of four elements printed in
the cylindrical substrate, once again to adjust its dimensions
so that the array resonates in the central frequency. Finally,
it is necessary to replicate the array twice and place them
separated by 120◦ in the substrate to build a three sectorial
antenna (120◦ sectorial antenna).
Since it is an extended abstract only array simulations will
be presented.
Figure 11 shows the array |S11 |. It is seen that the bandwidth
covered is higher but it did not increase as much as it was
necessary. It is covering 30 MHz of the 60 MHz needed.
Unfortunately, the array is too narrow and it will not cover
all the band needed.
Since the array is not covering all the bandwidth needed,
it is not possible to move on for a 120◦ sectorial antenna. It
would be necessary a thicker substrate or stacked patches in
order to obtain the bandwidth necessary.
The prototype chosen for fabrication and test is the 180◦
sectorial antenna because it has better results in all simulations
and needs less resources to be built. The 120◦ sectorial would
need three substrate ”sheets” (each array) of RT Duroid 5880
and that would increase the cost.
IV. FABRICATION AND M EASUREMENTS
A. Fabrication
This section describes the procedure to assemble the sectorial antenna in the cylindrical surface. It is a complex process
because of RT Duroid 5880 plates rigidity. Bending these
plates requires some strength and it must be done carefully
because the copper from the patches and the splitter can break
during this procedure.
The first step is to turn planar the layout extracted from
CST. In order to do that, the antenna is exported to AutoCAD.
This software tool will allow to draw the antenna in a planar
surface. Since the arrays are similar, it is only needed to draw
half of the side area of the cylinder. In order to simulate
the antenna in CST, it was considered a line with 50 Ω
characteristic impedance during simulations because CST does
not allow the user to build a port in an oblique plane. This
line of 50 Ω is only an extension to the border of the cylinder
in order to obtain an horizontal or vertical port, as shown in
figure 12. This line is erased in AutoCAD and replaced by
a hole in the middle of the line with 100 Ω characteristic
impedance which will be the input for the SMA connector.
The red circle is the place where this hole is placed to allow
using the SMA connector.
Another important thing is that during the process of turning
the structure planar so it can be printed in the plates, every
dimensions must be the same. For example, a piece of paper
with four squares drawn in form of cylinder and then it is
cut in the middle which results in two planar piece of papers
(square dimensions maintain the same dimensions).
The next step is to print the antennas in the substrate through
chemical procedures. The result is two plates, each one with
an array in the top and ground plane in the bottom. Each plate
has the same dimensions and is half of the cylinder’s side area.
Before move on to bending, it is necessary to find a way
of bend the plates and stuck them to the pipe. For that, it was
EXTENDED ABSTRACT FOR OBTAINING THE MASTER’S DEGREE IN ELECTRICAL AND COMPUTER ENGINEERING, OCTOBER 2014
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1) S Parameters: In this section, the S parameters of the
antenna are presented. To perform this measurement, a VNA
was used. Usually, these measurements can be done outside
the anechoic chamber but since the antenna has two arrays, it
is better to do it inside of the chamber to avoid any undesired
reflection. Figure 14 shows the setup used to perform this
measurement.
Fig. 12: Line extension to the border and vertical port.
made a test with Araldite glue which turns out to be important
because the tension caused by the plate bending made the plate
unstick from the pipe.
In order to gain more friction the pipe is sanded and it
is drilled in all side area so that the glue can create some
kind of spikes that support the collage. This sanding procedure
is not wanted because it was not taking into account during
simulations. This sanding procedure reduces the curvature
radius and consequently, it may change some characteristics
on the antenna. During the collage procedure it is necessary
to guarantee that the arrays are diametrically oppose.
Figure 13 shows the final result of the prototype. Some
pipe is needed to stuck in the structure that is supporting the
antenna. The next section explains all the measurements done
and it is made a comparison between the results achieved in
the simulations and in the anechoic chamber.
Fig. 13: Prototype in anechoic chamber.
B. Test
This section consists on explaining every procedure and
measurement. Every measurement are performed in the anechoic chamber using a Vector Network Analyzer (VNA).
Radiation patterns, S parameters and gain of the prototype are
presented. Each one will be compared with the simulations
presented in section III-B2.
Fig. 14: S parameters measurement.
Figure 14 shows the VNA and the prototype well isolated to
avoid reflections. It was used a VNA device to do the calibration automatically and during |S11 | and |S22 | measurements, a
charge of 50 Ω was used to avoid coupling between antennas
(first in second array SMA connector and second in the first
array SMA connector, respectively).
Figure 15 shows |S11 | and |S22 | obtained from the simulations as well as experimental.
Fig. 15: Comparison of simulation and experimental |S11 | and
|S22 | results.
Red and black curves represent the simulations while blue
and magenta represent the measurements.
The results on figure 15 show that there is a shift on the
frequency. |S11 | and |S22 | should overlap and they are a bit
shifted. This may happen because during the chemical process,
overetching may occur and consequently, the antenna becomes
resonating at a higher frequency. This will be explained in
detail in section V.
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comparison with gain of the standard horn and using the Friis
equation,
GainAU T = GHORN − P rHORN + P rAU T
Fig. 16: Comparison of simulation and experimental |S12 |
results.
In figure 16, red and black curves result from simulations
while blue and magenta result from measurements. The results
16 show a similarity between the measurements and simulations. |S12 | is below -50 dB which is a very good value. It
shows that there is no coupling between the two arrays.
2) Gain: All the measurements were done with an horn
antenna Model 08240-10 which has a nominal gain of 10 dB.
The following table shows how the gain varies with frequency.
FREQ
[GHz]
1,72
1,80
1,90
2,00
2,10
2,20
2,30
2,40
2,50
2,60
2,61
(15)
Array 1 has a gain of 11.84 dBi while array 2 has a gain of
11.51 dBi. Note that the accuracy of the standard horn gain is
±25 dB. As the gain obtained in the simulations is 12.04 dBi
it may be concluded that there is a good agreement between
simulation and experimental results.
3) Radiation patterns: In this section, radiation patterns
from the prototype are presented. Each measurement is explained through a diagram that describes the setup inside the
anechoic chamber. Since the antenna arrays have different
resonant frequencies, the radiation pattern will be measured
in a frequency that is the average between the two resonating
frequencies obtained in the measurements (2.239 GHz).
In order to measure the H plane (φ = 90◦ , θ =
[−180◦ , 180◦ ]), the setup is configured according to figure 18
inside the anechoic chamber. The prototype will rotate 360◦
around the positioner. The horn antenna is positioned with
angles indicated in the figure. Figure 19 shows the radiation
patterns from array 1 and figure 20 shows the radiation patterns
from array 2 obtained from simulations and measurements.
GAIN dB
(acc. +/- 0.25 dB)
8,4
8,8
9,2
9,6
10,0
10,4
10,7
11,1
11,4
11,7
11,7
TABLE II: Standard horn gain.
Fig. 18: Setup for H plane measurement.
The 3dB Beamwidth for E and H Planes at low frequency
is approximately 63◦ while it is at high frequency is approximately 48◦ .
Figure 17 shows the curve that relates gain and frequency.
Fig. 19: H plane radiation patterns from array 1 (simulated
and measured).
Fig. 17: Horn 08240-10 gain calculation.
The gain of horn antenna for the resonant frequency (2.239
GHz) is 10.52 dBi. The gain of the array was obtained by
Figures 19 and 20 presents radiation patterns in H plane.
Note that blue curve represents the simulation and red curve
the measurement. The results are very good. There is a really
good agreement between simulated and measured results. H
EXTENDED ABSTRACT FOR OBTAINING THE MASTER’S DEGREE IN ELECTRICAL AND COMPUTER ENGINEERING, OCTOBER 2014
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plane in array 1 simulated has a maximum gain of 12.04 dBi
while H plane in array 1 measured has a maximum gain of
11.83 dBi. H plane in array 2 simulated has a maximum gain of
12.05 dBi while H plane in array 2 measured has a maximum
gain of 11.4 dBi.
Fig. 23: E plane radiation patterns from array 2 (simulated and
measured).
Fig. 20: H plane radiation patterns from array 2 (simulated
and measured).
In order to measure E plane (θ = 90◦ , φ = [−180◦ , 180◦ ]),
the setup is configured according to figure 21 inside anechoic
chamber. The prototype will rotate 360◦ . The horn antenna
is positioned with angles indicated in the figure. Figure 22
shows the radiation patterns from array 1 and figure 23 shows
the radiation patterns from array 2 obtained from simulations
and measurements.
Note that both arrays have better measured results in terms
of directivity than in simulations. E plane in array 1 simulated
has a maximum gain of 12.04 dBi while E plane in array 1
measured has a maximum gain of 11.74 dBi. E plane in array
2 simulated has a maximum gain of 12.05 dBi while E plane
in array 2 measured has a maximum gain of 11.62 dBi.
V. C ONCLUSIONS AND F UTURE W ORK
This section highlights the main results and conclusions of
the work done and presents some suggestions for future work.
A. Achievements
Fig. 21: Setup for E plane measurement.
Fig. 22: E plane radiation patterns from array 1 (simulated and
measured).
Figures 22 and 23 present radiation patterns in E plane. The
results are also very good. There is a really good agreement
between simulated and measured results.
The main objective of this thesis was to study microstrip
patch antennas with cylindrical geometry to be used in base
stations of mobile communication systems. Depending on the
traffic profile of the cell, three different scenarios have been
considered:
1) omnidirectional antenna for a single sector antenna;
2) 180◦ two sectors antenna;
3) 120◦ three sectors antenna.
For the structures 1 and 3 only proof of concept has been
addressed. It was shown that an almost omnidirectional radiation pattern can be obtained with a printed ring ”wraparound”
element printed around a cylinder with 10 mm radius. It was
also shown that with a larger cylinder (70 mm radius) it is
possible to cover each of the three sector (120◦ each) with 3
arrays, each pointing to the middle of the its sector. Structure
number 2 has been designed, optimized and realidated with
experimental results.
The prototype produced consists in two arrays diametrically
oppose. Its fabrication is a complex process because of RT
Duroid 5880 rigidity. It was used a PVC plumbing pipe of 55
mm of surface curvature radius to be a form. This material
does not interfere with the electromagnetic characteristics
of the antenna. Since the substrate (RT Duroid 5880) is a
very rigid material, the pipe was sanded and drilled in all
side area in order to create some spikes during the collage
creating an extra friction. This sanding procedure may have
changed surface curvature radius which can be crucial in such
a sensitive antenna.
During simulations, it was considered a perfect cylinder
which in practice did not happen because it was used two
EXTENDED ABSTRACT FOR OBTAINING THE MASTER’S DEGREE IN ELECTRICAL AND COMPUTER ENGINEERING, OCTOBER 2014
plates of substrate, each one with an array which lead to
some imperfections when the plates were glued. For example,
plates may not be perfectly diametrically oppose. Also during
simulations, to simulate the antenna in CST, it was considered
a 50Ω line transmission during simulations because CST does
not allow the user to build a port in an oblique plane. This line
of 50Ω was only an extension to the border of the cylinder
in order to obtain an horizontal or vertical port. It was erased
in AutoCAD and replaced by a hole in the middle of the
100Ω line transmission which was the entrance for the SMA
connector.
These imperfections on simulations and fabrication may
explain some discrepancy between measurements and simulations. The frequency shift in |S11 | and |S22 | parameters measurements are completely normal. During the etching process,
the copper patches could have been overetched resulting in
smaller dimensions and consequently have a higher resonance
frequency. Through equation (3) from section II-A1, it is
possible to estimate the length of copper to be added to each
patch in order to obtain the central frequency, 2.14 GHz. Using
2.239 GHz (Resonating central frequency) and the length of
the patch used during simulations, it is possible to calculate
the new length for each new patch. The estimation is around
2 mm. |S12 | is very low. It is below -50 dB. The gain is also
concordant with simulations. Array 1 presents 11.84 dBi and
array 2 presents 11.51dBi which is quite similar to simulations.
Radiation patterns have also shown a good agreement between
simulation and experimental results.
B. Future Work
Finally, the author believes that the thesis has met the main
objectives of the worked proposed. In the authors view, there
is a good base on this thesis for someone that decides to
study conformal microstrip antennas to be used in mobile communications systems’ base stations. However the following
unsolved problems: increased the (impedance) bandwidth of
the ”wraparound” antenna to cover the whole LTE1 band and
increase the (impedance) bandwidth of the 3 sectorial antenna
to cover the whole LTE1 band are good challenges for future
work.
Moreover, the design of an array of ”wraparound” elements,
to provide the adequate vertical plane radiation pattern, is also
an interesting problem that deserves a detailed analysis.
R EFERENCES
[1] C. A. Balanis, Antenna Theory. John Wiley and Sons, 2005.
[2] C. Balanis, Modern Antenna Handbook. John Wiley and Sons, 2008.
[3] W. F. Richards, “”microstrip antennas”,” in Antenna Handbook: Theory,
Applications and Design. Van Nostrand Reinhold Co., 1988, ch. 10.
[4] C. M. Krowne, “Cylindrical-rectangular microstrip antenna,” IEEE Transactions on Antennas and Propagation, vol. 31, no. 1, pp. 194–199,
January 1983.
[5] C. Krowne, “Cylindrical-rectangular microstrip antenna radiation efficiency based on cavity q factor,” IEEE Antennas Propagat. Int. Symp.
Soc. Dig., pp. 11–14, June 1981.
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