www.powerelectronics.com
MARCH 2013
Vol. 39, No. 3
TECHNOLOGY
®
THE ENGINEER’S SOURCE FOR POWER AND ENERGY EFFICIENCY DESIGN INFORMATION
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42V, 2µA IQ Low Dropout Switcher
750mA
Output Current
Start-Up and Dropout Performance
9
8
VOUT
LT3973
7
Voltage (V)
VIN
6
5
4
VIN
VOUT
530mV
3
2
1
0
Time (s)
530mV Maximum Dropout
®
The LT 3973 is the newest member of our growing family of ultralow quiescent current high voltage monolithic buck
regulators. It consumes only 1.8µA of quiescent current while regulating an output of 3.3V from a 12V input source. A high
efficiency switch is included on-chip along with the catch diode, boost diode and all necessary control and logic circuitry.
A minimum dropout voltage of 530mV is maintained when the input voltage drops below the programmed output voltage,
®
providing a regulated output to the downstream load. Its low ripple Burst Mode operation maintains high efficiencies at low
output currents while keeping output ripple below 10mVP-P.
Features
Efficiency Curve, VOUT = 5V
• Ultralow Quiescent Current:
1.8µA IQ at 12VIN to 3.3VOUT
90
• Integrated Boost and Catch Diodes
• Excellent Start-Up & Dropout
Performance
Efficiency (%)
• Input Voltage Range: 4.2V to 42V
www.linear.com/product/LT3973
VIN = 12V
1-800-4-LINEAR
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• Low Ripple Burst Mode Operation
Info & Free Samples
VIN = 24V
70
VIN = 36V
60
50
• 750mA Output Current
40
• Adjustable Switching Frequency:
200kHz to 2.2MHz
30
0
0.1
0.2
0.3
0.4 0.5
Load Current (A)
0.6
0.7
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All other trademarks are the property of their respective owners.
EDITOR’Sviewpoint
The Dreamliner Nightmare
T
HE CLOSEST most electrical engineers come
to chemistry is probably through courses they
took in college. Right now they are probably
glad they went into the electronics rather than
chemistry because chemists haven’t been able
to solve current problems with lithium-ion batteries. These
problems started with a fire on a Japan Airlines 787 in
Boston. A similar situation occurred a week before when
smoke poured from the same battery system on an All
Nippon Airways plane, forcing it to make an emergency
landing.
Following these events, the U.S. Federal Aviation
Administration (FAA) issued an emergency directive,
grounding all Boeing 787s operated by American airlines
until Boeing can prove the batteries are safe. Other country’s
regulators then followed suit.
“The battery failures resulted in release of flammable
electrolytes, heat damage and smoke on two 787 airplanes,”
the FAA said. “These conditions, if not corrected, could
result in damage to critical systems and structures, and the
potential for fire in the electrical compartment.”
The National Transportation Safety Board (NTSB) is
examining the lithium-ion battery from the 787 that caught
fire in Boston, along with “black box” data from the airplane.
NTSB currently believes the battery did not suffer an overcharging. Japanese investigators concluded the same thing
after looking at the 787 that made an emergency landing in
Japan.
The Dreamliner is the first airliner to rely on lithium-ion
batteries for electrical power. It has two of these, about 10
in wide, 14 in long and 8 in high; they weigh 63 pounds.
One is located in the rear electrical equipment bay, near
the wings. It is used to start the auxiliary power generator, a
small engine that is primarily used to power the plane when
it is on the ground. The second battery powers up the pilot’s
computer displays and serves as a back-up for flight systems.
Other airliners use traditional mechanical and hydraulic
systems that divert power from the engines to run electrical
equipment. Where batteries are used, they have been traditional ones such as lead-acid or nickel-cadmium. Compared
with the larger 777, the 787’s electrical system is five times
more powerful and can produce enough electricity to power
500 homes.
2
Power Electronics Technology | March 2013
Dr. Peter Harrop, Chairman, IDTechEx said the larger a
lithium-ion battery is, the more there is to go wrong. Those
making safe small versions for phones or tablets cannot
necessarily make safe big ones. To some extent, improved
temperature performance from different cathodes can correlate with improved safety, though no lithium-ion cell is
inherently safe.
Lithium-based batteries are popular because lithium is
the lightest of all metals, has the greatest electrochemical
potential and provides the largest energy density for weight.
The energy density of lithium-ion is typically twice that of
the standard nickel-cadmium cell. The load characteristics
are reasonably good and behave similarly to nickel-cadmium
in terms of discharge. Li-ion’s 3.6 V/cell allows battery packs
with only one cell. Most of today’s mobile phones run on a
single cell.
Li-ion has no memory (like NiCad) and no scheduled
cycling required to prolong battery’s life. And, its selfdischarge is less than half of nickel-cadmium.
Despite its advantages, lithium-ion has drawbacks. It is
fragile and requires a protection circuit to limit the peak
voltage of each cell during charge and prevent the voltage
from dropping too low on discharge. In addition, cell temperature is monitored to prevent temperature extremes.
Aging is a concern with most Li-ion batteries. Some
capacity deterioration is noticeable after one year, whether
the battery is in use or not. The battery frequently fails after
two or three years. Other chemistries also have age-related
degenerative effects, including NiMH when exposed to high
ambient temperatures.
Storage in a cool place slows the aging of Li-ion cells.
Manufacturers recommend storage temperatures of 15°C
(59°F). In addition, the battery should be partially charged
during storage; the recommendation is a 40% charge.
If the chemists can’t get it right, maybe electrical engineers can come up with a foolproof technique that safeguards the batteries and the associated equipment.
SAM DAVIS, Editor-in-Chief
www.powerelectronics.com
march 2013
Editorial
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March 2013 | Power Electronics Technology
3
FOR DESIGNERS AND SYSTEMS ENGINEERS
w w w. p o w e r e l e c t r o n i c s . c o m
MARCH 2013 • Vol. 39, No. 3
CoverStory
DESIGN FEATURES
CURRENT TRENDS
eGaN® FET-Silicon Power
14
Shoot-Out Vol. 13, Part 1:
LED Drivers for
Impact Of Parasitics
36
Incandescent Bulb
Replacement Is A BOM
PET INNOVATIONS
The latest in this series of articles on
eGaN FETs compares the effects of parasitic inductance on performance of eGaN
FETs and MOSFETs in point-of-load
buck converters switching at 1 MHz.
Designing R2CD
24
Snubbers Using Standard
Recovery Diodes
By adding an extra resistor, the R2CD
configuration improves performance
across the board, lowers cost, and
increases immunity against variations in
circuit parameters.
Layout Power Supply
28
Boards to Minimize EMI Part 3: EMI Basics
Challenge
LED replacement of incandescent lighting for residential use involves IC design
that requires “penny pinching” to a
degree not found in many other areas.
DEPARTMENTS
2 EDITOR’S VIEWPOINT
6 INDUSTRY HIGHLIGHTS
38 NEW PRODUCTS
39 PATENTS
40 PRODUCT MARKETPLACE
40 ADVERTISER INDEX
Good layout from first prototyping on
actually saves significant resources in
EMI filters, mechanical shielding, EMI
test time and PC board runs.
IC Pair Improves
33
Transmitting And
Receiving Of Wireless Power
8
p.
Recently-introduced wireless power
receiver and transmitter ICs comply with
the Wireless Power Consortium (WPC)
1.1 standard.
On-Chip MCU Supervises
Operation
Of Power Conversion
Controller
IC
By Sam Davis, Editor-in-Chief
A hybrid, mixed-signal controller IC features
analog pulse-width modulation (PWM)
current mode operation with an integrated
microcontroller (MCU).
COVER DESIGN: Anthony Vitolo
4
Power Electronics Technology | March 2013
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INDUSTRY
Polymer Film Harvests Energy
M
IT ENGINEERS have created a new polymer film that
can generate electricity by drawing on a ubiquitous
source: water vapor.
The new material changes its shape after absorbing tiny
amounts of evaporated water, allowing it to repeatedly curl
up and down. Harnessing this continuous motion could drive
robotic limbs or generate enough electricity to power microand nanoelectronic devices, such as environmental sensors.
“With a sensor powered by a battery, you have to replace
it periodically. If you have this device, you can harvest
energy from the environment so you don’t have to replace
it very often,” says Mingming Ma, a postdoc at MIT’s David
H. Koch Institute for Integrative Cancer Research and lead
author of a paper describing the new material in the Jan. 11
issue of Science.
“We are very excited about this new material, and
we expect as we achieve higher efficiency in converting
mechanical energy into electricity, this material will find
even broader applications,” says Robert Langer, the David
H. Koch Institute Professor at MIT and senior author of
the paper. Those potential applications include large-scale,
water-vapor-powered generators, or smaller generators to
power wearable electronics.
Other authors of the Science paper are Koch Institute
postdoc Liang Guo and Daniel Anderson, the Samuel A.
Goldblith Associate Professor of Chemical Engineering and
a member of the Koch Institute and MIT’s Institute for
Medical Engineering and Science.
The new film is made from an interlocking network of
two different polymers. One of the polymers, polypyrrole,
forms a hard but flexible matrix that provides structural
support. The other polymer, polyol-borate, is a soft gel that
swells when it absorbs water.
Previous efforts to make water-responsive films have used
only polypyrrole, which shows a much weaker response on
its own. “By incorporating the two different kinds of polymers, you can generate a much bigger displacement, as well
as a stronger force,” Guo says.
The film harvests energy found in the water gradient between dry and water-rich environments. When the
20-micrometer-thick film lies on a surface that contains even
a small amount of moisture, the bottom layer absorbs evaporated water, forcing the film to curl away from the surface.
Once the bottom of the film is exposed to air, it quickly
releases the moisture, somersaults forward, and starts to
curl up again. The continuous motion converts the chemical
energy of the water gradient into mechanical energy.
Such films could act as either actuators (a type of motor)
or generators. As an actuator, the material can be surprisingly
powerful: The researchers demonstrated that a 25-milligram
film can lift a load of glass slides 380 times its own weight,
or transport a load of silver wires 10 times its own weight.
GLOBAL MARKET FOR THERMAL MANAGEMENT TO REACH $10.1 BILLION IN 2017
A
ccording to a new technical market
research report, The Market For
Thermal Management Technologies
(SMC024H) from BCC Research, the market for thermal management technologies
was valued at $6.7 billion in 2011 and
reached $7 billion in 2012. Total market
value is expected to reach $10.1 billion in
2017, increasing at a five-year compound
annual growth rate (CAGR) of 7.6%.
Thermal management hardware
(e.g., fans and blowers, heat sinks, etc.)
account for about 80% of the total thermal
management market. The other main thermal management product segments -software, interface materials, and substrates
- each account for between 5% and 7% of
6
the market.
The largest end markets for thermal
management products in 2011 were
the computer industry (50.8% of total
revenues), telecommunications (16.8%),
and medical/office equipment (11.5%).
Computers are projected to increase their
market share to 57.1% by 2017, while
telecommunications applications’ share
drops to 13.6%. Medical/office equipment’s share should remain steady at
around 11.5%.
The North American market should
maintain its number one position throughout the period under review, with a market
share of about 37%, followed by AsiaPacific with around 25%. The Asia-Pacific
Power Electronics Technology | March 2013
countries (except Japan) are not only the
second-largest market in absolute terms,
but they also have the highest projected
growth rate (i.e., a CAGR of 9% between
2012 and 2017).
The study includes these aspects:
• Identifies thermal management technologies and products with the greatest
commercial potential in the near to midterm (2012-2017)
• Analyzes the key drivers and constraints
that will shape the market for thermal
management technologies and products
over the next five years
• Estimates the current and future
demand for thermal management technologies and products
www.powerelectronics.com
designfeature
SAM DAvIS, Editor-in-Chief, PET
On-Chip MCU Supervises Operation
Of Power Conversion Controller IC
U
sing traditional analog control circuits,
A hybrid, mixed-signal controller IC
the
MCP19111
features analog pulse-width modulation
from
Microchip
(PWM) current mode operation with an
Technology regulates the output of
integrated microcontroller (MCU).
a synchronous, stepdown dc/dc converter with a 4.5 to 32 V input range. Also
on-chip is a version of Microchip’s PIC® MCU mid-range core
that enables customization of device operating parameters,
start-up and shut down profiles, protection levels and fault
handling procedures. The MCP19111 (Fig. 1.) is housed in a
space-saving, 28-pin, 5 mm x 5mm QFN package with integrated synchronous drivers, an internal linear regulator, and 4
kW flash memory.
To complete a full-fledged power management system,
Microchip developed a high-speed MOSFET family, optimized specifically for use with
the MCP19111. The family includes the MCP
87018, MCP87030,
MCP87090 and MCP
87130. They are 25
V-rated, 1.8 mΩ, 3 mΩ, 9
mΩ and 13 mΩ logic-level MOSFETs. MCP87030
and MCP87018 MOSFETs
are offered in a 5 x 6 mm,
8-pin PDFN package.
The
MCP87090 and MCP87130
MOSFETs are offered in both a
5 x 6 mm, 8-pin PDFN package,
as well as a 3.3x3.3 mm, 8-pin
PDFN package
Digitally-controlled supervision of power management allows
MCU register settings to configure
the device, rather than adding
or modifying external hardware.
The MPC19111’s low power, 8-bit
MCU easily performs all necessary
supervisory functions. Supervision
programmability permits the
8 Power Electronics Technology | March 2013
www.powerelectronics.com
Mcp19111 Digitally Enhanced Power Converter
Analog Control And Power Stage
8-Bit
MCU
designer to optimize efficiency, protect the analog section, provide 15
general purpose I/O connections, and include
some established PMBus
instructions using an I2C
interface.
VIN
Internal Bias Supply
Slope Compensation
VREF
Current Sense
Microchip
MCP87XXX
PWM
Generator
Synchronous
Mosfet
Driver
GPIO
Synchronous
Buck Topology
Mosfet Family
VOUT
Microchip
MCP87XXX
Error Amp
Comm
Mosfet Family
Internal Power
Interface
Adjustable
Fig. 2 shows a typiCompensation
Network
cal application for the
MCP19111, which has
two internal linear regulators that generate 5 V Fig. 1. Block diagram of the MCP19111 shows the integrated MCU and controller IC.
rails. One is an on-chip 5
V rail that powers internal analog circuits. Located at the
VDD pin is a second 5 V rail that powers the MCU.
IDRIVE = [QG(High) + QG(LOW)] ×FSW
The internal gate drivers can drive two external
N-Channel MOSFETs in a synchronous buck topology.
Where:
The gate of the floating MOSFET is connected to the
IDRIVE = Drive current in A
HDRV pin. The source of this MOSFET is connected to
QG(HIGH) = Total gate charge of the high-side
the PHASE pin. The HDRV pin source and sink current is
MOSFET in nC
configurable. Setting a bit in an internal register allows the
QG(LOW) = Total gate charge of the low-side
high-side to source and sink 1 A peak current. By clearing
MOSFET in nC
this bit, the source and sink peak current is 2 A.
FSW = Switching frequency in MHz
The low-side MOSFET gate connects to the LDRV pin
Synchronous MOSFET dead time occurs when one
and the source of this MOSFET connects to PGND. The
drive signal goes low and the complimentary drive signal
drive strength of the LDRV pin is not configurable. This
goes high. The MCP19111 can adjust both the high-side
pin can source 2 A of peak current and peak sink current is
and low-side driver dead times independently by using
4 A. This helps keep the low-side MOSFET off when the
register settings that enable 4 ns increments.
high-side MOSFET turns on.
The MCP19111 can disable the entire synchronous
Current required to drive the external MOSFETs is:
driver or just one side of the synchronous drive signal. A
register setting disables the
VIN
entire synchronous driver
GPA6
when the HDRV and LDRV
VIN
GPB1
signals are set low and the
PHASE pin floats. Clearing
GPA0
HDRV
TRACK
BOOT
the disable bit allows normal
GPA2
PGOOD
+VOUT
PHASE
operation.
GPA3
CNTL
Register settings also conLDRV
GPB7
ADDR1
V
figure
the output voltage,
DD
–VOUT
MCP19111
ADDR0
GPB6
eliminating the need for an
VDR
SYNC
GPA1
external resistor divider to
+ISEN
SMBUS ALERT GPA4
set the output voltage. Plus,
–ISEN
SCL
the MCP19111 contains a
GPA7
+VSEN
SDA
unity gain differential ampliGPB0
–VSEN
fier used for remote sensICCDAT
PGND
GPB4
ing of the output voltage.
MPLAB x ICD
ICDCLK
GPB5
GND
PROGRAMMER
Connecting the amplifier’s
MCLR
GPA4
+VSEN and -VSEN pins directly at the load allows better
Fig. 2. typical MCP19111 application drives two external synchronous MoSFets .
load regulation.
www.powerelectronics.com
March 2013 | Power Electronics Technology
9
POWERcontroller
ICs
You can configure
the chip’s
+
–
switching
fre–VSEN
–
+
quency
from
100 kHz to 1.6
VREF
MHz. A timer
Fig. 3. internal compensation network with the output module
generdifferential amplifier.
ates the HDRV/
LDRV switching
frequency. Typically, the MCP19111 controller IC is set
to operate at a 300 kHz switching frequency.
A register setting adjusts the controller IC’s compensation zero frequency and gain. Fig. 3 shows the internal
compensation network with the output differential amplifier showing the +VSEN and –VSEN pins.
In current mode control systems, slope compensation
must be added to the PWM circuit to prevent subharmonic oscillation when operating with duty cycles greater
than 50%. The MCP19111 adds a negative slope to the
error amplifier output signal before it is compared to the
current sense signal. The amount of slope added is controlled by a register.
Output current sensing can use either a resistor placed
in series with the output, or the series resistance of the
inductor. Using an inductor series resistance requires a
filter to remove the large AC component of the voltage
that appears across the inductor and leave only the small
AC voltage that appears across the inductor resistance. Fig.
4 shows the inductor current sense filter. You can find RS
and CS from:
+VSEN
(2)
Where:
L = Inductance of the output inductor in henries
RL = Series resistance of the output inductor in ohms
RS = Current sense filter resistor in ohms
CS = Current sense filter capacitor
When the current sense filter time constant is set equal
to the inductor time constant, the voltage appearing across
CS approximates the current flowing in the inductor multiplied by the inductance.
Protection Features
An analog-to-digital Converter (ADC) allows conversion
of an analog input signal to its 10-bit binary representation. The ADC uses analog inputs, which are multiplexed
into a single sample and hold circuit. The output of the
sample and hold connects to the controller input. The
controller generates a 10-bit binary result via successive
approximation and stores the right justified conversion
result in the ADC result registers.
You can monitor the output voltage measured between
10 Power Electronics Technology | March 2013
the +VSEN and -VSEN pins using the internal ADC. When
this ADC reading matches a user-defined power good
value (determined by firmware), it can toggle a GPIO
(general purpose I/O) to indicate the system output voltage is within a specified range. You can use firmware to
configure delays, hysteresis and time-out values.
To facilitate system prototyping, various internal signals
can be measured by configuring the MCP19111 in bench test
mode. To accomplish this, the ATSTCON<BNCHEN>
bit is set. This configures GPA0 as the ANALOG_TEST
feature. The ADC/multiplexer provides access to measurement of the internal analog signals.
The MCP19111 features a hardware overtemperature
shutdown protection set at +160°C typically. No firmware
fault-handling procedure is required to shutdown the
MCP19111 for an overtemperature condition. Typically,
if the internal temperature of the MCP19111 reaches
+140°C the MCP19111 clears a register bit. This bit
remains cleared until set by firmware.
Firmware can control soft start of the output voltage.
Internal registers settings can produce very long soft start
times.
You can configure the MCP19111 to track another
voltage signal at start-up or shutdown. The ADC can read
a GPIO that has the desired tracking voltage applied to
it. Then, firmware compares the internal output voltage
reference to this ADC reading.
Register settings and flags allow the designer to configure other MCP19111 functions, including:
• Input Under-Voltage Lockout
• Output Over-Current
• Current Sense AC Gain
• Output Compensation
• Slope Compensation
• Output Voltage Configuration
• Output Under-Voltage
• Output Over-Voltage
Analog Peripheral Control
You can configure various analog peripherals that enable
customizable operation.
• Diode Emulation Mode
• High-Side Drive Strength
• MOSFET Driver Dead Time
• Output Voltage Sense Pull-Up/Pull-Down
• Output Under-Voltage Accelerator
• Output Over-Voltage Accelerator
Analog Blocks Enable Control
Using enable bits located in the ATSTCON register you
can enable or disable various analog circuit blocks:
• Output Over-Voltage Enable
• Output Under-Voltage Enable
• Relative Efficiency Measurement Control
• Slope Compensation Control
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POWERcontroller
ICs
VIN
• Current Measurement Control
• Internal Temperature Measurement Control
• Relative Efficiency Circuitry Control
• Signal Chain Control
Device Calibration
Calibration words are stored in flash program memory
that is readable and writable during normal operation
(full VIN range). This memory is not directly mapped in
the register file space, but is indirectly addressed through
the Special Function Registers used to read and write this
memory. Read-only memory locations 2080h through
208Fh contain factory calibration data, including:
• Offset Calibration for The Output Voltage Remote
Sense Differential Amplifier
• Overtemperature Shutdown Threshold
• Internal Bandgap Voltage Reference for the ADC
• Buffer Amplifier Offset of the Output Voltage Regulation
Reference Set Point
• Error Amplifier Offset
• ADC Reading From The Internal Temperature Sensor
• Output Voltage Difference Amplifier Offset
• Unity Gain Buffer Offset Voltage
Memory Organization
• There are two types of memory:
• Program Memory
• Data Memory
• Special Function Registers (SFRs)
• General Purpose RAM
The MCP19111 has a 13-bit program counter capable
of addressing an 8K x 14 program memory space. Only the
first 4K x 14 (0000h-0FFFh) is physically implemented.
Special Function Registers and peripheral functions
control the MCU’s desired operation. These registers are
static RAM.
In the MCP19111, the PWM module generates the
system clock, which controls the switching frequency and
sets the maximum allowable duty cycle. The PWM module does not continuously adjust the duty cycle to control
the output voltage. The analog control loop and associated
circuitry performs the adjustment.
Instruction Set
The MCP19111 instruction set is highly orthogonal and
has three basic categories:
• Byte-oriented operations
• Bit-oriented operations
• Literal and control operations
Each instruction is a 14-bit word divided into an
opcode, which specifies the instruction type, and one or
more operands, which further specify instruction operation. One instruction cycle comprises four oscillator periods. For a 4 MHz oscillator frequency this yields a normal
instruction 1 μs execution time. All instructions execute
within a single instruction cycle, unless a conditional test
12 Power Electronics Technology | March 2013
–ISEN
+ISEN
RS
CS
HDRV
PHASE
To Load
L
RL
LDRV
Fig. 4. Current filter removes the AC component of the inductor voltage.
MCP19111 evaluation Board includes the IC and external MoS.
is true, or the program counter is changed as a result of an
instruction.
MPLAB IntegrAted deveLoPMent
A Graphical User Interface (GUI) aids in developing
MCP19111 firmware. You can use this GUI to quickly
configure the MCP19111 for basic operation. You can
also add customized or proprietary features to the GUIgenerated firmware. The MCP19111 also features firmware debug support.
Also available is the MCP19111 Evaluation Board
(ADM00397) that includes Microchip’s power MOSFETs
(Fig. 5). Offered with standard firmware, the evaluation
board is user-configurable through the MPLAB® X IDE
GUI plug-in. The combined evaluation board, GUI and
firmware allow power-supply designers to configure and
evaluate the performance of the MCP19111 for their
target applications.
The MPLAB IDE software brings an ease of software
development not usually seen with 8/16/32-bit MCU
applications. The MPLAB IDE is a Windows® operating
system-based application containing:
• A single graphical interface to all debugging tools
• Simulator
• Programmer (sold separately)
• In-Circuit Emulator and Debugger (sold separately)
• A full-featured editor with color-coded context
• A multiple project manager
• Customizable data windows with direct edit of contents
• High-level source code debugging
• Mouse over variable inspection
• Drag and drop variables from source to watch windows
• Extensive on-line help
• Integration of third party tools, such as IAR C Compilers.
The MPLAB IDE allows you to:
• Edit your source files (either C or assembly)
• One-touch compile or assemble, and download to
emulator and simulator tools (automatically updates all
project information)
• Debug using either source files (C or assembly), Mixed
C, and assemblyMachine code
www.powerelectronics.com
designfeature
DaviD Reusch, Ph.D., Director, Applications, Efficient Power Conversion Corporation
eGaN® FET-Silicon Power Shoot-Out
Vol. 13, Part 1: Impact Of Parasitics
eGaN® FETs have shown
their ability to achieve higher
efficiencies and switching
frequencies than possible
with silicon MOSFETs in this
series of articles. This article
studies the effect of parasitic
inductance on performance
of eGaN FETs and MOSFETs in
point-of-load buck converters
switching at 1 MHz.
I
n a traditional hard switching transition, the switching losses are impacted by
two device parameters, QGD, known as the Miller charge, which controls the
voltage rising and falling speed; and QGS2, which is the portion of the gate to
source charge from the device threshold voltage to the gate plateau voltage,
which controls the current rising and falling speed. The turn off period, shown
in Fig. 1a, begins with a decrease of gate drive voltage; when the gate to source
voltage reaches the plateau, the voltage across the device will begin to rise, being
driven by the gate current, IG.
During the voltage rising period, the device encounters both current and voltage in the device, resulting in switching loss. For the voltage rising period, the
device parameter determining loss is QGD. When the device voltage reaches the
input voltage, the current in the device will begin to fall and more switching loss
in the device will be incurred. For the current falling period, the device parameter
determining loss is QGS2. The power loss during the turn off switching transition
can be given by:
(1)
Where:
VIN =Input voltage (V)
IOFF = Turn-off current (A)
QGD = Miller charge (C)
QGS2 = Gate-to-source charge from device threshold voltage to gate
plateau voltage (C)
IG = Gate current (A)
For the turn on switching losses, the same principles apply, as shown in Fig. 1b.
Minimizing the QGD and QGS2 parameters decrease switching losses incurred in a
hard switching application. The turn on loss is given by:
(2)
TVR
VIN
TCF
TCR
VIN
VDS
TVF
VDS
IDS
IOFF
IDS
VGS
VGS
VPL
VTH
VPL
VTH
(a)
ION
QGD
QGS2
T
(b)
QGS2
QGD
T
Fig. 1. Ideal hard switching showing: (a) Turn-off transition and (b) Turn-on transition.
14 Power Electronics Technology | March 2013
www.powerelectronics.com
FOM = (QGD + QGS2) * RDS (ON) (nC*Ω)
35
40 V MOSFETs
30
Where:
ION = turn on current (A)
25
Figure of merit (FOM) is widely
QGS2
20
QGS2
used to compare the performance
[5]
of competing power devices .
15
The FOM of hard switching appli25 V
40 V
cations such as a synchronous buck
MOSFETs
10
eGaN FET
converter is defined as the product
QGS2
QGD
QGS2
QGD
QGS2
of the dynamic loss parameters,
5
QGD
QGD
QGD
QGD + QGS2, and static losses,
0
RDS(ON). When comparing 40 V
BSZ097N04LSG BSZ040N04LSG BSZ060NE2LS BSZ036NE2LS
EPC2015
eGaN FETs to 40 V MOSFETs
currently on the market, eGaN Fig. 2. 40 V device Figure of Merit comparison with VDS = 12 V, IDS = 20 A.
FETs offer a significant reduction
in FOM, as shown in Fig. 2. For designs requiring lower
To evaluate the performance of the 40 V eGaN
input voltages, for example a 12 V input buck converter,
FET against different combinations of 40 V and 25 V
lower voltage MOSFETs can be utilized. The FOM of a
MOSFETs, similar power loop designs were tested for the
25 V Si MOSFET is comparable to a higher rated 40 V
eGaN FET and MOSFETs. Fig. 3 shows the PCB layout for
eGaN FET.
the designs with the high frequency loop highlighted in red
From an FOM comparison, the eGaN FET should
in Fig. 3a. This conventional PCB layout places the input
achieve higher efficiency than the equally rated 40 V
capacitors and devices on opposite sides of the PCB, with
MOSFET devices and similar efficiency to the 40% lower
the capacitors being located directly underneath the devicrated 25 V MOSFETs. In practical applications, FOM
es to minimize the physical loop size, leading to reduced
is just one of the contributors to
achieving higher efficiency. The
others are die size optimization [6],
package parasitics, and PCB layout
parasitics. To enable the high
switching speed available from low
FOM, low parasitic packaging and
PCB layout is required. eGaN
FETs were developed in advanced
land grid array (LGA) packages
(a)
(b)
(c)
that not only have low internal
inductance, but enable the user to Fig. 3. Conventional vertical power loop PCB layouts including: (a) side view showing high frequency loop in red,
design ultra-low inductance into (b) MOSFET layout top view, and (c) eGaN FET layout top view.
their board. This shootout will
cover the impact of package and
CONFiGuraTiON
TOp SwiTCh parT NO.
SyNChrONOuS rECTiFiEr parT NO.
PCB layout parasitics on converter
eGaN FETs
EPC2015
EPC2015
performance and compare the in
40 V MOSFETs
BSZ097N04LSG
BSZ040N04LSG
circuit performance of eGaN FET
25 V MOSFETs
BSZ036NE2LS
BSZ036NE2LS
and MOSFET devices.
Table: Device parameters for 40 V eGaN FET 40 V MOSFETs, and 25 V MOSFETs
CONFiGuraTiON
TOp SwiTCh
(QGD+QGS2) (NC)
rDS(ON)
(mΩ)
FOM (pC-Ω)
SyNChrONOuS rECTiFiEr
rDS(ON) (mΩ)
40 V eGaN FET
EPC2015
2.4#
3.2
7.6
EPC2015
3.2
40 V MOSFET Pair 1
BSZ097N04LSG
3.3#
8.2*
27.0
BSZ040N04LSG
3.4*
40 V MOSFET Pair 2
BSZ040N04LSG
9.0#
3.4*
30.5
BSZ040N04LSG
3.4*
25 V MOSFET Pair 1
BSZ060NE2LS
1.6#
5.1*
8.3
BSZ036NE2LS
3.0*
25 V MOSFET Pair 2
BSZ036NE2LS
2.7#
3.0*
8.1
BSZ036NE2LS
3.0*
# QGD at 12 V, QGS2 at 20 A
www.powerelectronics.com
* MOSFETs driven at 8 V
March 2013 | Power Electronics Technology
15
eGaN
fets
EFFICIENCY (%)
PCB parasitic inductance. Space is left in between the
devices for the switching node connection which is connected to the output inductor in a buck converter.
The eGaN FET and MOSFET prototypes, shown in
Figs. 3b and 3c, used similar part layouts and identical board
91
90
89
88
87
86
85
84
83
82
81
80
79
78
77
76
builds to ensure the comparison would only be influenced
by the devices. For the MOSFET device, the smallest
package was chosen, a 3 x 3 mm TSDSON-8, to compare
against the eGaN FET 4.1 x 1.6 mm LGA package. For
the drivers, the eGaN FET used the LM5113, designed to
meet the driving requirements of the eGaN FET, while the
MOSFET employed an ISL2111 MOSFET driver.
TOP
SWITCH
LS
L
L LOOP
40 V eGaN FET
40 V MOSFET Pair 1
40 V MOSFET Pair 2
25 V MOSFET Pair 1
25 V MOSFET Pair 2
2
4
6
8
10
12
14
16
18
20
CIN
SYNCHRONOUS
RECTIFIER
DRIVER
COUT
22
OUTPUT CURRENT (IOUT)
Fig. 4. Comparing efficiency of 40 V eGaN FET and 40 V and 25 V silicon
MOSFETs at VIN = 12 V, VOUT = 1.2 V, FSW = 1 MHz, LBUCK = 300 nH.
Fig. 5. Synchronous buck converter showing parasitic inductances.
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Power Electronics Technology | March 2013
www.powerelectronics.com
eGaN
fets
BS
SOURCE CONTACTS
SU
DRAIN CONTACTS
POWER LOSS (W)
GA
TE
TR
AT
E
For the device comparisons, MOSFETs with similar
FET provided higher efficiency than all of the benchon resistance were selected for the synchronous rectifiers
mark MOSFETs. The 40 V eGaN FET can outperform
and two different criteria were used
to compare the top switch. The first
criterion for the MOSFET top switch
5.5
selection was to minimize dynamic
loss parameters, QGD + QGS2, in an
5
effort to offer the lowest switchLS
MOSFET
ing losses at higher switching frePACKAGING
4.5
quencies. The second criterion for
LLOOP
MOSFET top switch selection was
4
eGaN FET
to select similar on resistance charPACKAGING
acteristics as the 40 V eGaN FET, to
3.5
offer similar conduction losses. The
selected devices and their characteris3
tics are shown in the table.
0 0.3 0.6 0.9 1.2 1.5 1.8 2.1 2.4 2.7 3
Fig. 4 compares the efficienPARASITIC INDUCTANCE (nH)
(a)
(b)
cy of the 40 V eGaN FET, 40 V
MOSFETs, and 25 V MOSFETs,
with the device parameters for these Fig. 6. Parasitic Inductance vs Power Loss for eGaN and MOSFET packaging showing (a) Impact on power loss
parts shown in the table. At a switch- VIN = 12 V, VOUT = 1.2 V, IOUT = 20 A, FSW = 1 MHz, top switch EPC2015, synchronous rectifier EPC2015, and
ing frequency of 1 MHz, the eGaN (b) the eGaN FET land grid array package.
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18
Power Electronics Technology | March 2013
www.powerelectronics.com
®
eGaN
fets
the best pair of 25 V MOSFETs
by almost 1% and the best pair of
40 V MOSFETs by almost 3%. To
explain the performance gains of
the eGaN FET over the MOSFETs
with similar FOM and a more optimized die size selection, the influence of package parasitics and PCB
layout parasitics must be considered.
For the eGaN FETs, switching losses
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20
Power Electronics Technology | March 2013
can be reduced at higher frequencies
by using the EPC2014 for the top
switch, an eGaN FET with a smaller
switching charge.
CONVERTER PARASITICS
Previously, it was shown that devices
with similar switching charges and on
resistances performed differently in
circuit. This can be seen by the greater
than one percent difference in efficiency between the 40 V eGaN FET
and the lower rated 25 V MOSFET
pair 2. The reason for the improved
performance of the eGaN FET when
compared to a MOSFET with similar
characteristics is the use of superior
eGaN FET packaging. In a practical
buck converter, there are two major
parasitic inductances (see Fig. 5),
which have a significant impact on
converter performance:
The first is common source inductance, LS, the inductance shared by
the drain to source power current
path and gate driver loop.
The second is high frequency
power loop inductance, LLOOP,
which is the power commutation
loop and comprised of the parasitic
inductance from the positive terminal of the input capacitance, through
the top device, synchronous rectifier,
ground loop, and input capacitor.
The common source inductance,
Ls, has been shown to be critical
to performance because it directly
impacts the driving speed of the
devices [7]-[9]. The common source
inductance is mainly controlled by
the package inductance of the device
and varies from package to package
[10], [11].
For the eGaN FET, the
LGA package (Fig. 6b) offers low
common source inductance, reducing loss, as shown in Fig. 6a.
The high frequency loop inductance, LLOOP, impacts the switching commutation time and the peak
drain to source voltage spike of the
devices. The high frequency loop
inductance is controlled by the PCB
layout and package inductance. In
www.powerelectronics.com
LOOP INDUCTANCE (nH)
applications utilizing low package parasitics, e.g.
TOP LAYER
eGaN FETs, the PCB layout dominates the high
[12]-[15].
INNER LAYER 1
BOARD
frequency loop inductance
THICKNESS
The impact of parasitic inductance on power
INNER LAYER 2
loss for an eGaN FET based buck converter is calBOTTOM LAYER
(a)
culated and shown in Fig. 6a [16], it can be seen that
by introducing common source and high frequency
3.5
loop inductance the loss increases. Understanding
Si MOSFET
3
the impact of parasitic inductance on performance,
VERTICAL LOOP
the eGaN FET made the reduction of package
2.5
parasitics a high priority. For the eGaN FET, a
eGaN FET
2
device with a higher voltage lateral structure, all
VERTICAL LOOP
of the connections are contained on the same side
1.5
of the die. This allows for the die to be mounted
1
directly to the PCB, minimizing the total parasitics
to the internal bussing and external solder bumps.
0.5
To further decrease parasitics, the drain and source
0
connections are arranged in an interleaved LGA
50
40
20
30
60
70
package, providing multiple parallel connections to
BOARD THICKNESS (mil)
(b)
the PCB from the die. The result is a device package inductance in the range of a couple hundred Fig. 7. Vertical power loop design: (a) PCB cross section and (b) simulated high frequency
picohenries [5],[11].
loop inductance vs board thickness.
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March 2013 | Power Electronics Technology
21
egan
fets
91
90
22 Power Electronics Technology | March 2013
EFFICIENCY (%)
89
PCB Layout
With the significant reduction in package related inductance provided by the eGaN FET, the common source
inductance is minimized and is no longer the major contributor to parasitic loss. Instead, it is the high frequency
loop inductance, controlled by PCB layout, thus making
layout using the eGaN FETs critical to high frequency
performance.
To compare the performance of different PCB layouts for both eGaN FETs and MOSFETs, two different
board builds were considered based on the PCB layout
in Fig. 3. For the layout comparison, a 4 layer PCB was
used with two ounce copper on each layer and an overall
board thickness of 62 and 31 mils were tested. In the
conventional vertical power loop design, the loop inductance is heavily dependent on the board thickness as the
power loop is contained on the top and bottom layers of
the PCB. As the board thickness increases so does the
high frequency loop inductance, leading to higher losses
and consequently lower efficiency.
Fig. 7a shows the cross section of a circuit board, while
Fig. 7b shows the simulated high frequency loop induc-
88
87
86
85
40 V eGaN FET
40 V MOSFET
25 V MOSFET
31 MILS
62 MILS
84
83
82
81
80
2
4
6
8
10
12
14
16
18
20
22
OUTPUT CURRENT (IOUT)
Fig. 8. Comparing efficiency vs output current for 40 V eGaN FEt, 40 V MoSFEt,
and 25 V MoSFEtS with VIN = 12 V, Vout = 1.2 V, FSW = 1 MHz, and LBuCK = 300 nH.
tance of the eGaN FET and MOSFET based PCB designs.
Due to the eGaN FET’s smaller size and reduced package
parasitics, the loop inductance is reduced around 50%
when compared to the MOSFET design. For the eGaN
FET, the PCB layout dominates the loop inductance, with
the inductance increasing 80% when the board thickness
increases from 31 to 62 mils. As a result, the 62 mil board
designs all suffer an efficiency drop of at least 1% (Fig. 8).
www.powerelectronics.com
While figure of merit is an important metric in determining the best performing device, the package and layout
parasitics are also a major contributor to loss. In this
article, eGaN FETs and MOSFETs were compared using
similar traditional PCB layouts. The eGaN FET, combining low FOM, low package parasitics, and a small footprint
reducing PCB parasitics, outperformed MOSFETs rated
for much lower voltages. As FOM and packages improve,
the PCB layout becomes critical to high efficiency.
The next part of this series on eGaN FETs explores
the topic of PCB layout optimization to further improve
the performance achievable with these high-performance eGaN FETs.
RefeRences
[1] J. Strydom, “The eGaN® FET-Silicon Power Shoot-Out Vol. 7: Buck
Converters”, Power Electronics Technology, Vol. 38, No. 2, February, 2012.
[2]J. Strydom, “The eGaN® FET-Silicon Power Shoot-Out Vol. 8: Envelope
Tracking Power Electronics Technology, Vol. 38, No. 5, May, 2012.
[3] M. de Rooij and J. Strydom, “eGaN® FET – Silicon Shoot-out Vol. 9:
Wireless Power Converters,” Power Electronics Technology, Vol. 38, No.
7, July 2012.
[4] D. Reusch and J. Strydom, “The eGaN® FET-Silicon Power ShootOut Vol. 10: High Frequency Resonant Converters,” Power Electronics
Technology, Vol. 38, No. 9, September 2012.
[5] J. Strydom, “The eGaN® FET-Silicon Power Shoot-Out Vol. 1:
Comparing Figure of Merit (FOM),” Power Electronics Technology, Vol.
36, No 9, September 2010.
[6] J. Strydom, “The eGaN® FET-Silicon Power Shoot-Out Vol. 11:
Optimizing FET On-Resistance,” Power Electronics Technology, Vol. 38,
No. 10, October 2012.
[7] A. Elbanhawy, “Effects of parasitic inductances on switching performance,” in Proc. PCIM Eur., May 2003, pp. 251–255.
[8] G. Nobauer, D. Ahlers, J. Sevillano-Ruiz, “A method to determine
parasitic inductances in buck converter topologies,” in Proc. PCIM Eur.,
May 2004, pp. 37–41.
[9] B. Yang, J. Zhang, “Effect and utilization of common source inductance
in synchronous rectification,” in Proc. IEEE APEC’05, Mar. 2005, vol. 3,
pp. 1407–1411.
[10]M. Pavier, A. Woodworth, A. Sawle, R. Monteiro, C. Blake, and J.
Chiu, “Understanding the effect of power MOSFET package parasitic on
VRM circuit efficiency at frequencies above 1 MHz,” in Proc. PCIM Eur.,
May 2003, pp. 279–284.
[11]D. Reusch, D. Gilham, Y. Su and F.C. Lee, “Gallium nitride based
multi-megahertz high density 3D point of load module,” APEC 2012.
pp. 38-45. Feb. 2012.
[12] T. Hashimoto, T. Kawashima, T. Uno, Y. Satou, N. Matsuura, “System
in package with mounted capacitor for reduced parasitic inductance in voltage regulators,” Applied Power Electronics Conference and Exposition, 2008.
APEC 2008. Twenty-Third Annual IEEE, pp.187-191, 24-28, Feb. 2008.
[13] Y. Kawaguchi, T. Kawano, H. Takei, S. Ono, A. Nakagawa, “Multi
Chip Module with Minimum Parasitic Inductance for New Generation
Voltage Regulator,” Power Semiconductor Devices and ICs, 2005.
[14] A. Ball, M. Lim, D. Gilham, F.C Lee, “System design of a 3D integrated non-isolated Point Of Load converter,” Applied Power Electronics
Conference and Exposition, 2008. Twenty-Third Annual IEEE, pp.181-186,
24-28 Feb. 2008.
[15] D. Reusch, F.C. Lee, Y. Su, D. Gilham, “Optimization of a High
Density Gallium Nitride Based Non-Isolated Point of Load Module,” Energy
Conversion Congress and Exposition (ECCE), IEEE, Sept. 2012.
[16] D. Reusch, “High Frequency, High Power Density Integrated Point of
Load and Bus Converters,” PhD Dissertation, Virginia Tech, 2012.
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March 2013 | Power Electronics Technology
23
DESIGNfeature
LARRY MEARES, Intusoft
Designing R2CD Snubbers Using
Standard Recovery Diodes
RCD snubbers are widely used
to limit peak voltage stress
in switched-mode power
supplies. By adding an extra
resistor, the R2CD configuration improves performance
across the board, lowers
cost, and increases immunity
against variations in circuit
parameters.
S
nubbers play an important role in switch-mode power supplies.
To understand use of the snubber, we have to look at the performance of a switch-mode power supply. Fig. 1 illustrates the
two types of snubbers used in a switch-mode power supply:
RCD and R2CD. When a switch-mode power supply, SMPS,
switches OFF, the parasitic leakage inductance [1] that couples
the primary and secondary winding of a transformer must be
charged or discharged. In that brief instant, the resulting voltage transient must
be stabilized to prevent circuit destruction caused by the sudden change in current transmitted to the leakage inductance.
The test circuit shown in Fig. 2 describes the snubber circuit. FET, Q1, charges
the simulated leakage inductance when it is turned ON and releases the charge
into the snubber circuit when it is turned OFF. The parameter blocks in Table
1 represent either RCD or R2CD (used when reverse recovery time, trr, is > .5
microseconds) configurations. Snubber parameters are adjusted for each case
to yield the same results. The FET should be the same device used in the final
design in order to account circuit affects imposed by the actual device. The drain
voltage reaches about 2 kV using this test circuit when the snubber is removed
and avalanche does not occur.
If the snubber diode is conducting reverse current when it turns OFF, a
negative transient voltage spike occurs at Q1 drain (typical RCD behavior). This
spike results from discharging the residual leakage inductor current into Q1 and
results in a damped ringing behavior when switch capacitance is included in the
RR
27k
+
C1
100 nF
R
100k
C3
2.7 nF
V1
350
12 µ HY Leakage Inductance
X1
R1
68
D1
MUR160
RCD Snubber
X2
MBR20100
C2
R5
D4
1N4007
R2CD Snubber
X5
MTP6N60m
+
V2
Rsense
Fig. 1. RCD and R2CD snubbers for ON Semi Demo, NCP1216 70 Watt flyback transformer.
24
Power Electronics Technology | March 2013
www.powerelectronics.com
(1)
Where:
1< k < 2
Most of the power loss occurs in this resistor. The
turn-off time of a diode is determined by the time
required to remove its stored charge, possibly less than
trr. For an R2CD snubber, if the half period of resonance
is small compared to trr, then the diode turns off with no
switching transient. As trr increases, there is no penalty,
mitigating the dearth of specifications for these diodes.
It’s especially noteworthy that EMI generated by the
switching transient of a properly designed R2CD snubber is actually less than the RCD version. Fig. 4 shows
the simulation waveforms illustrating how the R2CD
snubber works.
Three snubber configurations have been compared
using a kilowatt level full bridge DC-DC converter. The
output bridge drives a buck type L-C filter in a manner
www.powerelectronics.com
4.00 µ
0
3.00 µ
–5.00
–10.0
Charge in coulumbs
5.00
1
iv3
2
charge
1
2.00 µ
1.00 µ
–15.0
0
2
1.25 µ
time in seconds
0
2.50 µ
Fig. 3. Current and Charge are zero at the end of the switching transient.
1 V1rcd
2 Iv3rcd
Plot 1
V1r2cd, V1rcd in volts
400
200
400
-200
-400
3 Vir2cd
4 Iv3r2cd
16.0
iv3r2cd, iv3rcd in amperes
POWER LOSS IN RESISTOR
The R2CD snubber has an extra resistor, R1, in series
with the diode. The trick in designing a resonant R2CD
snubber is selecting R1 to damp the R1-L-C resonance
so that:
Plot 1
Iv3 in amperes
model. However,
if a long trr diode
WR
C
is used, it’s pos{C}
{R}
+
sible to design
V3
the circuit so
IV3
+ V1
that the current
L
DUT
{L}
{Vs}
is zero when the
TT = {TT}
diode drops out
CJO = {CJO}
of conduction; a
WR1
R1
{R1}
high efficiency
IQ1_1
resonant condiR2
5
Vsnub
Q1
tion.
STB30NM50N
+
Standard
V2
Ton
recovery diodes
exhibit a long
storage time that
is characterized Fig. 2. Snubber test circuit examines theory
as trr in the data
sheets. By convention, the standard recovery diodes have
trr specified as greater than 0.5 µS; trr typically ranges
from 2 µS to 5 µS, increasing with higher voltage rating. When the transient current in the snubber diode is
shorter than trr, the stored charge can be removed just as
the diode voltage approaches zero. Fig. 3 shows the current waveform and its associated charge. Thus, an R2CD
snubber operates properly for any trr that’s greater than
one-half the period of snubber resonant frequency.
12.0
3
8.00
4.00
0
4
1.00µ 2.00µ 3.00µ 4.00µ 5.00µ
time in seconds
Fig. 4. The R2CD snubber (red and green traces) produces less EMI than the RCD
snubber.
that produces a 120 VRMS full wave rectified sine wave.
The input bridge is connected to a combination 48 Volt
battery and solar panel battery charger. Bi-directional
power conversion is required so that motor loads can
be supplied as well as battery charging. Normally, all
transistors in the output bridge are turned on so that the
load current flows through the transformer secondary.
When power is needed, opposite output bridge switches
are turned off and corresponding input bridge switches
connect to the battery/charger system. Thus, the initially
charged transformer leakage inductance must be discharged into a snubber. In this case, the snubber protects
the high-side driver components because the switches
are avalanche rated.
The RCD and R2CD snubbers are the same ones
evaluated in the test simulation. A resonant snubber
based on [2] was included in the evaluation. Fig. 5 shows
March 2013 | Power Electronics Technology
25
SWITCHED-MODEsupplies
the test results.
Additional simulation and tests were run
on a two Watt housekeeping power supply
and are summarized in Fig. 6. The simulation
parameters were adjusted to match the test
results. VR1 is the voltage across snubber resistor R1. The major adjustments were for leakage
inductance (adjusted in the transformer model
by changing the insulation thickness) and varying diode trr.
1
96.0
Eff
3
Eff#a
4
Eff#b
Resonant
Eff#a Eff#b Eff
94.0
4
Plot 1
DISCUSSION
92.0
3
Simulation Vs. Test: It is important to make
test and simulation results agree. Once the
RCD
R2CD
1
simulation results match the test results; then
simulation parameters can be varied in order
90.0
to evaluate circuit performance for varying
Operating points
Max Grid Tie, 1.2kW
tolerances and temperatures. Some of these
Average Backup 350W
parameters, trr for example, can’t be varied
Peak backup 1.75kW
88.0
by part substitution. The equations describing
snubber loss are approximate and don’t include
the complex behavior of switched current,
1.00k
200
600
1.40 k
1.80k
voltage and time. So like it or not, simulation is
pin
a required design tool!
R2CD Topology: Variations in the R2CD Fig. 5. R2CD snubber has best overall efficiency below 1 kW.
topology yield the same results. These variations include moving R1 to be anywhere in series with
where magnetizing inductance is in parallel with a series
C; for example on either side of the diode, the ground
combination of leakage inductance and reflected windleg or in series with C. Similarly, R can go anywhere
ing resistance. An LCR mulltimeter will report vastly
after the snubber diode. Placing R1 in the ground leg
different results at 1 kHz vs. 100 kHz. For the circuit
simplifies measuring snubber current. C can be conused in Fig. 5, the result is 168 µH vs. 5 µH. In any event,
nected to ground or the output. Connecting to the outthe short circuit driving point impedance needs to be
put increases conducted EMI and reduces the required
compared to the simulation result in order to evaluate
voltage rating.
or adjust the transformer model.
Leakage Inductance: The power to be snubbed is ,
EMI reduction: The reduced EMI is caused by the resonant charge characteristics (Fig. 2). To get the best result
R1 must be set based on L and C (Equation 1) so that
the only degree of freedom in the design is the selection
so that minimizing leakage inductance is the first priof C. As C increases, voltage clamping is reduced but
ority. Leakage inductance is difficult to predict [1]
efficiency is reduced. R1 can be cut in half (k = 1 in
because magnetic device geometry depends on winding
Equation 1) without much change in the EMI charactertechniques, Therefore, is necessary to measure leakistic in order to reduce the maximum voltage.
age inductance by shorting out the secondary windEfficiency: To calculate the RCD snubber power loss;
ings. Measuring leakage inductance is surprisingly more
First, Calculate the power loss in a Zener clamped snubcomplex than it appears. Consider an equivalent circuit
ber with no switch capacitance:
TABLE 1: COMMON DIODE DATA
PART NUMBER
RM
ENERGY
COST
1N4007 x3
5.3
3.70 µJ
$0.17
6A10DCT
5.4
3.99 µJ
$0.26
20ETS12
9.5
10.9 µJ
$1.79
26
Power Electronics Technology | March 2013
(2)
(3)
(4)
www.powerelectronics.com
vr1, CH3 in volts
Plot1
v(1), CH1 in volts
ery characteristic [4]. Forward recovery
occurs when the diode is switched on
1 V(1)
2 Vr1
3 CH1
4 CH3
because of the low initial conductivity of
the intrinsic region. For the R2CD snub120
85.0
ber, the initial turn-on resistance, Rm,
can be used as a figure of merit. Table
1
85.0
65.0
3
1 lists the characteristics and cost for
several common diodes. Rm and energy
40.0
45.0
were based on test using a 5 A current
pulse. Multiplying energy by operating
W1 & W2 Simulation
frequency gives the power dissipation
W3 & W4 Test
0
25.0
caused by forward recovery. The loss
is expected to grow proportional to the
-40.0
5.00
square of current. Detailed test data is
2
available [3].
9.966m
9.968m
9.970m
9.972m
9.974m
The 1n4007 used three devices in
time in seconds
parallel and cost data for 500 quantiFig. 6. Two Watt R2CD snubber simulation compares favorably with lab test results.
ties. All others used 100 quantities for
pricing. Digi-Key was used for the price
estimates. All trr values were in the acceptable range for
(5)
the application. Notice the newer, “modern” technology
diode had greater forward recovery loss and costs more.
Then replace the zener with a resistor having the
The 20ETS12 data was representative of 2 other part
same average current.
numbers (40EPS08 and D6020L) that were even more
expensive. For reference, the DSEP8-12A fast recovery
(6)
diode used in an RCD snubber costs $1.37.
Test Circuit: The circuit in Fig. 2 simulates snubber
loss so that testing can be accomplished at lower power.
(7)
The actual circuit (Buck or Flyback) operates at much
higher power.
Where:
CONCLUSION
VM = Maximum or snubbed voltage
Vs = Input voltage
The simulation results speak loudly! The R2CD snubnVs = Flyback voltage
ber is more efficient, lower cost and produces less EMI
F = Frequency
than a standard RCD snubber. The popular misconcepL = Leakage inductance
tion that diode storage time must equal the resonant ½
Is = Switched current
period is replaced by the requirement that storage time
PZ = Zener Power
be greater than the resonant 1/2 period. More hardware
PR = Resistor Power
tests are available along with the production and test
The R2CD snubber loss is just PL, so the RCD snubdrawings [3]. The R2CD snubber is surprisingly immune
ber is flawed by not having available the proper voltage,
to circuit parameter variations while limiting peak voltnVs, to connect the bleed resistor.
ages for wide variations in R1, R, C and trr.
Some of the energy of the switched leakage inductance is dissipated in the switching transistor. The R2CD
REFERENCES:
[1] Magnetics Designer Manual. Published by Intusoft, available in PDF
configuration dissipates less of this energy in the switchformat from
ing transistor. Actual values depend on the transistor, its
http://www.intusoft.com/lit/Magdes.pdf pg 157
drive circuit and the direction of power flow.
[2} L.G. Meares, “Improved Non-Dissipative Snubbers for Buck Regulators
For the R2CD snubber, R is used to control the final
and Current-Fed
Inverters”, Proceedings of Powercon 9, July 1982, pp B-2,1-8.
value of charge. Simulation can be used to find a value
[3] http://www.intusoft.com/snubbetTest.htm
for R that minimizes EMI.
[4] Cliff L. Ma and P.O. Lauritzen “A Simple Power Diode Model with
Diode Selection: The R2CD snubber diode relies on
Forward and Reverse
unspecified data sheet parameters. Two important
Recovery”. IEEE Transactions on Power Electronics, vol 8, No.4, October
parameters are storage time and the forward recov1993.
www.powerelectronics.com
March 2013 | Power Electronics Technology
27
designfeature
Christian KueCK, Linear Technology Corp.
Layout Power Supply Boards to
Minimize EMI - Part 3: EMI Basics
PC-board layout sets the
functional, electromagnetic
interference (EMI), and thermal behavior of a power
supply. Good layout from
first prototyping on actually
saves significant resources
in EMI filters, mechanical
shielding, EMI test time, and
PC board runs. This segment
reviews some of the basic
fundamentals of EMI and EMC.
E
lectromagnetic far field impedance is about 377Ω = 120π or
29,9792458 × 4 × πΩ for the vacuum velocity of light. Any
electromagnetic wave far enough from its source (rule of thumb
>wavelength/2 × π) has a 377Ω relationship between its magnetic and electric field. Closer to the source, it can be a perfectly
matched antenna, which transforms its input power source to the
right 377Ω electromagnetic field. Or, there is significant mismatch
and the antenna starts mainly as a magnetic field source or an electric field
source.
The magnetic field source has a lower impedance of 377Ω. The electric field
source has a higher than 377Ω impedance. The graph in Fig. 23 shows that,
regardless if it starts as an electric or a magnetic field source, the electromagnetic
field balances itself to its far field impedance at a distance of:
(1)
Zw
where λ = Wavelength
Nonisolated switch mode power supply units have primarily magnetic field
sources, since the impedances of the EMI-relevant loops with high di/dt are
much lower than 377Ω unless you have very low current high voltage power
supplies. So minimizing the AC magnetic fields on any nonisolated power supply
unit will be the key to success.
Any isolated power supply unit will have AC loops
with lower than 377
10000
Ω, where the same
5000
Electric
magnetic field mini3000
Field Source
mization as on non2000
DIPOLE
isolated PSUs will be
1000
required. However,
due to the very nature
500
of isolation, we need
300
200
higher impedances
between the isolation
100
barrier. On the isola50
Magnetic
tion barrier, which is
Field Source
30
mostly done with a
20
1
transformer, we try to
Near Field
Far Field
2π
10
get MΩ of isolation.
0.02 0.05 0.1
0.2
0.01
0.5
1
2
5
10
0
On the isolation barr /λ
Fig. 24. The AC current flows around an area rier, the electric AC
Fig. 23. Regardless if it starts as an electric or a magnetic field source, the
and creates the magnetic field part of a normal field dominates and
electromagnetic field balances itself to its far field impedance.
dipole antenna.
requires a different
28 Power Electronics Technology | March 2013
www.powerelectronics.com
strategy. Here we try to get as low
capacitive field coupling as possible.
So we try to get as much distance as
possible and to minimize the size of
any conductive material.
DIPOLE ANTENNA EFFECT OF THE HOT LOOP
When analyzing what the hot loop
does, magnetic dipole antennas give
a good clue.
The AC current flows around an
area and creates the magnetic field
part of a normal dipole antenna, as
shown in Fig. 24).
Magnetic antennas with loop
diameters <<λ have very low radiation resistance. The range: µ to
m .
(2)
RR = Radiation resistance in ohms
F = Area of magnetic loop
N = Number of turns (= 1 in most
layouts)
λ = Wavelength
with
IW
and N = 1 for all
practical layout loops
H
(3)
I
c = Speed of light ≈300000km/s
f = Frequency
The radiation resistance is low
(mΩ) for typical dimensions of a
PC-board power supply unit.
Increasing the radiation resistance
improves the matching and increases emitted radiation proportional to
Fig. 25. If the current is DC, it will look like this.
Because I is constant, the resulting H is constant
and Iw is zero. In a case that I has an AC content,
which means there is di/dt, then the resulting magnetic field H changes.
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March 2013 | Power Electronics Technology
29
pc boardlayout
the radiation resistance. The parameter we can influence the most with
layout is the area of the magnetic
loop. The emitted radiation is proportional to the square of this area.
Skin EffEct
To understand the effect of shielding, we need to dig a bit into
the electromagnetic properties of
the materials used. Electric cur-
fig. 26. forces that move the current density to
the outside of a conductor are called skin effect.
rent, whether or not it is tied to a
conductor, will always flow on the
path of least impedance. For higher
frequencies, this will be the path
of least inductance. This means the
current will also flow on the path
of its lowest losses. Electric conduction material minimizes any internal magnetic AC fields by creating
eddy currents that oppose AC fields
internal to the conducting material.
Viewed from the outside, this looks
like the skin effect, because the current density is forced to the outside
of the conductor.
For another example, assume
that the current, I, flows uniformly
through a cylinder. This is a typical
case for DC current.
If the current is DC, then it will
look like the shape in Fig. 25. I is a
δ
fig. 27. Wall thickness of a pipe, which would give
the same resistance at Dc as a full cylinder wire at
Ac, is called skin depth.
30 Power Electronics Technology | March 2013
www.powerelectronics.com
constant, so the resulting H is constant and IW is zero. In
the case that I has an AC content, which means there is
di/dt, then the resulting magnetic field H changes. The
changing magnetic field H creates induction voltage and
since we are inside a conductor, the induction voltage
creates an induced current IW, often referred to as eddy
current. Eddy currents will create losses. If we assume
that nature minimizes these losses, the only way to
minimize this is to change the original uniform current
distribution and let all current flow only at the surface
of the cylinder. Now the magnetic field H is only at the
surface of the cylinder. This way, the eddy currents IW
are moved to the outside and the return path is cut off,
omitting most of the eddy currents and its losses.
SKIN EFFECT AND SKIN DEPTH
We can think of forces (Fig. 26) that move the current
density to the outside of a conductor. This is called skin
effect. The wall thickness of a pipe, which would give
the same resistance at DC as a full cylinder wire at AC,
is called skin depth (Fig. 27). Since the forces moving the
current density to the outside are a result of the induc-
www.powerelectronics.com
H
H additional
Fig. 28. Large diameter pipe wall conducting all the current on its surface.
Total inductance of a given length of this wire is the complete volume integral
of its magnetic field up to infinity or whatever physical size you assume for the
universe.
March 2013 | Power Electronics Technology
31
PC BOARDlayout
100 µm
tion law, the skin depth
goes down with rising
frequency. Skin depth
goes down with rising
conductivity and goes
10 µm
down with increasing
magnetic permeability.
For another example, assume we have a
pipe wall conducting
all the current on its
1 µm
surface, as shown in
the large diameter in
Fig. 28. Total inductance of a given length
of this wire (1 meter
or 1 foot whatever is 100 nm
most suitable) is the
100k
10k
1M
complete volume integral of its magnetic
field up to infinity or
whatever physical size you assume for the universe.
Now we shrink the pipe diameter to the smaller inner
one shown in Fig. 28. The magnetic field is still the same
as the larger pipe to infinity. However, we now have an
additional magnetic field volume between the new small
pipe and the prior larger pipe diameter. So the total
integral of the magnetic field is now larger. This shows
that the inductance is larger for a thinner conductor of
a given length. Or, we can assume that the inductance
Opposite direction current
Outer magnetic
field cancels
Same direction current
Outer magnetic
field intensifies
Fig. 30. Opposite direction current forces will attract the currents to each other,
which is called proximity effect.
32
Power Electronics Technology | March 2013
Highly Doped
Silicon 10**20
Copper Clad 35 µm 2 oz
25 µm Bond Wire
17 µm 1 oz
8.5 µm 0.5 oz
TIN
Fig. 29. Graph of skin
depth over frequency
for some materials.
Au
Cu
Fe-Ni
u-Metal
Superconductor
(Typ 1.2 10 nm to 500 nm)
10M
100M
1G
10G
Frequency (Hz)
increases as more of the current moves from the outer
wall to the center of the wire. If we apply a voltage over
both ends of the wire, the resulting current distribution
is determined by the impedance. Since the inductance in
the center is higher, most current density moves to the
surface. The most extreme case of skin effect in conductors with zero resistance is shown in superconductors.
There, quantum effects prevent all current from being
bound to an outer layer of zero thickness. The thickness
where most current is concentrated on superconductors
is called London depth.
A graph of skin depth over frequency for some materials is shown in Fig. 29. We see that copper on typical
PC-board material is affected by skin effect in the range
of 5 MHz to 50 MHz. And, we see that even highly
doped silicon at the thickness typically used on ICs
is only affected in the terahertz region. Copper (Cu)
and gold (Au) are close together. Materials with high
magnetic permeability, such as Fe-Ni, have low skin
depth, even at audio frequencies. For this reason, material with high magnetic permeability is used to shield
audio transformers.
AC current through a good conductor will push current density to the outside. The current will flow where
the impedance, dominated by inductance, is lowest.
With regard to the impact of skin effect on layout and
components, we can derive simple guidelines. Better is
short and thick or wide. Reverse geometry capacitors
have lower ESL because they are shorter and thicker. If
we have opposite direction current, the same forces will
attract the currents to each other (Fig. 30). This is called
proximity effect.
www.powerelectronics.com
designfeature
Sam DaviS, Editor-in-Chief, PET
IC Pair ImprovesTransmitting And
Receiving Of Wireless Power
W
ireless Power Transfer relies on magnetic induction between
a planar receiver and transmitter coils. When the receiver
coil is positioned over the transmitter coil, magnetic coupling occurs when driving the transmitter coil. The resultant flux is coupled into the secondary coil, which induces
a voltage and current flows. The secondary voltage is rectified, and power is wirelessly transferred to a load. Two new
Texas Instruments ICs manage this transfer: one transmits and the other receives
the transferred power, as shown in Fig 1.
The power transfer receiver IC is a bq51050B secondary-side direct liion battery charge-controller (Fig. 2). This 20V receiver IC provides:
• Efficient ac/dc power conversion
• A digital controller that complies with the WPC 1.1 Standard
• The necessary control algorithms needed for li-ion and li-pol battery charging
The bq51050B’s self-contained charger eliminates the need for the separate battery charger circuit used in older generation systems. This inductor-free, single-stage
design delivers high efficiency and saves board space, compared with implementations requiring the separate charger IC.
The bq51050B integrates a low-impedance synchronous rectifier, low-dropout
regulator (LDO), digital control, li-ion charger controller, and accurate voltage
and current loops. The entire power stage (rectifier and LDO) utilize low-resistive
NMOS FET’s (100-mΩ typical RDS(ON)) ensuring high efficiency and low power
dissipation. Its features include:
• Wireless power receiver, rectifier and battery charger in one small package
• 4.2V or 4.35V battery output voltage options
• Support for up to 1.5A charging current
• 93% peak ac-dc charging efficiency
• 20V maximum input voltage tolerance, with input overvoltage protection clamp
• Thermal shutdown and overcurrent protection
• Temperature monitoring and fault
detection
Receiver
Transmitter
• Power stage output that tracks rectiPower
fier and battery voltage to ensure
maximum efficiency across the full
Power
Voltage
AC-DC
Load
Rectification
charge cycle
Stage
Conditioning
• Either small WCSP or QFN packages
Communication
The bq500410A wireless power transmitter features:
BQ500410A
Controller
Feedback
• Expanded “free-positioning” using a
Bq51K
three-coil, A6, transmit array
• Intelligent control of wireless power
Fig. 1. Wireless power transfer using receiver and transmitter circuits.
transfer
Recently-introduced wireless power receiver and
transmitter ICs are poised
to improve wireless power
transfer employed for charging li-ion batteries. These new
ICs comply with the Wireless
Power Consortium (WPC) 1.1
standard.
www.powerelectronics.com
March 2013 | Power Electronics Technology
33
WIRELESSpower
ICs
• Wireless Power Consortium (WPC) compliance
• Digital demodulation the reduces components
• Overcurrent protection
• A signal output that indicates the start of power transfer,
which can activate a ceramic buzzer
• An End-of-Power Transfer signal that causes an LED
indicator to illuminate
• LED indication of charging state and fault status
• Overcurrent monitoring threshold that can halt power
transfer for one minute.
• Power-On Reset (POR) that monitors the supply voltage
and sets the device startup sequence.
• A 48-pin, 7 mm x 7 mm QFN package
• Operating temperature range of –40 °C to 110 °C
TRANSFERRING POWER
Power transfer depends on coil coupling that depends on:
• Distance between coils
• Alignment
• Coil dimensions
• Coil materials
• Number of turns
• Magnetic shielding
• Impedance matching
• Frequency Duty cycle
Receiver and transmitter coils must be aligned for best
coupling and efficient power transfer. The closer the space
between the two coils, the better the coupling. However,
to account for housing and interface surfaces the practical
distance is set to be less than 5 mm, as defined within the
WPC Standard (see sidebar, “Wireless Power Consortium
1.1 Standard”). Shielding is added as a backing to both the
transmitter and receiver coils to direct the magnetic field
to the coupled zone.
You can control regulation by varying any one of the
coil coupling parameters. However, for WPC compatibility, the transmitter-side coils and capacitance are specified
and its resonant frequency point is fixed. Power transfer is
regulated by changing the frequency along the resonance
curve from 112 kHz to 205 kHz.
COILS
The bq500410A uses the A6 coil arrangement to achieve
greater than 70-percent efficiency. The WPC Standard
establishes coil and matching capacitor specification for
the A6 transmitter. Although the bq500410A is intended
to drive an A6 three-coil array, it can also be used to drive
a single coil. For single coil operation, the two outer coils
and associated electronics are simply omitted. Fig. 3 shows
the A6 three-coil configuration that allows 70 x 20 mm
charge surface area as well as the A1 single-coil 18 mm x
18mm “bull’s-eye” charge space. The 70 mm by 20 mm
charge area is 400-percent larger than 18-mm by 18-mm
area now being used. Use of the A6 coil configuration provides a “free-positioning” digital wireless power controller.
The performance of an A6 transmitter can vary based
■ WIRELESS POWER CONSORTIUM 1.1 STANDARD (JULY 2012)
THE WIRELESS POWER CONSORTIUM (WPC)
is an international group of companies
from diverse industries. They developed
the WPC Standard to facilitate cross
compatibility of compliant transmitters
and receivers. The Standard defines the
physical parameters and the communication protocol used in wireless power
transfer.
Power transfer involves two device
types. Those that provide the wireless
power are designated as Base Stations,
and those that consume wireless power
are called Mobile Devices. Power transfer always occurs from a Base Station
to a Mobile Device. The Base Station
contains a power transmitter with a primary coil. The Mobile Device contains a
power receiver with a secondary coil. The
primary and secondary coils form two
halves of a coreless resonant transformer.
34
Shielding at the bottom face of the primary coil and the top face of the secondary coil, as well as the close spacing of
the two coils, ensures that power transfer
occurs with an acceptable efficiency.
Typically, a Base Station has a flat
surface—referred to as the interface surface — on top of which a user can place
one or more Mobile Devices. This ensures
that the vertical spacing between primary
and secondary coils is sufficiently small.
FREE POSITIONING
“Free-positioning” using the A6 coil
arrangement does not require active participation in alignment of the primary and
secondary coils. One implementation of
free-positioning uses an array of primary
coils to generate a magnetic field at the
location of the secondary coil. Another
implementation uses mechanical means
Power Electronics Technology | March 2013
to move a single Primary Coil underneath
the Secondary Coil.
Main features in wireless power transfer
are:
• Contactless power transfer from a Base
Station to a Mobile Device, based on
near field magnetic induction between
coils.
• Transfer of about 5 W, using an appropriate secondary coil (having a typical
outer dimension of around 40 mm).
• Operation at frequencies in the
100…205 kHz range.
• Free-positioning enables arbitrary
placement of the Mobile Device on the
surface of a Base Station.
• A simple communications protocol
enables the Mobile Device to take full
control of the power transfer.
• Very low stand-by power (implementation dependent).
www.powerelectronics.com
Fig. 2. the bq5105x secondary-side 20V receiver
is a digital controller that provides ac/dc power
conversion, a WPC 1.1 communication protocol,
and an integral li-ion battery charger.
CCLAMP1
CBOOT 1
C1
TI
Wireless TX
Power
Coil
Transmitter
RX
Coil
BOOT 1
C2
AC1
AC2
CBOOT2
CCOMM2
CCLAMP2
CCOMM 1
BATT
COMM 1
D1
RECT
C3
bq5105xB
TS
BOOT2
COMM2
CLAMP2
CLAMP1
ILIM FOD
R1
C4
ROS
/CHG
TERM
EN2
PGND
R5
RFOD
on the design of the A6 coil set. For best performance with
small receiver coils under heavy loading, design the coil set
so that the distance between the centers of the outer coils
is on the low end of the specified tolerance (49.2 mm).
The WPC standard describes the dimensions, materials of the coils and information regarding the tuning of
the coils to resonance. The value of the inductance and
its associated resonant capacitor are critical for proper
operation and system efficiency. The resonant tank circuit requires a total capacitance value of 68 nF plus a 5.6
nF center coil, which is the WPC system compatibility
requirement. Capacitors must be rated for at least 100 V
and comprise a high quality C0G dielectric
The bq500410A drives three independent half-bridges.
Each half-bridge drives one coil of the A6 coil set. A
TPS28225 is the recommended driver IC that features
high-side drive capability and enables use of N-channel
MOSFETs throughout. You can derive the gate-drive supply from a primitive, active voltage divider.
The bq500410A supports both Parasitic Metal
Fig. 3. the bq500410a can employ either of two types of coil configurations. on
the left is the a1 single coil and on the right is the a6 three-coil configuration
used for free-positioning.
www.powerelectronics.com
Detection (PMOD) and Foreign
Object Detection (FOD) by conR4
tinuously monitoring the efficiency
PACK+
NTC
of the established power transfer.
+
PMOD and FOD protect against
–
power lost due to metal objects in
PACK–
the wireless power transfer path.
The bq500410A compares input
power, known losses, and the
Tri-State
amount of power reported by the
Bi-State
receiver IC. This yields an estimate
HOST
of unaccounted power presumed
lost due to misplaced metal objects.
Exceeding this loss generates a fault
and halts power transfer.
The FOD algorithm uses information from an in-system characterized and WPC1.1
certified receiver and it is more accurate than the previous
PMOD obtained from WPC1.0. While WPC1.0 required
merely the rectified power packet, WPC1.1 also uses the
received power packet that more accurately tracks power
used by the receiver.
WPC 1.1 is intended for 12 V systems, but the
bq500410A requires a 3.3 VDC input supply. Therefore,
use a buck regulator or a linear regulator to step down
from the 12V system input.
CoMMuniCation
Communication within the WPC is from the receiver
to the transmitter, where the receiver tells the transmitter to send power and how much. For regulation, the
receiver must communicate with the transmitter whether
to increase or decrease frequency. The receiver monitors
the rectifier output and using Amplitude Modulation
(AM) to send packets of information to the transmitter. A
packet consists of a preamble, header, actual message and
a checksum, as defined by the WPC standard.
The bq500410A starts power transfer by pinging the
surrounding environment looking for WPC compliant
devices waiting to be powered. If it finds a compliant
device it safely engages it, reads the packet feedback from
the powered device, and manages the power transfer.
The receiver sends a packet by modulating an impedance network. This AM signal reflects back as a change in
the voltage amplitude on the transmitter coil. The signal is
demodulated and decoded by the transmitter-side electronics and it adjusts the frequency of its coil-drive output
to close the regulation loop.
March 2013 | Power Electronics Technology
35
PET innovations
SAM DAVIS, Editor-in-Chief
Engineering LED Drivers for Incandescent
Bulb Replacement is a BOM Challenge
Fig. 1. Cutaway view of a typical LED
replacement bulb.
L
ED-BASED BULBS that replace their incandescent
counterparts require a significant amount of engineering (Fig. 1). In the 60 W range, the LED driver
board can take up more than 15% of the overall cost
of the bulb. Consumer price pressure on the LED driver
IC manufacturer has forced its designers to work with one
hand on a calculator and the other hand on an oscilloscope.
For example, designers have a choice between using a
power MOSFET or bipolar junction transistor (BJT) to
power the LEDs. The MOSFET may cost $.12 where the
BJT is only $.07. Therefore, the BJT wins the fight for a
lower bill of materials (BOM).
Price pressure also involves the use of power factor
correction (PFC). Worldwide regulations mandate PFC
and minimum power line harmonics. These regulations
vary by country and by application. The traditional PFC
circuit requires a two-stage circuit, which increases the
BOM. Therefore, some manufacturers have opted for a
less costly single-stage design. This simpler single-stage
technique lowers the cost, but it can impact the ripple cur-
Fig. 2. iW3626 non-dimmable LED driver provides capability for incandescent
lamp replacement.
36
Power Electronics Technology | March 2013
rent through the LED that causes
objectionable flicker. A single-stage
design has an inherent trade-off between the
PF and output ripple. It is possible for single-stage PFCs
to provide an adequate power factor with relatively low
output ripple, minimizing both heat and flicker, with little
impact on overall cost or size.
LED bulb operating temperature is another design-cost
consideration, particularly for residential use where the
bulb may be in a location with little or no cooling. This
requires some form of over-temperature protection (OTP)
to protect internal bulb circuits.
Isolated and non-isolated LED drivers may be employed.
Isolated types require a transformer, whereas the non-isolated version eliminates that cost. However, the non-isolated driver incurs other costs because it requires heavier
insulation and a more costly mechanical enclosure. This is
another design-cost trade-off.
DRIVER DESIGN
The iW3626 from iWatt takes into account these
design-cost tradeoffs, a non-dimmable LED driver for
incandescent lamp replacement (Fig. 2). It is a high performance, power factor corrected, AC/DC power controller
for LED luminaires (Fig. 3). The IC uses digital control
technology in a PWM flyback power supply to achieve relatively high power factor while minimizing LED current
ripple. It operates in quasi-resonant mode to provide high
efficiency along with a number of key built-in protection
features that minimize the external component count.
This simplifies EMI design and lowers the BOM.
As shown in Fig. 3, its features include:
• All-in-one, non-dimmable, low-cost off-line LED driver
• Supports universal input voltage range (90V – 277V)
up to 10W
• Supports isolated or non-isolated LED driver
www.powerelectronics.com
Output Current In Percentage of Nominal
• Isolated design without opto-coupler
L
• Supports wide range of LEDs with tight
current regulation
VOUT • Helps reduce light flicker
• Active start-up scheme enables fastest
+
possible start-up
N
VOUT +
• No audible noise over entire the LED’s
operating range
The iW3626 removes the need for a
secondary feedback circuit while achievU1
ing excellent LED current regulation over
iW3626
line and load variation. It also eliminates
VCC 1
6 OUTPUT
the need for loop compensation compo+
4 CS/PF
nents while maintaining stability under all
ASU 3
operating conditions.
FB/OTP 2
5 GND
iWatt’s proprietary technology maximizes the iW3626 performance in a small
SOT-23 package. The iW3626 offers
two multi-function pins, allowing users
to configure PFC and OTP threshold as Fig. 3. Application circuit for iW3626 employs transformer that enables non-isolated design.
required with no cost and size impact,
thereby providing design flexibility.
Configurable PFC is achieved using iWatt’s patent(%)
pending hybrid control method that adaptively switches
100
between constant current and TON control modes within
90
an AC cycle. TON is the duration of the switch in the ON
position (the width of a PWM cycle). This hybrid control
80
method shapes the TON width, effectively setting the
70
amount of time allocated for the constant current control
60
versus the TON control. The IC’s PFC circuit provides:
• Tight current regulation over line/load ranges (± 3%)
• Power factor adjustable from >0.7 to >0.9
• Low current ripple
• Cycle-by-cycle regulation
0
• Built-in compensation for AC line voltage variation
0 90
100 110 120 130 140 150
°C
Fig. 4 shows how the iW3626 reacts to its junction
Junction Temperature TJ
temperature, which is an indication of the temperature
inside the sealed LED bulb. When junction temperature
Fig. 4. The iW3626 over-temperature protection derating characteristics relationship between output current vs. junction temperature.
reaches a point set by the system designer, the iW3626
LED driver automatically reduces the current drive to
the LED. This lowers the power dissipation and results in
built-in protection features include LED open/short,
cooler overall operation. This reduces the risk of thermal
input over-voltage, over-current, and current-sense resisrunaway and ensures the temperature rating of the elector short protections.
trolytic capacitors in the system is not exceeded, thereby
The table below lists the characteristics of three of
helping ensure predictable bulb operating life. Additional
iWatt’s drivers for LED bulb replacement.
TABLE: NON-DIMMABLE AND DIMMABLE LED DRIVERS FOR LED BULB REPLACEMENT
PART NO.
POWER (W)
POWER
FACTOR
TOPOLOGY
SWITCHING
FREQUENCY
LED
DRIVER
PACKAGE
DIMMING
(%)
iW3616
3 - 12
>0.95
2-Stage
200 kHz
FET
SO-14
1 - 100
iW3617
3 - 25
>0.95
2-Stage
200 kHz
FET
SO-14
1 - 100
iW3626
3 - 10
Config >0.7 − >0.9
1-Stage
72 kHz
BJT
SOT-23
No
www.powerelectronics.com
March 2013 | Power Electronics Technology
37
NEWproducts
■ Automotive
AEC-Q200-Qualified MPMA Precision
Matched-Pair Resistor Networks
Wide Band Current Monitor
VISHAY INTERTECHNOLOGY, INC. released the new MPMA
series of precision matched-pair resistors. MPMA resistor networks
are AEC-Q200-qualified and packaged in a compact molded
surface-mount SOT-23.
Each MPMA network is
constructed using moisture-resistant thin film
tantalum nitride resistor
film with enhanced passivation on a high purity
alumina substrate. The
MPMA device is resistant to moisture at + 85 °C, 85 % relative
humidity, and 10% rated power per MIL-STD-202, method 202.
Offering higher precision matching capability than discrete SMT
chips, the AEC-Q200-qualified dividers provide low TCR tracking
of ± 2 ppm/°C and tight ratio tolerance to ± 0.05 %, with excellent
long-term ratio stability over time and temperature.
The MPMA series provides a resistance range from 250 ohm to
50 kohm with divider ratios from 1:1 to 50:1. Offering a rugged
38
With a Pearson™ Current Monitor you can make precise
amplitude and waveshape measurement of AC and pulse
currents from milliamperes to kiloamperes. Current can be
measured in any conductor or beam of charged particles,
including those at very high voltages.
A typical model gives you an amplitude accuracy of +1%,
-0%, 20 nanosecond rise time, droop of 0.8% per
millisecond, and a 3dB bandwidth of 1 Hz to 20 MHz.
Other models feature 1.5 nanosecond rise time, or a droop
as low as 0.05% per millisecond.
Contact Pearson Electronics
for application information.
Pearson Electronics
4009 Transport St. Palo Alto, CA 94303 USA
Telephone: (650) 494-6444 FAX (650) 494-6716
www.pearsonelectronics.com
Power Electronics Technology | March 2013
molded case construction, the MPMA networks offer power ratings
of 100 mW at + 70 °C per resistor, extremely low noise of < - 30
dB, low voltage coefficients of < 0.1 ppm/V, and a operating temperature range of - 55 °C to + 155 °C. Available with lead (Pb)-free
terminations, the resistor networks are RoHS-compliant. Price,
depending on quantity and tolerance, ranges from $0.50 to $1.
Samples are available now, with lead times of eight to 10 weeks
for production quantities.
Vishay Intertechnology
Malvern, PA
http://www.vishay.com
■ Triple-Output
45 Watt Power Supplies with Medical
and IT Equipment Safety Approvals
EMERSON NETWORK Power announced two new triple-output open-frame ac-dc power supplies, the NPT43-M and the
NPT44-M. Extending Emerson’s successful NPT40-M series, these
new models are suitable for a wide range of IT equipment, medical,
light industrial, instrumentation and process systems, as well as
low-power dental and laboratory equipment.
NPT40-M series power supplies are rated for 45 watts power
output with convection cooling and up to 55 watts with forced air
cooling. The NPT43-M offers 5, 15 and -15Vdc regulated outputs
while the NPT44-M offers 5, 12 and 24 Vdc regulated outputs.
The expanded NPT40-M series carries a comprehensive set of
worldwide IT equipment (ITE) and non-patient contact and nonpatient critical medical safety approvals. These power supplies
can accommodate operating temperatures from minus 20 to 50
degrees Celsius at full power and as high as 80 degrees Celsius
with de-rating. They feature an industry standard 2 x 4 inch (51
x 102 mm) footprint, and a height of less than 1 inch (25 mm).
This series has a wide-range universal input capable of accommodating any ac voltage in the range 90 to 264 Vac. It can also
operate from any dc input in the range 127 to 300 Vdc, enabling
it to be used virtually anywhere in the world. The NPT42-M power
supply requires less than 74 W of input power, and inrush current
is less than 50 A peak at 230 Vac input.
These power supplies fully comply with the international EN
61000-3-2 standard for harmonic emissions. They feature built-in
EMI filters (CISPR 22 Class B) and meet rigorous international
EMC standards, including FCC Class B, EN 55022 class B and
VDE 0878PT3 Class B for conducted noise. Safety approvals
include TUV/UL/CSA 60950 and 60601-1, CB certificate, CE
mark (LVD) and CQC mark.
The NPT40-M series power supplies are fully protected against
short-circuit conditions. Their main output is also protected against
overvoltage conditions, and primary-side total power monitoring
protects the overall power supply against overload. An optional
LPX50 enclosure kit is available for added component protection.
Emerson Network Power
St. Louis, MO
http://www.emersonnetworkpower.com
www.powerelectronics.com
POWERelectronics
SAM DAVIS, Editor in Chief
POWER CONDITIONING UNIT
Issued: February 5, 2013
United States Patent 8,369,113
A power conditioning unit for delivering power from a power source to a
mains utility supply, the power conditioning unit comprising a plurality of
input terminals for connecting to the power source, a plurality of output
terminals for connecting to the mains utility supply, a voltage increasing
converter connected to the input terminals, a voltage reducing converter
connected to the voltage increasing converter and a dc-to-ac converter
connected to the voltage reducing converter and to the output terminals.
Inventors: Rodriguez; Cuauhtemoc (Impington, GB)
Assignee: Enecsys Limited (Cambridge, GB)
Appl. No.: 11/718,879
Filed: November 4, 2005
PCT Filed: November 04, 2005 PCT No.: PCT/GB2005/050197 371(c)
(1),(2),(4) Date: May 21, 2009 PCT Pub. No.: WO2006/048688 PCT Pub.
Date: May 11, 2006
ISOLATED DC-TO-DC POWER CONVERTER TOPOLOGY
Issued: February 5, 2013
United States Patent 8,369,116
New utility of an existing class of DC galvanically isolated current sourcing
circuit topologies for power conversion simultaneously allows improvement in its secondary circuit(s) to power conversion efficiency and
reduction in working voltage magnitudes, or simply reduction in working
voltage magnitudes, with resulting benefits for reduction in manufacturing cost, reduction in size and weight, and increase in market acceptance,
or may simply allow secondary circuit(s) to enable easier provisioning
of safety, improvement in reliability, or improvement in efficiency. The
magnitude of DC output voltage is optimized at higher value for greater
efficiency, while simultaneously optimizing the secondary circuit’s working voltage maximum magnitude at a lower value for greater safety. The
method requires full cycle current-compliant input impedance of the
secondary power source, whereby the secondary of the DC galvanically
isolating device behaves in a mode of being a full cycle voltage-compliant
current source.
Inventors: Maroon; Raymond Peter (Encinitas, CA)
Appl. No.: 12/715,108
Filed: March 1, 2010
POWER AMPLIFIER
Issued: February 12, 2013
United States Patent 8,373,508
A pre-driver for an amplifier comprising a load network in which the
following elements are connected in the following order: a resistor-an
inductor-a capacitor. Also described are a power amplifier comprising
such a pre-driver, a method of fabricating a pre-driver for an amplifier, and
a method of performing power amplification.
Inventors: Acar; Mustafa (Eindhoven, NL), van der Heijden; Mark Pieter
(Den Bosch, NL), Apostolidou; Melina (Enschede, NL), Vromans; Jan Sophia
(Maastricht, NL)
Assignee: NXP B.V. (Eindhoven, NL)
Appl. No.: 13/141,719
Filed: November 30, 2009 PCT Filed: November 30, 2009 PCT No.: PCT/
IB2009/055406 371(c)(1),(2),(4) Date: June 23, 2011 PCT Pub. No.: WO2010/073155 PCT Pub. Date: July 01, 2010
METHODS AND SYSTEMS FOR DIRECT CURRENT
POWER TRANSMISSION
Issued: February 12, 2013
United States Patent 8,373,307
A direct current (DC) power transmission system is described. The DC
power transmission system includes a first plurality of series connected
www.powerelectronics.com
power collection systems and at least one superconducting DC conductor
coupled to the plurality of series connected power collection systems and
configured to transmit power generated by the plurality of power collection systems to a remote load.
Inventors: Sihler; Christof Martin (Hallbergmoos, DE), Roesner; Robert
(Unterfoehring, DE), Haran; Kiruba Sivasubramaniam (Clifton Park, NY),
Bose; Sumit (Niskayuna, NY)
Assignee: General Electric Company (Niskayuna, NY)
Appl. No.: 13/116,652
Filed: May 26, 2011
METHODS AND DEVICES FOR ESTIMATION OF INDUCTION
MOTOR INDUCTANCE PARAMETERS
Issued: February 12, 2013
United States Patent 8,373,379
Methods and devices are presented herein for estimating induction motor
inductance parameters based on instantaneous reactive power. The induction motor inductance parameters can be estimated from motor nameplate
data and instantaneous reactive power without involving speed sensors or
electronic injection circuits. In one embodiment, the method includes:
measuring voltages and currents; converting the measured voltages and
currents into discrete-time voltage and current samples by analog-todigital converters; synthesizing a complex voltage from the discrete-time
voltage samples; synthesizing a complex current from the discrete-time
current samples; acquiring and storing motor nameplate data; detecting
instantaneous rotor speed by calculating an instantaneous rotor slot harmonic frequency with respect to an instantaneous fundamental frequency;
calculating.
Inventors: Gao; Zhi (Wake Forest, NC), Turner; Larry A. (Chapel Hill, NC),
Colby; Roy S. (Raleigh, NC)
Assignee: Schneider Electric USA, Inc. (Palatine, IL)
Appl. No.: 12/909,589
Filed: October 21, 2010
THERMOELECTRIC ELEMENT
Issued: February 12, 2013
United States Patent 8,373,057
A thermoelectric element includes at least one thermopair and a pnjunction. The thermopair has a first material with a positive Seebeck coefficient and a second material with a negative Seebeck coefficient. The first
material is selectively contacted by way of a conductor with the p-side of
the pn-junction, and the second material is selectively contacted by way
of a conductor with the n-side of the pn-junction.
Inventors: Span; Gerhard (Wattens, AT)
Appl. No.: 12/216,031
Filed: June 27, 2008
ENERGY STORAGE SYSTEM FOR ELECTRIC
OR HYBRID VEHICLE
Issued: February 5, 2013
United States Patent RE43,956
A battery load leveling system for an electrically powered system in which
a battery is subject to intermittent high current loading, the system including a first battery, a second battery, and a load coupled to the batteries.
The system includes a passive storage device, a unidirectional conducting
apparatus coupled in series electrical circuit with the passive storage device
and poled to conduct current from the passive storage device to the load,
the series electrical circuit coupled in parallel with the battery, such that
the passive storage device provides current to the load when the battery
terminal voltage is less than voltage on the passive storage device, and a
battery switching circuit that connects the first and second batteries in
either a lower voltage parallel or a higher voltage series arrangement.
Inventors: King; Robert Dean (Schenectady, NY), Richter; Timothy Gerard
(Wynantskill, NY), Salasoo; Lembit (Schenectady, NY)
Assignee: General Electric Company (Schenectady, NY)
Appl. No.: 13/025,102
Filed: February 10, 2011
March 2013 | Power Electronics Technology
39
Defense Electronics serves electronic design engineers working in defense and
aerospace markets with the latest technology-based news, design, and product
information. It reviews the latest advances in electronics technologies related to military
and aerospace electronic systems, from the device and component levels through
the system level, also covering the latest developments in the software needed to
simulate those defense/aerospace systems and the test equipment needed to analyze
and maintain them. It is the industry’s most trusted source of technical information for
electronic engineers involved in military/aerospace circuit and system design.
WE SUPPORT
THE DESIGN PROCESS
FROM INTENT TO
ACTION
m
ectronicsmag.co
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2012
MBER 2010
R/NOVE/APRIL
OCTOBEMARCH
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Electronics Techno
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16 | COTS GEAR FOR MILITAR
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A Special Section to
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ADVERTISERindex
Advanced Power Electronics Corporation ............................. 23
Ametherm, Inc. ................................................................... 18
Applied Power Systems ........................................................ 30
Avnet ................................................................................. IBC
CKE, Products by Dean Technology ...................................... 21
Coilcraft ............................................................................. BC
Crane Aerospace and Electronics .......................................... 3
CUI, Inc. ............................................................................. 20
EBG Resistors, LLC .............................................................. 23
International Rectifier, Inc. ................................................. IFC
IXYS .................................................................................... 16
IXYS Colorado ..................................................................... 29
Linear Technology Corporation .............................................. 1
Mean Well .......................................................................... 13
Mouser Electronics ......................................................... 5, 11
Payton America ................................................................... 22
Pearson Electronics ............................................................ 38
Positronic .............................................................................. 7
Powerex, Inc........................................................................ 31
TDK-Lambda ....................................................................... 17
Trim-Lok .............................................................................. 19
This Advertiser Index is printed as a courtesy only.
Power Electronics Technology is not responsible for errors or omissions.
40
Power Electronics Technology | March 2013
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“Superinductors”
That’s what engineers are calling our
new ultra-low DCR power inductors
Superconductors pass current
with virtually no resistance.
Our new XAL/XFL inductors
do much the same. Their DCR is
incredibly low: often half that of
similar size parts.
And their current handling
is equally impressive. Coilcraft’s
proprietary core material has a
soft saturation characteristic that
126%
higher
DCR
38%
higher
DCR
48%
higher
DCR
Competitors’ 4.7uH inductors
have much higher DCR per mm 3
than Coilcraft’s XAL5030.
prevents drastic inductance drops
during current spikes.
Unlike competitive parts,
these inductors don’t suffer from
thermal aging. And we give you
far more footprint options to
maximize PCB density.
To see what else makes our
new XAL/XFL inductors so super,
visit coilcraft.com/xal.
®
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