Soft switching forward-flyback converter with
one magnetic component and current doubler
rectification circuit
N. Frohleke,
P. Wallmeier,
J. Richter,
H. Grotstollen
Institute f o r Power Electronics and Electrical Drives
Department of Electrical Engineering, University of Paderborn
Warburger Strasse 100,33098 Paderborn, G e m n y
Tel.: 49-5;?51-603157,Fax: 49-5251-603157, E-mail: nfroel @pblea.uni-paderborn.de
Abstract - An active clamped forward-flyback converter
comprizing a current doubler scheme, which allows to boost
the application range of MOSFETs for efficient rectification in
low-voltage power supplies, is investigated first approximatively and second incl. commutations. Design equations and
issues are given for the magnetical integration of filter
inductors on the outer core legs of the transformer . The circuit
allows soft switching in the relevant operation range, while
keeping the reactive power low. Thus, the resulting efficiency
of nearly 90% using Schottky diodes is increased compared to
hard switching or resonant topologies.
involved analysis when applied in active clamped forwardflyback converters. But previous studies on the forward
converter with conventional rectification like [3], the thorough
evaluation of SR efficiency improvements given in [4] and the
integrated magnetics in [SI are a valueable base for this study.
I. INTRODUCTION
Since a number of years several efforts have been made to
decrease the power consumption of data processing electronic
circuits, despite their ever iincreasing clock frequency by
reducing the supply voltage below 5 V. Subsequently lowvoltage power supplies with reasonable effciency are required
more and more due to increased availability of low-voltage
logic IC’s.
I t
‘Q2
iQI
Fig. 1 Circuit diagram of the active-clamped
forward-flyback converter.
Since the resonant transition switching technique [1][2] with ZVS, being applied when using MOSFETs at elevated
11. ANALYSIS OF THE CIRCUIT
switching frequencies, and ZCS or ZVS, when IGBTs are
applied at elevated power levels - proved as appropriate means
A. Operation principle and simplified analysis
to raise conversion efficiency and to reduce EMI, questions
Using the following simplifying assumptions yields an
arise concerning the rectification components, which
contribute to more than half of the total losses and its drive analytical description of the circuit: Switching of diodes and
scheme. The latter depends liirgely on the power electronic MOSFETs is assumed to be ideal . Parasitics like stray induccircuit, when self-driven synchronous rectifiers (SR’s) are tances and component capacitances are ignored. Then the
applied [3]. As SR’s are only available with on-resistances in converter’s operation can be described by two intervals, based
the range of more than 6... 10 n d 2 at 25°C so far, yielding less on the explanation in [6].
losses than Schottky diodes with a blocking voltage of 30 V up
In an energizing interval the main switch Q1 and the
to currents of about 40 A only, it seems logical to conduct
respective Schottky diode or transistor Qq are conductive.
comparative investigations on current doubler rectification
About half of the energy is transferred directly towards the
schemes, when higher output icurrents are required. However,
secondary, magnetizing Lfl. The other 50% yield an
the treated rectification scheme (see Fig. 1) causes a quite increasing magnetization of the transformer and will supply
0-7803-3544-9196$5.00 0 1996 IEEE
1161
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the secondary circuit during the second interval, when
auxiliary switch Q2 and the corresponding Schottky diode,
resp. transistor Q3 are active. Clamp capacitor Ccl is approximately charged to a constant voltage U,, = U,/ (1 - D ) ,used
for the demagnetization of the transformer. At steady state the
auxiliary switch current iQ2(t) does not carry any mean value,
guaranteeing a constant ucl(t) at the beginning of this interval.
'prim
~ Q 4 = - =Uu c , - u 1
=
1
U,
z(--ul)
U 1-D
(4)
= 1u2
-D
A loss representing resistor Rv is assumed to approximate
conduction
losses
of
diodes
and
transistors:
R , = RQ1,2/ii2 + RQ3,4 . The threshold voltage of Schottky
diodes U , is considered as well. With ii as turns ratio and R,
as the load resistance the dc-voltage ratio results as:
UlJ2
__
U,
R,
-.D--.R,+R,
R,
UU,
R,+R,
U,
0'
I
I
I
I
I
0.2
0.4
0.6
0.8
1
D
Fig. 3 Blocking voltage of rectifiers and secondary voltage
versus duty cycle, U2 = 5.83V
The following equation gives an optimum turns ratio by
specification of the input ( Ulmin, U l m a x )and output voltage
range (ulmin,
uzmaX).
Optimum means, that the rectifiers are
charged by the same maximum blocking voltage:
'Q3max
-
'Q4ma.x.
Ulmin'lmax
0
0.2
0.4
0.6
0.8
I
Ulmin+ Ulmnx
1
1-uZmax
I
(5)
3. Analysis with non-ideal devices and commutations
Load current = 60 A, RL = 83 mQ
For the approximate analysis of the active clamped forwardflyback converter with magnetically integrated current doubler
rectification the following assumptions are set: ideal switching
of transistors and diodes and short commutations. On the
primary side parasitic capacitances of the power MOSFETs
The choose of turns ratio is strongly related to the blocking (Cess) are taken into account in order to approximate the
voltage the diodes have to withstand: A higher blocking diode switch-on and -off losses, since zero-voltage switching (ZVS)
causes higher on-state losses. And since the rectifiers on the does not occur through the entire working range. The subsesecondary side for output voltages between 2,l V - 5 V cause quent listed network topologies/intervals occur.
about 50% - 70% of the total losses, the blocking capability
Interval 1 (Energizing): The energizing interval is initiated
should be minimized.
by the termination of the commutation of D3 -> D4 (see
The potential drops on the power semjconductors are interval 8). Main switch Q1 is already switched on, while
considered by a modified output voltage U 2 . Eqn. (1). is MOSFET Qq is conductive, too. Since Q2 is blocking and
rewritten then as:
voltage uCl(t) equals uQZ(t) from the previous interval, the
remaining network is not affected by the capacitances.
- R,+R,
- DU,
Fig. 2 Output-input voltage ratio versus duty cycle for
varius levels of approximation:
a)ideal, b) R , = 16mR, c) R , = lOmQ, U , = 0.23V
U2 =
-U 2 + U F *
RL
U2 = U
The blocking voltage across e3is inspected only in the first
interval, while as Q4 conducts in the first intervall, the
maximum blocking voltage occurs in the 2nd interval:
Fig. 4
1162
Equivalent circuit for Interval 1, to 5 t 5 tl
Interval 3 (Zero-voltage switch-on of auxiliary switch Q2):
A positive current iQ2(t) is discharging CossQ2, until the
(6) inverse diode DQZ of the auxiliary transistor becomes
conductive and terminates this interval. Q , is switched on
actively with a short delay by the gate drive. As soon as the
transformer voltage uprimreaches zero the freewheeling of the
secondary diodes D, and D, is initiated. The resulting equiv( 8 ) alent cirsuit and summarized results are given as follows:.
Application of Kirchhoffs laws and a notation with e.g.
Index ltO describing quantity 1 at instant to yields:
UQl(t] =
ov
N1 = L OS
L + LmL p +L,L,,,/u
..2
+LpLop+LaPLas
Interval 2 (Commutation from Q1 -> Q2): When the main
transistor is switched off by the controller the drain - source
capacitance becomes effective. The voltage across the main
switch evolves sinusoidally, resulting in a drop of the transformer voltage. Consequently MOSFET Q4 blocks at once
(according to assumptions stated above) and the current
commutates to the inverse diode: resp. to a parallel connected
Fig. 6
diode D,. Simultaneously D3 begins to conduct, initiating the
commutation on the secondary side ,whereby the 3. interval is
started.
UCI
Equivalent circuit for Interval 3, t2 I t I t3
u Q l ( t ) = ( A 3 e s i n ( o O 3 . ( t - t , ) ) +K3)U
(11)
A3
i, (t) = uz3 cos (WO,. ( t - t , ) )
Fig. 5
Equivalent circuit for I[nterval2, tl I t I t2
With il(t) dividing according to the capacitance ratio the three
mesh and one node equations together with initial conditions
yield the following general solutions:
uQl
(t) =
(Jm.
i , (t) = (AA:
-. +.:
sin (oo2.( t - tl)
+ q,) + K,) ii
cos ( W,),. ( t - t,)
+ q2))/U
z2
The coefficients are calculated as follows:
z3 =
/(:),
A, = Z 3 . i i I l t 2 , K3 =
U1
U
Interval 4 (Commutation on the secondary side): The trans(9) former voltage (uprim)grows according to proportional to the
turns ratio uprim= U . U,/ (1 - D) .
The commutation on the secondary side is terminated as
(10) current is(t) attains the magnitude of filter current iLn(t). Then
the total current iLfl(t)+ iLf2(t)is conducted by diode D3.
Fig.7 depicts the equiv. circuit followed by calculative results.
1
Z2 =
@,
..2
Lm/u
K2 = --U1
p 2 9
Lfl +Los + Lm/U
U
UCl
Fig. 7
A, = Z,.UIlt,, B , = -K2
with A, < 0 we obtain:
p
',
with A , 2 0 we obtain:
'p2 =
=
+ atan
~t
E)
uQl
atani
1163
Equivalent circuit for Interval 4,t3 I t I t4
(t) =
(Jm.
s i n ( o W +( t - t , ) + q 4 )+K4)ii
(13)
oo4=
1
-
Z4 =
Jc4L4’
ForA,<O:
(p,
=
Interval 6 (Commutation of Q2 -> Ql): This interval is
introduced by the switch-off of the auxiliary switch Q 2 . The
main switchs current and voltage are depicted as follows:
&,
7c+atan
A, = Z,. i i I l f 3 , B, =
i
‘Clt3
i- 1‘
..
9
;forA420: (p4 = atan
Fig. 9
Equivalent circuit for Interval 6,
t5 5 t
t6
Interval 5 (Transformer demagnetisation): The clampingcapacitor voltage uCl(t) subtracted by U1 is applied across the
transformer. The capacitor voltage adjusts in a way reflecting
the magnetizing current iLm(t)at steady state. This is indicated
also by the original term ,,current mirror demagnetization“
given by Vinciarelli. Neglection of CossQl yields a resonant
circuit, consisting of C a and the resulting inductance of L,
and filter inducance Lf2. Initially, current il(t) is positive,
charging the clamp capacitor until the current decays to zero.
At this instant the highest voltage occurs across Q1. This is the
latest instant to switch-on the auxiliary switch. The input
current reaches negative values, whereby the transistor voltage
is lowered in order to reduce charge-down losses of the main
transistor. Fig.8 depicts the effective circuit state, followed by
the resulting expressions for uQl ( t ) and i, ( t ) .
As soon as the negative input current il(t) affects the transformer voltage to be zero, a secondary sided commutation is
initiated
Fig. 8
UCtU
Equivalent circuit for Interval 5 ,
u Q l ( t ) = ( ~ ~ . c o s ( w o( fs- f , .)
t4 I
t I t5
+ q S )+ K s ) i i
(15)
Interval 7 (Switch-on of the main switch): In order to
reduce charge-down losses and eventually EMC-problems,
associated with the switch-on of transistors under non-zerovoltage conditions, the inspection of the transistor voltage is of
predominant interest. The respective equivalent circuit is
UCl
uQ2
Fig. 10 Equivalent circuit for Interval 7, t, 5 t 5 t7
given in Fig.10, followed by calculative results.
u Q l ( t ) = (A7.sin(oO7. ( t - t 6 ) ) +K7)ii
A7
il(t) = -uz7
.cos(oo7.
1164
(t-t,))
(19)
a
zoom of
commutauon intervals
interval 5
interval 1
QI --> Qz
Qz --> QI
1
Z, =
/@,
A, = Z , . UIlt6, K, = U1
U
A practical analyis could be achieved with the following
assumptions:
..2
'ossQl
= 'ossQ2
Lcp =
" 'Cl,
..2
L ~ "s Lm3
For this, the main switch voltage is calculated approximately
to:
-1
i(t)
70
[AI
6o
50
tl
t4
t5
t6
t8
t6
t8
40
30
20
10
0
From the various means to accomplish ZVS, MOSFETs
with low output capacitance are selected, since they allow to
shorten soft-switching intervals and additionally the chargedown losses are reduced. Contrary, the means of increasing the
resonant inductance results in the negative consequence of
reduced turns ratio of the transformer, due to lenghtend
commutations. Alternatively, a higher current il(k) yields a
stronger discharging of the transistors capacitance, which is
supported also by the transformer with integrated filter
inductors.
t4 t5
tl
t5 t8
tl
t4
interval 5
1
ov
t it4
+ U, ( t -
t,)
t4
t5
t6
t4
t5
-
'8
L m +L c s u
Fig. 12 Calculated waveforms at a) U ,
and U2 = 5 V , I2 = 60A.
1165
t8
t8
2
0
-1
-2
(22)
t6
commutation intervals
Q1-z QZ
QZ --> QI
tl
[AI
i, ( t ) = I l t ,
-
t5
zoom of
b
interval 1
Equivalent circuit for Interval 8, t7 5 t I fg
( t ) =-
tl
-2
An evaluation of the symbolic treatment of the circuit by
Mathematica is given in Fig. 12, showing essential voltage and
current waveforms. Relevant design equations incl. loss determination for the individual components are derived in [7],
using symbolic calculations.
UQ,
t5t8
2
1
0
-1
Interval 8 (Secondary sided commutation): After switching
on the main transistor the transformer voltage equals the input
voltage, affecting a fast commutation from Q3 to Q 4 . With
iQ3 becoming zero, this interval terminates. The respective
equivalent circuit is depicted in Fig. 11, followed by results for
uQl(t), il(t).
Fig. 11
tl
= 200V, b)
t6
t8
U, = 400V
III. F
TEGRATION ON
The specific rectification circuit (,,hybridge") allows an
integration of the filter inductors on the core of the transformer, aiming on cost and volume reduction. In addition the
filter and secondary winding can be integrated, using one
mutual current path, whereby currents compensate each other
in half of the winding (see Fig. 14). By this means copper
losses can be minimized.
IL
1U,
i L m ( f 0 ) = 2ii 2 L mDTS
I,
iLm(tl) = 2ii
1 U1
+ -DTS
2Lm
(27)
As the load current is set and the secondary turns are chosen
to one for minimization of copper losses the only remaining
design variable of the first term of equation (25) is the air gap,
which is the major disadvantage of this concept. Means to
reduce the second term are constrained to the switching
frequency and the core cross section.
energizing
demagnetisation
iLfl= -is
iLB=is
Fig. 13 Principle of the transformer with integrated
filter inductors.
(N, number of respective turns; 6,magnetic flux)
The assembly depicted in Fig. 13, Fig. 14 leads to an
asymmetrical magnetic flux density. In order to study the
effects of filter currents on the design of the single magnetic
component, taking care of the input voltage range, the
following simplifying assumption was set concerning the air
gap:
,y'
iLfl 4-iLD
iLf1
i- iLf2
>> ' F e / p r
The analysis of the transformer is based on the simplified
analysis of the circuit, given in 1I.A. With losses being
included by consideration of k,.
k , = RLOCId/
(RLoad+Rv) (RLoa,..loadresistance).
The maximum magnetic flux density is derived via a reluctance model, using
Fig. 14 Reduction
of copper losses
using a mutual current branch for filter
and secondary
winding.
0
and the relations for the currents (eqn. (25)-(27)):
1166
IV. EXPERIMENTAL, VERIFICATION
was adjusted at 10% of the load current.
a
A. Experimental set-up
b
Measurement results are obtained by a prototype forwardflyback converter for the following specifications:
200V to 400V
Input voltage range, U1:
Output ratings, P2 at U2/12: 300W at 5V/60A
Switching frequency, f,:
380 kHz
Components used:
Main switch Q , : 2*BUZ307, Auxiliary switch Q,:BUZ 307
RDS(,,)
= 5.2a11w0c, Cos, = 9 0 ~ F
Schottky diodes: 82CNQ30, U, =: 0.23V, rF = 3.5mQl
Fig. 16 Input- and main switch current at 30 A load current
L, = 282pH, Lfl = Lrz = 291nR, L,,+u L,, = 5.1pH
Fig. 15a: None of the primary sided MOSFETs are
conductive after switching off the auxiliary switch. Only the
output capacitances are effective at this instant. The input
current il is shared by the capacitances of Q, and Q,. This
behavior is indicated by the marked cycle "A". The diffference
in current Ai is charging the output capacitance of Q,.
Fig. 15b: If the delay time is too low, Q , and Q , are
conducting simultaneously, which leads to a partial discharge
of the clamp capacitor, indicated by the current peak of the
main transistor.
(Measured with disconnected transformer)
C. Calculated and measured results
B. Practical measurements
Fig. 17 depicts the calculated efficiency using Schottky
diodes while Fig. 18 shows the measured efficiency. The
strange result indicating a higher measured efficiency as calculated is due to the following:
Synchronous rectifiers (SR): IRL3803, R,,(~~, = 8.3nr12'loooc
Diodes paralleled to SR: 20CTQ30, U, = 1V ' ~ A
Clamp capacitor: 10nF, lOOOV
Integrated magnetics:
ETD39, reduced height = 23mm, core material: N67
primary: 22 turns, 2.4mm*0.25mim
secondary + filter: 1 turn, 9*5Smm*.2Smm
air gap: 1, = 0.5"
..2
a
n k
nun
I~SMSIS
b
HI np.
First, the real RD,(,,) of power MOSFETs under use were
measured less than the assumptions for calculations. The same
applies also for Schottky diodes concerning on-state-losses.
Second, the core losses were calculated at a temperature of
25OC, while the actual core losses are lower at increased
switch-on time
operation temperature.
~
, ~ ~~ ~ , ~ ~, = 5 V / d i v .
uos,Q1= lOOV/div., ~ ~ ~ , ~ ~ = S V / u
d i v .~= lOOV/div.,
t :I lOOns/div.
t = 400nddiv.
Fig. 15 Main switch voltage uel(t) at 200 V input voltage
Fig. 15a: According to the quasi zero-voltage-switching, the
drain-source-voltage is low when switching on. At
U,, = U,, the MOSFET's Miller-capacitance is predominantly charged by the gate-current, with UGs remaining
constant.
Fig. 15b:The main switch is switched on with a constant delay
after the turn off of the auxiliary switch. The measured oscillogram was taken at rated power. Q, is not optimally switched
on at higher load current, since its optimum switch-on instant
II
I
I
1
I
100
150
200
250
%
SO
Pou,[WI-
Fig. 17 Calculated efficiency using Schottky diodes
1167
300
92
intervals. During commutation the externally connected low
power Schottky diodes produce higher on-state losses than the
SR’s. Paralleling of SR can of course extend this range, but
costs rise too for the power supply module, since the cooling
sinks are not included.
1
%
V. CONCLUSIONS
50
100
150
200
250
300
POUT[-
In this publication an approximative and a thorough
analysis of the soft switching forward-flyback converter is
given which comprises only one magnetical component (with
the filter inductors integrated on the transformers core) and a
current doubler rectification circuit.
Due to the low reactive power and soft switching in the
essential operation range, the resulting efficiency is increased
The higher efficiency for lower input voltage (s. Fig. 17 and by about 5% compared to hard switching PWM controlled or
Fig. 18) results as the on-state losses of the Schottky diodes resonant converters. The profitable application range of
are higher due to increased RMS-value of the current, as the Synchronous Rectifiers, which can compete with low-loss
charge-down losses as well as core and copper losses of the Schottky diodes, is extended due to the current doubler rectifimagnetical components are higher at increased input voltage. cation circuit. But still Schottky diodes seem to be the only
Hence, the circuit shows similar efficiency characteristics like components for low-voltage rectification, if output currents are
the asymmetrical PWM controlled bridge (AB) topologies [11, in excess of 35 Ampere and if costs count.
which also belong to the resonant transition class of
converters. The gained efficiency increase of the inspected
forward-flyback converter compared with the just mentioned
AB is about 3%, mostly affected by the application of
Schottky diodes with a blocking voltage of only 3 0 V in the
previous circuit.
Fig. 18 Measured efficiency using Schottky diodes
REFERENCES
[l] N.Frohleke, A. Fiedler, H. Grotstollen, B. Margaritis: Investigation of PWM controlled, resonant transition converters with
asymmetrical duty cycle, IEEE IECON 1995, pp.429-434
[2] G. Hua, E C. Lee: Evaluation of switched -mode power
88
conversion technologies, Proc. of the 1st int. Power Electronics
and Motion Control Conference (IPEMC), 1994, Beijing, pp.
86
12-26
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I
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84
[3]
J.A Cobs, J. Uceda: Low output voltage dc/dc conversion,
/DR
IEEE IECON 1994, pp. 1676-1681
[4]
M.M.Jovanovic, M.F. Zhang, F.C. Lee: Evaluation of
I
I
I
I
I
Synchronous-Rectification efficiency improvement limits in
forward converters, IEEE Trans. Ind. Electronics, vo1.42, no.4,
50W lOOW 150W 200W 250W 3 6 W
Aug. 1995, pp. 387-397
10A
20A
30A
60A
40A
50A
[5] C. Peng, M. Hannigan, 0. Seiersen: A NEW EFFICIENT HIGH
P O U P Io,,
FREQUENCY RECTIFIER CIRCUIT, HFPC, June 1991,
pp. 237-243.
Fig. 19 Comparing measured efficiency between Schottky
[6] I. D. Jitaru: HIGH EFFICIENCY DC-DC CONVERTER,
diode rectification (DR) and synchronous rectification
Power Conversion, June 1994, pp. 109-116
(SR) for an input voltage of 300 V
[7] J. Richter: Quasi zero voltage switching forward-flyback
Fig. 18 shows the efficiency graphs for an input voltage of
converter with active clamp and current doubler rectification,
Diploma thesis (in German), University of Paderborn, Institude
300 V versus load current using only one synchronous rectifier
for Power Electronics and Electrical Drives, Dec. 1995.
(IRL3803) for Q3 and Q4in comparison with Schottky diodes.
Note, that a comparative advantage for synchronous rectifiers
results only below 30 A to 40 A and that additional losses are
caused with high output currents due to longer commutation
I
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1168