Improved Performance Analysis of Clamp Circuits with Flyback

International Journal of Emerging Technology and Advanced Engineering

Website: www.ijetae.com ( ISSN 2250-2459, Volume 2, Issue 1, January 2012)

Improved Performance Analysis of Clamp Circuits with

Flyback Converter

T.Halder

Kalyani Government Engineering College, Kalyani, Nadia, West Bengal, PIN-741235, INDIA tapas_haldar@yahoo.com

,



Abstract --This paper presents the overall improved performances of clamp circuits based on software simulation that can be used to repress these transients on both the primary and secondary side. The Flyback dc-dc converter topology results in important cost and space savings for single or simple several output power supplies with high output voltage for power levels up to 40W with natural isolation.

Flyback topology accumulates and flies energy using a couple inductor or transformer, which due to realistic restrictions and leakage of the couple inductor can cause abnormal voltage transient spikes during the switching cycle across the power switch.

Key words

—Flyback Converter, Clamp circuits, Design,

Analysis and Result

I. I NTRODUCTION

Some published literatures reveal soft switching schemes to reduce switching losses, and EMI for high switching power converter applications. In many cases and the articles [1-9] elucidate the performances of the dc to dc power converters using soft switching techniques where auxiliary switch, supplementary control unit and passive components are enforced to boost up the circuit efficiency of the power converters by virtue of which circuit complicacy and surplus cost involvement arise haphazardly.

Even some papers [2, 3] disclose upto 96% efficiency without counting switching losses of auxiliary switch and related other passive components. In this paper, simple snubber or clamp circuits are discussed without auxiliary circuit to progress the performances of flyback power converter. The leakage inductance of the flyback power transformer of an isolated dc-dc converter may significantly affect its efficiency and cost. Using lossless recovery clamp circuits improves efficiency and avoids problems arising from transformer leakage-inductance spikes. In the dualtransistor forward (full-bridge and half bridge) converters, where the switching transistors are connected in series, energy stored in the transformer leakage inductance may be recycled through the clamping diodes that are an integral part of this architecture.

However, in the “single-ended” topologies, such as forward, flyback, push pull and current-fed, in which the switching transistors are connected to electrical ground, the energy stored in the leakage inductance may significantly affect a converter’s performance. If no measures are taken to recover the energy stored in this inductance, the energy will be dissipated either in the converter components or in the snubber or the clamp circuits, absorbing voltage spikes caused by the leakage inductance. Energy stored in the leakage inductance of the power transformer during each switching cycle may be represented by loss equation. Clamp circuits are an essential part of power electronics. Clamp circuits or snubbers are small networks of parts in the power switching circuits whose function is to control the effects of circuit reactance. Clamp circuits progress the performance of the switching converter circuits with power converters and consequence in higher reliability, higher converter circuit efficiency, higher switching frequency, smaller size, lower weight, and lower EMI (Electromagnetic

Interference). The fundamental objective of a resistor capacitor and diode (RCD) clamp is to absorb energy from the reactive elements in the circuit. The benefits of this may comprise circuit damping, controlling the rate of change of voltage or current or clamping voltage overshoot. In performing these functions a clamp circuits limits the amount of stress which the switch must tolerate and this increases the reliability of the switch. When a clamp circuit is appropriately designed and implemented the power switch will have lower average power dissipation, much lower peak power dissipation, lower peak operating voltage and lower peak operating current. This article describes some of the various non dissipative passive current snubber or clamp circuits. Theses clamp network elements are very inadequate to resistors, capacitors, inductors and diodes.

Active clamp include transistors or other active switches, often necessitate a major amount of additional circuitry and introduce another level of parasitic which must be dealt with (usually with a passive snubber). However, active clamp is apt in some typical applications. A fine example of an active clamp is that they are used, how they function, how they are designed and what their limitations are clampers may be either passive or active networks with flyback converter.

1


II. C LAMP O


R S NUBBER F UNDAMENTALS

In most cases, however, the optimum efficiency level is reached with small snubbers long before a full resonant

Each snubber or clamp in this discussion will be shown in an example circuit which is as generic as possible. Fig. 1,

Fig. 2 and Fig. 3 show the basic buck, boost and flyback converter circuits respectively drawn so that the switch is approach is necessary.

The effect of a snubber or clamp circuit on the switch may be viewed as changing, or shaping, the load line on the always grounded. The figures which follow will generally show a snubber in relation to a generic grounded switch so

Fig. 1, Fig. 2 and Fig. 3 may be used to pertaining the switch for a particular purpose. An inductor in series with the switch, a current snubber or clamp circuit, presents the switch with an inductive load at turn-on so that it switches clamper to most other converter topologies. In this approach the switch and the clamp or snubber circuits may be thought of as a single unit, a snubbed switch, which may on with zero current. This is a modification of the load which would typically be somewhat capacitive at turn-on.

The same is true at turnoff with a voltage snubber. At turnoff the load on the switch will typically look inductive so a capacitor in shunt with the switch, a voltage snubber, will be used in almost any switching regulator topology. Not all snubber or clamp circuits are pertinent a particular problem.

Fig. 2 are clamped inductive loads and these circuits do not need voltage clamp snubbers since that working function in change the load line to be capacitive so that the switch can turn off at zero voltage. inherent in the topology.

III. N ON DISSIPATIVE C URRENT S NUBBER OR C LAMP

The essential theory of a current snubber is that energy is collected in an inductor as part of its function of controlling the rate of rise of current in the switch. Non-

Fig. 1 Buck converter

Fig. 2 Boost Converter dissipative current snubbers are very analogous to their dissipative equivalents. The basic functionality is the not identical; an inductor is placed careful way in series with the switch to control the rate of rise of current. In a nondissipative current snubber the energy stored in the inductor on each cycle is transferred back to the input or ahead to the output instead of being dissipated. There are several methods for removing the energy from then inductor on each cycle and some of the more ordinary schemes are discussed in this part.

Fig. 3 Flyback Converter

The snubber or active clamps which are apt in these power converter topologies are voltage and current rate of rise control snubbers. Controlling the rate of rise voltage and current, clamping voltage and controlling resonances reduces stress on the switch. Snubbers can control the voltage and current to the point where switching occurs at zero voltage and zero current and this raises the consistency of the power stage significantly. Taken to the extreme zero voltage and current switching becomes resonant power conversion. This becomes necessary when the circuit parasitic become large relative to the power level. High voltage outputs are one example.

IV. F LYBACK R ESET C URRENT C LAMP

One of the clearest ways to get the energy out of the inductor is to put an additional winding on it. This allocates the energy to be aimed at somewhere and endows with a controlled overvoltage state on the switch set by the turns ratio and the recovery voltage. Fig.4 shows the basic snubber and Fig.5 and Fig.6 show two ways of hooking the snubber to the generic converter of Fig. 2. The configuration preferred depends on whether it is a buck, flyback (Fig. 5) or boost (Fig.6) converter. In all cases the energy is transferred to the load although other connections could transfer the energy back to the input.

The design of this type of snubber or clamp circuit is rather simple.

The value of the primary inductance is the same as it would be for a dissipative snubber and the voltage which is added to the switch during reset is determined by the-

2



VI. I SOLATED S INGLE S WITCH ZVS-ZCS F LYBACK

C ONVERTER

Fig. 4 Schematic Flyback converter

The flyback from the Fig.7 is the simplest among isolated

ZVS converters. The above analytical results are still substitution output voltage with the value is given by

Vo= N

S

/N

P

*V s

(1)



V. D ESIGN S PECIFICATIONS O F T HE F LYBACK C ONVERTER

The flyback converter and the RCD Clamp are designed in the conventional way at discontinuous mode of operation

(DCM) to find out the value of active and passive components of the converter.

Supply voltage 90-250V dc

Power =40W

Output voltage= 24V

Primary inductance, L p

= 400uH

Secondary inductance, L s=

42uH

Rating of MOSFET 800V, 5A

Diode UFR

R1= 10K and 10W

C1=0.1 uF and 800V disc type

BJT 800V, 1A

Output capacitor of the flyback converter 200uF and 40V

Switching frequency, f= 70 kHz

L1=10uH

3

Fig. 7 Basic Flyback converter

In the above formulae N s

, N p

and V s

are secondary, primary turns of the flyback transformer and supply voltage respectively.

Voltage across flyback diode (V d

) is given by

V d

=V

0*

N (2)

Where, turns ratio of the flyback transformer, N= N s

/N p

VII. R EALIZATION OF ZCS A ND ZVS

ZCS can eliminate both the switching losses at turn-off and reduce the switching losses at turn-on. As a relatively large capacitor is connected across the output diode during resonance, the converter operation becomes insensitive to the diode’s junction capacitance. The major limitations associated with ZCS when power MOSFETS are used are the capacitive turn-on losses. Thus, the switching loss is proportional to the switching frequency. During turn-on, considerable rate of change of voltage can be coupled to the gate drive circuit through the Miller capacitor, thus increasing switching loss and noise. Another limitation is that the switches are under high current stress, resulting in high conduction loss. It should be noted that ZCS is particularly effective in reducing switching loss for power devices (such as MOSFET) with large tail current in the turn-off process. ZVS eliminates the capacitive turn-on loss. It is suitable for high-frequency operation. For singleended configuration, the switches could suffer from excessive voltage stress, which is proportional to the load.


For both ZCS and ZVS, output regulation of the resonant converters can be achieved by variable frequency control.


ZCS operates with constant on-time control, while ZVS

Worst case takes place when the drain to source voltage,

V

DS

(t) and I d

(t) drain current instantly rises up at the switch opening (case-1).

If some clamp exists to slow the voltage rise, the operates with constant off-time control. With a wide input and load range, both techniques have to operate with a wide switching frequency range, making it not easy to design idealized case-2 can happen with more favourable loss accounts, resonant converters optimally. P

SW

, off = f sw

*[

 t



0

I

D

( t ) V

DS

( t ).

dt ] (3)

The most important hindrance in reducing the size and

Fig. 8 Hard switching and soft witching waveforms

Turns ratio and either the source or load voltage. The major problem with this type of snubber is the leakage inductance between the primary and secondary of the inductor. This leakage inductance can cause a large voltage spike across the switch. This type of snubber has been used effectively but it is generally used with high power converters where the rise and fall times are slow or where rate of rise voltage snubber or clamp are also used. Simple

RC damping networks are usually needed across both the primary and secondary of the inductor.

VIII. L OSS M ODELING O F T HE F LYBACK C ONVERTER

In order to compute the switching power loss, considering the overlapping section as a triangle of small time (∆t) width shown below, the average power dissipated by the MOSFET (M1) at turn off and turn on condition is simple triangle area under the curve, as shown below. case-1 case-2 weight of Switch Mode Power Supply (SMPS) are the switching losses, which are frequent, switching voltage and current conditional. The switching power losses in power switch like power MOSFET. Firstly, turn-on losses,

∆P1= 0.5f

sw

C p

V

2 sw (4)

where f sw and V sw

are the switching frequency and voltage across the main switch of the flyback converter respectively, caused by a discharge of both parasitic capacitance (C p

) of the power switch and the primary winding of the power transformer and secondly, losses determined by the turn-off dynamics of the main switch of the flyback converter as shown below.

∆P1=1/6f sw

I sw

V sw

∆t (5)

For example, in conventional single switch flyback converter with pulse width modulation (PWM) and constant switching frequency of 70 KHz of switch current I sw

=2A are very important issue in power converter scheme here. In order to minimize the switching losses, snubber circuits constructing a turn-off network are usually used. They are based on reducing the voltage front slope during turn-off, but commonly (particularly, RCD (Resistance Capacitance and diode) generating surplus losses of

∆P3=0.5f

sw

C1V

2 sw

(6)

An ordinary clamping circuit added for control of parasitic burst of voltage can also be sources of losses, caused by the primary leakage inductance of the flyback power transformer

∆P3=0.5f

sw

L e

I

2 sw

(7)

Here, Le is the leakage inductance connected in series of the flyback transformer at the primary side. These drawbacks are absent in the majority of quasi-resonant ZV-

ZCS-converters, in which turn-on losses are absent, and the favourable trajectory of turn-off of power switch can be ensured by artificial increase of parasitic capacitance playing a greatly practical role here. Since a stage of switching losses and electromagnetic interference (EMI) directly depends on current and voltage differentials at switching intervals, ZV-ZCS converters by these parameters are the dominant of acknowledged.

4



Inopportunely the most important shortcoming of conventional types of ZC-ZVS converters [8] is resonance

Note that the highest voltage transient occurs when switching the highest level of current. nature of switch voltage, whose amplitude is reliable of load current, minimal value of which is determined by the capability of resonance capacitor recharge. Such characters

This voltage overshoot may cause excessive power dissipation in the MOSFET even if it is within safe operating limits of the device and thus reduce the total create involved using them as off-line power supplies with wide load range.

There is scrupulous set of ZVS-converters with quasisquare wave voltage waveforms, lacking the above drawback, which are preeminent for implementation as offline power converters. As is recognized, a characteristic efficiency.

Also, it may cause instability in the loop due to (dv/dt) induced spurious turn-on of the MOSFET as a result of the transient voltage spike. The effective spike voltage V peak is dependent on the input current, parasitic capacitance (C p

) of the flyback transformer and parasitic drain to source capacitance (C oss

) of the power MOSFET (M1). feature of the ZV-ZCS-converters is commonness of a converse current prior to a direct current of power switches on an interval of conductivity. This resource significantly simplifies a snubber circuit, which is abridged to the sole capacitor connected in parallel to a primary winding of the power transformer of the flyback converter, and the losses allied with its recharge are not present. The most identified of them are classical resonant converters, operated above resonance frequency [5]. Since the switching current of the final is quasi-sinusoidal, it will not survey because the proposed presently innovative proportional charge controlling practice for power switches are not pertinent at this point.

X. S ECONDARY L EAKAGE I NDUCTANCE AND THE

D IODE

R ECTIFIER

The transformer secondary leakage inductance may couple with the reverse recovery current I

REC

of the output rectifier diode to cause ringing when the diode turns off.

The transformer secondary leakage inductance L2 and the capacitance C

D

of the rectifier determine this resonant frequency. The ringing may generate significant radiated and conducted noise. There is typically very little loss in the

IX. P RIMARY L EAKAGE I NDUCTANCE

S WITCH

A ND T HE MOSFET resonant circuit so this network will cause many cycles of ringing after the spike. The ringing can therefore affect the current sense signal used by the controller in a flyback configuration. The overshoot caused by this ringing may exceed the diode voltage rating and cause damage to the diode. The maximum positive voltage across the rectifier can be estimated as:

The primary leakage inductance L

1 including L e

in a flyback does not participate in the primary to secondary energy transfer and so it has a negative impact on efficiency. Since the leakage inductance does not find a path for the current built up in it during the switch on-time, it leads to a voltage spike at the turn-off of the MOSFET and also delays the transfer of power from the primary to the secondary. The parasitic elements in the transformer and the switch determine the magnitude of this voltage spike. At high output voltages the circuit parasitic become large relative to the amount of output power delivered. The primary leakage inductance, L

LP

, and the primary winding capacitance in the transformer, C

P

, and the output capacitance, C

OSS

, of the MOSFET form a parasitic LC network. The peak voltage is given by:

V

PEAK



I p



C

P

L

1



C

OSS





V in



V o

N (8)

V

PEAK , S

 transformer.

I

REC

XI. P

L

2

 

D

OWER L



V in

OSS O

N

F

(9)

The peak voltage of the rectifier diode of the flyback converter arises due to presence of the reverse recovery current of the rectifier diode, effective capacitance of the rectifier diode, input voltage and turns ration of the flyback

RCD C LAMP

In the clamp, Resistor capacitor and diode (RCD) mode the purpose of the snubber is to clamp the voltage during turn-off at the drain of the MOSFET as shown in the Fig. 8 in the encircled part. The parallel R1C1 circuit may be returned to ground or to a voltage other than ground (i.e.,

Where I

P

is the current level circulating at the time the

MOSFET turns OFF, N is the ratio of secondary to primary turns, V in

is the input voltage, and V

O

is the output voltage.

The square root term in the above equation represents the characteristic impedance of the parasitic LC network. input voltage if the drain can go above input voltage) since this will reduce the power dissipation in the resistor. The

MOSFET switch itself will have to sustain the peak power dissipation during turn-off.

5



The value of the capacitor, C

CLAMP

, and resistor, R

CLAMP

, is based on the energy stored in the parasitic inductance, as this energy must be discharged into the RC network during each cycle. The voltage across the capacitor and resistor sets the clamp voltage, V

CLAMP

. The turn-off waveform at the drain of the MOSFET switch is as shown in Fig. 8.

Fig. 9 Flyback converter with RCD Clamp

XII. B ASIC O PERATION O F T RANSISTORIZED C LAMP

C IRCUIT

Fig. 8 Schematic Voltage across MOSFET during off condition

This implies that the power dissipated in the clamp at turn-off is given by

P

CLAMP

= 0.5 V

CLAMP

I

CLAMP

Δt f (10)

Where, f is the switching frequency. However,

Δt = (L1 I

P

) /(V

CLAMP

- V

O

N) (11)

The power dissipated in the clamp is through the resistor.

Therefore

R1=R

CLAMP

= [2 V

CLAMP

(V

CLAMP

- V

O

N)] /( L

1

I

P

² f) (12)

It is important to minimize the ripple, V ripple

, superimposed on V

CLAMP

to keep the MOSFET drain voltage close to the clamp voltage. The minimum value for the capacitor,

C

CLAMP

, is therefore,

C1=C

CLAMP

= V

CLAMP

/ (V ripple

R

CLAMP

f) (13)

The capacitor, C

CLAMP

, should have a low ESR and low inductance to handle the high peak currents during turn-off.

The RC time constant of the clamp should be chosen much larger than the switching period of the MOSFET. The series diode used in the clamp should be able handle the peak currents and must be a fast turn-on type. Topology with a clamped inductive load since it does not reduce ringing.

The encircled part of the Fig. 8, C

CLAMP and R

CLAMP are shown as C1 and R1 respectively. RCD snubber dissipate the energy due to presence of the resistance but it help to protect the switch against the spurious spike coming from the leakage energy of the flyback transformer. It also reduces converter efficiency.

Fig. 10 Flyback converter with Transistorized clamp circuit

The main difference of the isolated compared to basic topology is large value of leakage inductance Ls of the high frequency power transformer, which can cause a destructive surge voltage on the power switch. To tyrannize this outcome, it is invented by the innovative snubber circuit, which causes no power losses in converter due to its energy recovery character in Fig. 9 as shown below. From the linearised simple diagrams, the expression for evaluating the surge voltage can easily be derived on condition of zero voltage balance in L1 and zero current balance in C1 during the period assuming the equal forward voltage drop on the diode D1 and the base-emitter leads of transistor Q1, we finally receive very simple expression for calculating by the flowing empirical equation as shown below.

L1=7*Le*∆V

BE

/9∆V s

(14)

6



Where, ∆V

BE

, is the change of base to emitter voltage of the power transistor (Q1). When the power MOSFET of the flyback converter is on, the leakage energy (∆P3) will be discharged by the inductor L1. When emitter voltage will be negative and collector terminal voltage will be positive.

Then this transistor (Q1) will be on, store energy will send to source. When power switch of flyback converter is off without consideration of leakage voltage, the voltage (V off

) across it, given by

V off

=V in

+NV o

(15)

Energy stored by the capacitor Coss is given by

E

2

=0.5C

oss

V

2 off

(16)

E

2

energy is slowly discharged through inductor, L1, giving rise negative emitter voltage for which the transistor

(Q1) will be on to send the leakage energy to source. The capacitor (C oss

) enhances the zero voltage switching operation and to improve the converter efficiency.

The similar circuit with transistor (Q1) can be placed across the output diode of the flyback converter to reduce the partial conduction loss of diode by pronounced ZCS operation. C1 is the voltage dumping capacitor to maintain a voltage source when discharge current coming from the inductor (L1). Diode D1 acts here voltage blocking diode to prevent the current path via source to flyback MOSFET when the flyback MOSFET is on. Neglecting on resistance of MOSFET, voltage is being nearly zero across he flyback switch when the flyback switch is on.

Fig. 12 Switching waveforms of the Flyback converter

Similarly, Fig. 12 depicts the gate pulse, drain current of the main power switch of the MOSFET (M1) and voltage across M1 For a load 1A for input voltage 190V dc.

The encircled part of the Fig. 10 describes the transistorized snubber circuit of the flyback converter to achieve the zero current switching at turn on condition of main switch (M1) of the flyback converter and zero voltage switching (ZVS) at turn off condition of the main

MOSFET of the flyback converter. The Fig. 11 and Fig. 12 that are simulated waveforms confirm the ZCS at turn on condition and ZVS at turn off condition of the MOSFET

(M1). The leakage energy is recovered by the placement of transistorized snubber circuit with less current and voltage stress across the switch.

XIII. R ESULT

Fig. 11 switching waveforms of the Flyback converter

Fig. 11 depicts the gate pulse, drain current of the main power switch of the MOSFET (M1) and voltage across M1 For a load 0.8A for input voltage 110V dc.

7

Fig. 13 Load vs. efficiency covered for using RCD Clamp



In the Fig. 13, X-Axis represents percentage efficiency and Y-Axis represents the output power of the flyback converter using hard switching.

Fig. 14 Load vs. Efficiency curved for using transistorized clamp

In the Fig. 14, X-Axis represents percentage efficiency and Y-Axis represents the output power of the flyback converter using soft switching scheme (i.e. using transistorized clamp circuit). At different load of the flyback converter efficiency attained 94%. The above experimental graphs show the confirmation of the ZV-ZCS with high efficiency of the prototype advanced flyback converter

XIV. C ONCLUSION

This scheme removes first order (drain circuit) switching losses for high efficiency and consistency. The dissimilar

ZVS and ZCS design optimization method facilitates exercise of a superior MOSFET expire immensity for

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Author: reduced voltage and current stress across the switch, conduction losses, switching losses and higher conversion efficiency and lesser EMI. Output flyback converter diode’s

T.Halder is presently attached with Research Scholar in

Jadavpur University and Assistant Professor in Kalyani reverse recovery effects and losses are reasonably eradicated. Comparatively trouble-free power phase with

Government Engineering College. His interest includes low parts count compared to all other isolated ZVS and

ZCS topologies. Second order transfer function performs power electronic converters & Electrical Machine. He has several years’ industrial and teaching experience in power and energy sector and power electronics system. He has like a typical DCM flyback converter but with enhanced damping and can lodge ample line voltage ranges and output voltage regulations. Predestined to become an industry standard topology as a higher circuit efficiency published a paper in international journal of electronics,

Taylor & Francis, USA and three in IEEE conferences, six national conferences and one international conference. His

BEE and MEE degree from the Jadavpur University, upto 94%, light weight ,compact size and higher frequency alternative to conventional off line flyback converter simple implementation using timing signals from ZVS and ZCS

Kolkata-32, India drive which can also in attendance compel power phase.

References

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Improved Performance Analysis of Clamp Circuits with Flyback

Improved Performance Analysis of Clamp Circuits with

Flyback Converter

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