LT1996
Precision, 100µA
Gain Selectable Amplifier
U
FEATURES
DESCRIPTIO
■
The LT®1996 combines a precision operational amplifier
with eight precision resistors to form a one-chip solution
for accurately amplifying voltages. Gains from –117 to
118 with a gain accuracy of 0.05% can be achieved without
any external components. The device is particularly well
suited for use as a difference amplifier, where the excellent
resistor matching results in a common mode rejection
ratio of greater than 80dB.
■
■
■
■
■
■
■
■
■
■
■
Pin Configurable as a Difference Amplifier,
Inverting and Noninverting Amplifier
Difference Amplifier
Gain Range 9 to 117
CMRR >80dB
Noninverting Amplifier
Gain Range 0.008 to 118
Inverting Amplifier
Gain Range –0.08 to –117
Gain Error: <0.05%
Gain Drift: < 3ppm/°C
Wide Supply Range: Single 2.7V to Split ±18V
Micropower Operation: 100µA Supply
Input Offset Voltage: 50µV (Max)
Gain Bandwidth Product: 560kHz
Rail-to-Rail Output
Space Saving 10-Lead MSOP and DFN Packages
The amplifier features a 50µV maximum input offset
voltage and a gain bandwidth product of 560kHz. The
device operates from any supply voltage from 2.7V to 36V
and draws only 100µA supply current on a 5V supply. The
output swings to within 40mV of either supply rail.
The internal resistors have excellent matching characteristics; variation is 0.05% over temperature with a guaranteed matching temperature coefficent of less than 3ppm/°C.
The resistors are also extremely stable over voltage,
exhibiting a nonlinearity of less than 10ppm.
U
APPLICATIO S
■
■
■
■
The LT1996 is fully specified at 5V and ±15V supplies and
from –40°C to 85°C. The device is available in space
saving 10-lead MSOP and DFN packages. For an amplifier
with selectable gains from –13 to 14, see the LT1991 data
sheet.
Handheld Instrumentation
Medical Instrumentation
Strain Gauge Amplifiers
Differential to Single-Ended Conversion
, LTC and LT are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners. Patents Pending.
U
TYPICAL APPLICATIO
Rail-to-Rail Gain = 9 Difference Amplifier
15V
450k/81
VM(IN)
∆VIN
VP(IN)
INPUT RANGE
±60V
RIN = 100kΩ
450k/9
+
450k/9
4pF
–
+
LT1996
450k/27
450k/81
40
35
PERCENTAGE OF UNITS (%)
450k
450k/27
–
Distribution of Resistor Matching
VOUT = VREF + 9 • ∆VIN
SWING 40mV TO
EITHER RAIL
LT1996A
G = 81
30
25
20
15
10
450k
5
4pF
0
–0.04
VREF
–15V
0
–0.02
0.02
RESISTOR MATCHING (%)
0.04
1996 TA01
1996 TA01b
1996f
1
LT1996
W W
U
W
ABSOLUTE
AXI U RATI GS
(Note 1)
Total Supply Voltage (V + to V –) ............................... 40V
Input Voltage (Pins P9/M9, Note 2) ....................... ±60V
Input Current
(Pins P27/M27/P81/M81, Note 2) .................. ±10mA
Output Short-Circuit Duration (Note 3) ............ Indefinite
Operating Temperature Range (Note 4) ...–40°C to 85°C
Specified Temperature Range (Note 5) ....–40°C to 85°C
Maximum Junction Temperature
DD Package ...................................................... 125°C
MS Package ..................................................... 150°C
Storage Temperature Range
DD Package .......................................–65°C to 125°C
MS Package ......................................–65°C to 150°C
MSOP–Lead Temperature (Soldering, 10 sec)...... 300°C
U
W
U
PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
TOP VIEW
P9
1
10 M9
P27
2
9 M27
P81
3
8 M81
VEE
4
7 VCC
REF
5
6 OUT
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 160°C/W
UNDERSIDE METAL CONNECTED TO VEE
(PCB CONNECTION OPTIONAL)
LT1996CDD
LT1996IDD
LT1996ACDD
LT1996AIDD
DD PART MARKING*
ORDER PART
NUMBER
TOP VIEW
P9
P27
P81
VEE
REF
1
2
3
4
5
10
9
8
7
6
LT1996CMS
LT1996IMS
LT1996ACMS
LT1996AIMS
M9
M27
M81
VCC
OUT
MS PACKAGE
10-LEAD PLASTIC MSOP
TJMAX = 150°C, θJA = 230°C/W
MS PART MARKING*
LBPC
LTBPB
*Temperature and electrical grades are identified by a label on the shipping container. Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Difference amplifier configuration, VS = 5V, 0V or ±15V;
VCM = VREF = half supply, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
∆G
Gain Error
VS = ±15V, VOUT = ±10V; RL = 10k
G = 81; LT1996AMS
G = 27; LT1996AMS
G = 9; LT1996AMS
MIN
TYP
MAX
UNITS
●
●
●
±0.02
±0.03
±0.03
±0.05
±0.06
±0.07
%
%
%
G = 81; LT1996ADD
G = 27; LT1996ADD
G = 9; LT1996ADD
●
●
●
±0.02
±0.02
±0.03
±0.05
±0.07
±0.08
%
%
%
G = 81; LT1996
G = 27; LT1996
G = 9; LT1996
●
●
●
±0.04
±0.04
±0.04
±0.12
±0.12
±0.12
%
%
%
GNL
Gain Nonlinearity
VS = ±15V; VOUT = ±10V; RL = 10k; G = 9
●
1
10
ppm
∆G/∆T
Gain Drift vs Temperature (Note 6)
VS = ±15V; VOUT = ±10V; RL = 10k
●
0.3
3
ppm/°C
CMRR
Common Mode Rejection Ratio,
Referred to Inputs (RTI)
VS = ±15V; G = 9; VCM = ±15.3V
LT1996AMS
LT1996ADD
LT1996
●
●
●
80
80
70
100
100
100
dB
dB
dB
VS = ±15V; G = 27; VCM = –14.5V to 14.3V
LT1996AMS
LT1996ADD
LT1996
●
●
●
95
90
75
105
105
105
dB
dB
dB
1996f
2
LT1996
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Difference amplifier configuration, VS = 5V, 0V or ±15V;
VCM = VREF = half supply, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
CMRR
Common Mode Rejection Ratio (RTI)
VS = ±15V; G = 81; VCM = –14.1V to 13.9V
LT1996AMS
LT1996ADD
LT1996
VCM
VOS
Input Voltage Range (Note 7)
Op Amp Offset Voltage (Note 8)
MIN
TYP
●
●
●
105
100
85
120
120
120
P9/M9 Inputs
VS = ±15V; VREF = 0V
VS = 5V, 0V; VREF = 2.5V
VS = 3V, 0V; VREF = 1.25V
●
●
●
–15.5
0.84
0.98
15.3
3.94
1.86
V
V
V
P9/M9 Inputs, P81/M81 Connected to REF
VS = ±15V; VREF = 0V
VS = 5V, 0V; VREF = 2.5V
VS = 3V, 0V; VREF = 1.25V
●
●
●
–60
–12.6
–1.25
60
15.6
6.8
V
V
V
P27/M27 Inputs
VS = ±15V; VREF = 0V
VS = 5V, 0V; VREF = 2.5V
VS = 3V, 0V; VREF = 1.25V
●
●
●
–14.5
0.95
1
14.3
3.84
1.82
V
V
V
P81/M81 Inputs
VS = ±15V; VREF = 0V
VS = 5V, 0V; VREF = 2.5V
VS = 3V, 0V; VREF = 1.25V
●
●
●
–14.1
0.99
1
13.9
3.81
1.8
V
V
V
15
50
135
µV
µV
15
80
160
µV
µV
25
100
200
µV
µV
25
150
250
µV
µV
0.3
1
2.5
5
7.5
nA
nA
50
500
750
pA
pA
50
1000
1500
pA
pA
LT1996AMS, VS = 5V, 0V
●
LT1996AMS, VS = ±15V
●
LT1996MS
●
LT1996DD
●
∆VOS/∆T
Op Amp Offset Voltage Drift (Note 6)
IB
Op Amp Input Bias Current
●
●
IOS
Op Amp Input Offset Current
LT1996A
●
LT1996
●
MAX
UNITS
dB
dB
dB
µV/°C
Op Amp Input Noise Voltage
0.01Hz to 1Hz
0.01Hz to 1Hz
0.1Hz to 10Hz
0.1Hz to 10Hz
0.35
0.07
0.25
0.05
µVP-P
µVRMS
µVP-P
µVRMS
en
Input Noise Voltage Density
(Includes Resistor Noise)
G = 9; f = 1kHz
G = 117; f = 1kHz
46
18
nV/√Hz
nV/√Hz
RIN
Input Impedance (Note 10)
P9 (M9 = Ground)
P27 (M27 = Ground)
P81 (M81 = Ground)
●
●
●
350
326.9
319.2
500
467
456
650
607.1
592.8
kΩ
kΩ
kΩ
M9 (P9 = Ground)
M27 (P27 = Ground)
M81 (P81 = Ground)
●
●
●
35
11.69
3.85
50
16.7
5.5
65
21.71
7.15
kΩ
kΩ
kΩ
1996f
3
LT1996
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Difference amplifier configuration, VS = 5V, 0V or ±15V;
VCM = VREF = half supply, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
TYP
MAX
UNITS
∆R
Resistor Matching (Note 9)
G = 81; LT1996AMS
G = 27; LT1996AMS
G = 9; LT1996AMS
●
●
●
MIN
±0.02
±0.03
±0.03
±0.05
±0.06
±0.07
%
%
%
G = 81; LT1996ADD
G = 27; LT1996ADD
G = 9; LT1996ADD
●
●
●
±0.02
±0.02
±0.03
±0.05
±0.07
±0.08
%
%
%
G = 81; LT1996
G = 27; LT1996
G = 9; LT1996
●
●
●
±0.04
±0.04
±0.04
±0.12
±0.12
±0.12
%
%
%
0.3
–30
3
∆R/∆T
Resistor Temperature Coefficient (Note 6)
Resistor Matching
Absolute Value
●
●
PSRR
Power Supply Rejection Ratio
VS = ±1.35V to ±18V (Note 8)
●
●
Minimum Supply Voltage
VOUT
ISC
Output Voltage Swing (to Either Rail)
Output Short-Circuit Current (Sourcing)
Output Short-Circuit Current (Sinking)
BW
–3dB Bandwidth
105
135
ppm/°C
ppm/°C
dB
2.4
2.7
V
40
55
65
110
mV
mV
mV
150
225
275
300
mV
mV
mV
No Load
VS = 5V, 0V
VS = 5V, 0V
VS = ±15V
●
●
1mA Load
VS = 5V, 0V
VS = 5V, 0V
VS = ±15V
●
●
Drive Output Positive;
Short Output to Ground
8
4
12
●
mA
mA
Drive Output Negative;
Short Output to VS or Midsupply
8
4
21
●
mA
mA
G=9
G = 27
G = 81
38
17
7
kHz
kHz
kHz
GBWP
Op Amp Gain Bandwidth Product
f = 10kHz
560
kHz
tr, tf
Rise Time, Fall Time
G = 9; 0.1V Step; 10% to 90%
G = 81; 0.1V Step; 10% to 90%
8
40
µs
µs
tS
Settling Time to 0.01%
G = 9; VS = 5V, 0V; 2V Step
G = 9; VS = 5V, 0V; –2V Step
G = 9; VS = ±15V; 10V Step
G = 9; VS = ±15V; –10V Step
85
85
110
110
µs
µs
µs
µs
SR
Slew Rate
VS = 5V, 0V; VOUT = 1V to 4V
VS = ±15V; VOUT = ±10V
0.12
0.12
V/µs
V/µs
IS
Supply Current
VS = 5V, 0V
●
●
0.06
0.08
100
110
150
µA
µA
130
160
210
µA
µA
●
VS = ±15V
●
Note 1: Absolute Maximum Ratings are those beyond which the life of the
device may be impaired.
Note 2: The P27/M27 and P81/M81 inputs are protected by ESD diodes to
the supply rails. If one of these four inputs goes outside the rails, the input
current should be limited to less than 10mA. The P9/M9 inputs can
withstand ±60V if P81/M81 are grounded and VS = ±15V (see Applications
Information section about “High Voltage CM Difference Amplifiers”).
Note 3: A heat sink may be required to keep the junction temperature
below absolute maximum ratings.
1996f
4
LT1996
ELECTRICAL CHARACTERISTICS
Note 4: Both the LT1996C and LT1996I are guaranteed functional over the
–40°C to 85°C temperature range.
Note 5: The LT1996C is guaranteed to meet the specified performance
from 0°C to 70°C and is designed, characterized and expected to meet
specified performance from –40°C to 85°C but is not tested or QA
sampled at these temperatures. The LT1996I is guaranteed to meet
specified performance from –40°C to 85°C.
Note 6: This parameter is not 100% tested.
Note 7: Input voltage range is guaranteed by the CMRR test at VS = ±15V.
For the other voltages, this parameter is guaranteed by design and through
correlation with the ±15V test. See the Applications Information section to
determine the valid input voltage range under various operating
conditions.
Note 8: Offset voltage, offset voltage drift and PSRR are defined as
referred to the internal op amp. You can calculate output offset as follows.
In the case of balanced source resistance, VOS, OUT = VOS • Noise Gain +
IOS • 450k + IB • 450k • (1 – RP/RN) where RP and RN are the total
resistance at the op amp positive and negative terminal respectively.
Note 9: Resistors connected to the minus inputs. Resistor matching is not
tested directly, but is guaranteed by the gain error test.
Note 10: Input impedance is tested by a combination of direct
measurements and correlation to the CMRR and gain error tests.
U W
TYPICAL PERFOR A CE CHARACTERISTICS
(Difference Amplifier Configuration)
Output Voltage Swing vs
Temperature
Output Voltage Swing vs Load
Current (Output Low)
200
OUTPUT VOLTAGE SWING (mV)
175
TA = 85°C
150
TA = 25°C
125
TA = –40°C
100
75
50
25
0
2
4
60
0
OUTPUT SHORT-CIRCUIT CURRENT (mA)
OUTPUT VOLTAGE SWING (mV)
TA = 85°C
–400
TA = 25°C
–500
–600
–700
–800
–900
–1000
0
1
2
3 4 5 6 7
LOAD CURRENT (mA)
8
TA = 25°C
600
TA = –40°C
400
25
50
75
100
VEE
125
9
10
1996 G04
0
1
2
3 4 5 6 7
LOAD CURRENT (mA)
1996 G02
25
TA = –40°C
–300
800
150
VS = 5V, 0V
20
15
SOURCING
5
0
–50 –25
9
10
Input Offset Voltage vs
Difference Gain
SINKING
10
8
1996 G03
Output Short-Circuit Current vs
Temperature
–100
–200
1000
TEMPERATURE (°C)
Output Voltage Swing vs Load
Current (Output High)
VS = 5V, 0V
TA = 85°C
200
1996 G01
VCC
VS = 5V, 0V
1200
OUTPUT LOW
(LEFT AXIS)
VEE
–50 –25
6 8 10 12 14 16 18 20
SUPPLY VOLTAGE (±V)
–40
–60
20
0
–20
OUTPUT HIGH
(RIGHT AXIS)
40
1400
INPUT OFFSET VOLTAGE (µV)
SUPPLY CURRENT (µA)
VCC
VS = 5V, 0V
NO LOAD
OUTPUT VOLTAGE (mV)
Supply Current vs Supply Voltage
VS = 5V, 0V
REPRESENTATIVE PARTS
100
50
0
–50
–100
–150
50
25
0
75
TEMPERATURE (°C)
100
125
1996 G05
9 18 27 36 45 54 63 72 81 90 99 108 117
GAIN (V/V)
1996 G06
1996f
5
LT1996
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Output Offset Voltage vs
Difference Gain
Gain Error vs Load Current
0.04
VS = 5V, 0V
REPRESENTATIVE PARTS
7.5
5.0
0
–2.5
0
–0.01
–0.02
–7.5
–0.03
9 18 27 36 45 54 63 72 81 90 99 108 117
GAIN (V/V)
REPRESENTATIVE UNITS
–0.04
1
0
2
3
LOAD CURRENT (mA)
4
CMRR (dB)
–3dB BANDWIDTH (kHz)
25
20
15
10
5
9 18 27 36 45 54 63 72 81 90 99 108 117
GAIN (V/V)
130
120
110
100
90
80
70
60
50
40
30
20
10
0
VS = 5V, 0V
TA = 25°C
100
GAIN = 27
90
GAIN = 9
80
0.01
10k
100k
1996 G13
GAIN = 81
GAIN = 9
GAIN = 27
10
0
10
100
1k
10k
FREQUENCY (Hz)
1M
100k
10
100
1k
10k
FREQUENCY (Hz)
Gain Error vs Temperature
GAIN = 9
VS = ±15V
GAIN = 9
VS = ±15V
0.025
60
0
–50 –25
100k
1996 G12
0.030
20
100
1k
FREQUENCY (Hz)
50
20
GAIN ERROR (%)
CMRR (dB)
0.1
10
60
30
40
GAIN = 9
1
70
40
80
GAIN = 27
VS = 5V, 0V
TA = 25°C
110
100
100
OUTPUT IMPEDANCE (Ω)
120
VS = 5V, 0V
TA = 25°C
GAIN = 81
125
1996 G09
CMRR vs Temperature
120
GAIN = 81
100
1996 G11
Output Impedance vs Frequency
10
50
25
75
0
TEMPERATURE (°C)
PSRR vs Frequency
1996 G10
1
0.10
0
–50 –25
5
PSRR (dB)
VS = 5V, 0V
TA = 25°C
1000
SR+ (RISING EDGE)
CMRR vs Frequency
30
0
SR– (FALLING EDGE)
0.15
1996 G08
Bandwidth vs Gain
35
0.20
0.05
1996 G07
40
GAIN = 9
VS = ±15V
VOUT = ±10V
0.25
0.01
–5.0
–10.0
0.30
SLEW RATE (V/µs)
0.02
2.5
Slew Rate vs Temperature
GAIN = 81
VS = ±15V
VOUT = ±10V
TA = 25°C
0.03
GAIN ERROR (%)
OUTPUT OFFSET VOLTAGE (mV)
10.0
(Difference Amplifier Configuration)
0.020
0.015
0.010
0.005
REPRESENTATIVE UNITS
50
25
75
0
TEMPERATURE (°C)
100
125
1996 G14
0
–50 –25
REPRESENTATIVE UNITS
50
25
75
0
TEMPERATURE (°C)
100
125
1996 G15
1996f
6
LT1996
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TYPICAL PERFOR A CE CHARACTERISTICS
Gain and Phase vs Frequency
40
VS = 5V, 0V
TA = 25°C
VS = 5V, 0V
TA = 25°C –20
GAIN = 9
–40
PHASE
(RIGHT AXIS)
40 GAIN = 81
30
30 GAIN = 27
20
0.01Hz to 1Hz Voltage Noise
0
GAIN (dB)
20 GAIN = 9
–80
GAIN
(LEFT AXIS)
–100
10
–120
PHASE (deg)
GAIN (dB)
–60
–140
0
10
–160
–180
0
0.5
1
10
100
FREQUENCY (kHz)
500
–10
0.1
1
10
FREQUENCY (kHz)
1996 G16
–200
400
U
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PI FU CTIO S
10 20 30 40 50 60 70 80 90 100
TIME (s)
1996 G21
Small Signal Transient Response,
Gain = 81
Small Signal Transient Response,
Gain = 27
50mV/DIV
50mV/DIV
10µs/DIV
0
1996 G17
Small Signal Transient Response,
Gain = 9
50mV/DIV
100
VS = ±15V
TA = 25°C
MEASURED IN G =117
REFERRED TO OP AMP INPUTS
OP AMP VOLTAGE NOISE (100nV/DIV)
Gain vs Frequency
50
(Difference Amplifier Configuration)
1996 G18
20µs/DIV
1996 G19
50µs/DIV
1996 G20
(Difference Amplifier Configuration)
P9 (Pin 1): Noninverting Gain-of-9 input. Connects a 50k
internal resistor to the op amp’s noninverting input.
P27 (Pin 2): Noninverting Gain-of-27 input. Connects a
(50k/3) internal resistor to the op amp’s noninverting input.
P81 (Pin 3): Noninverting Gain-of-81 input. Connects a
(50k/9) internal resistor to the op amp’s noninverting input.
VEE (Pin 4): Negative Power Supply. Can be either ground
(in single supply applications), or a negative voltage (in
split supply applications).
REF (Pin 5): Reference Input. Sets the output level when
difference between inputs is zero. Connects a 450k internal
resistor to the op amp’s noninverting input.
OUT (Pin 6): Output. VOUT = VREF + 9 • (VP1 – VM1) + 27 •
(VP3 – VM3) + 81 • (VP9 – VM9).
VCC (Pin 7): Positive Power Supply. Can be anything from
2.7V to 36V above the VEE voltage.
M81 (Pin 8): Inverting Gain-of-81 input. Connects a
(50k/9) internal resistor to the op amp’s inverting input.
M27 (Pin 9): Inverting Gain-of-27 input. Connects a
(50k/3) internal resistor to the op amp’s inverting input.
M9 (Pin 10): Inverting Gain-of-9 input. Connects a 50k
internal resistor to the op amp’s inverting input.
1996f
7
LT1996
W
BLOCK DIAGRA
10
M9
9
M27
8
7
M81
450k/81
6
VCC
OUT
450k
4pF
450k/27
450k/9
450k/9
–
OUT
+
LT1996
450k/27
450k/81
P9
2
P27
3
P81
450k
4
VEE
5
REF
1996 BD
U
1
4pF
W
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APPLICATIO S I FOR ATIO
Introduction
The LT1996 may be the last op amp you ever have to stock.
Because it provides you with several precision matched
resistors, you can easily configure it into several different
classical gain circuits without adding external components. The several pages of simple circuits in this data
sheet demonstrate just how easy the LT1996 is to use. It
can be configured into difference amplifiers, as well as into
inverting and noninverting single ended amplifiers. The
fact that the resistors and op amp are provided together in
such a small package will often save you board space and
reduce complexity for easy probing.
The Op Amp
The op amp internal to the LT1996 is a precision device
with 15µV typical offset voltage and 3nA input bias current. The input offset current is extremely low, so matching the source resistance seen by the op amp inputs will
provide for the best output accuracy. The op amp inputs
are not rail-to-rail, but extend to within 1.2V of VCC and 1V
of VEE. For many configurations though, the chip inputs
will function rail-to-rail because of effective attenuation to
the +input. The output is truly rail-to-rail, getting to within
40mV of the supply rails. The gain bandwidth product of
the op amp is about 560kHz. In noise gains of 2 or more,
it is stable into capacitive loads up to 500pF. In noise gains
below 2, it is stable into capacitive loads up to 100pF.
The Resistors
The resistors internal to the LT1996 are very well matched
SiChrome based elements protected with barrier metal.
Although their absolute tolerance is fairly poor (±30%),
their matching is to within 0.05%. This allows the chip to
achieve a CMRR of 80dB, and gain errors within 0.05%.
The resistor values are (450k/9), (450k/27), (450k/81)
and 450k, connected to each of the inputs. The resistors
have power limitations of 1watt for the 450k and (450k/81)
resistors, 0.3watt for the (450k/27) resistors and 0.5watt
for the (450k/9) resistors; however, in practice, power
dissipation will be limited well below these values by the
1996f
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maximum voltage allowed on the input and REF pins. The
50k resistors connected to the M9 and P9 inputs are
isolated from the substrate, and can therefore be taken
beyond the supply voltages. The naming of the pins “P9,”
“P27,” “P81,” etc., is based on their admittances relative
to the feedback and REF admittances. Because it has 9
times the admittance, the voltage applied to the P9 input
has 9 times the effect of the voltage applied to the REF
input.
Bandwidth
The bandwidth of the LT1996 will depend on the gain you
select (or more accurately the noise gain resulting from
the gain you select). In the lowest configurable gain of 1,
the –3dB bandwidth is limited to 450kHz, with peaking of
about 2dB at 280kHz. In the highest configurable gains,
bandwidth is limited to 5kHz.
Input Noise
The LT1996 input noise is comprised of the Johnson noise
of the internal resistors (√4kTR), and the input voltage
noise of the op amp. Paralleling all four resistors to the
+input gives a 3.8kΩ resistance, for 8nV/√Hz of voltage
noise. The equivalent network on the –input gives another
8nV/√Hz, and the op amp 14nV/√Hz. Taking their RMS
sum gives a total 18nV/√Hz input referred noise floor.
Output noise depends on configuration and noise gain.
Input Resistance
The LT1996 input resistances vary with configuration, but
once configured are apparent on inspection. Note that
resistors connected to the op amp’s –input are looking
into a virtual ground, so they simply parallel. Any feedback
resistance around the op amp does not contribute to input
resistance. Resistors connected to the op amp’s +input
are looking into a high impedance, so they add as parallel
or series depending on how they are connected, and
whether or not some of them are grounded. The op amp
+input itself presents a very high GΩ impedance. In the
classical noninverting op amp configuration, the LT1996
presents the high input impedance of the op amp, as is
usual for the noninverting case.
Common Mode Input Voltage Range
The LT1996 valid common mode input range is limited by
three factors:
1. Maximum allowed voltage on the pins
2. The input voltage range of the internal op amp
3. Valid output voltage
The maximum voltage allowed on the P27, M27, P81 and
M81 inputs includes the positive and negative supply plus
a diode drop. These pins should not be driven more than
a diode drop outside of the supply rails. This is because
they are connected through diodes to internal manufacturing post-package trim circuitry, and through a substrate
diode to VEE. If more than 10mA is allowed to flow through
these pins, there is a risk that the LT1996 will be detrimmed
or damaged. The P9 and M9 inputs do not have clamp
diodes or substrate diodes or trim circuitry and can be
taken well outside the supply rails. The maximum allowed
voltage on the P9 and M9 pins is ±60V.
The input voltage range of the internal op amp extends to
within 1.2V of VCC and 1V of VEE. The voltage at which the
op amp inputs common mode is determined by the
voltage at the op amp’s +input, and this is determined by
the voltages on pins P9, P27, P81 and REF. (See “Calculating Input Voltage Range” section.) This is true provided
that the op amp is functioning and feedback is maintaining
the inputs at the same voltage, which brings us to the third
requirement.
For valid circuit function, the op amp output must not be
clipped. The output will clip if the input signals are attempting to force it to within 40mV of its supply voltages. This
usually happens due to too large a signal level, but it can
also occur with zero input differential and must therefore
be included as an example of a common mode problem.
1996f
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Consider Figure 1. This shows the LT1996 configured as
a gain of 117 difference amplifier on a single supply with
RF
VCC
RG
–
5V
7
8
450k/81
VEXT
VINT
450k
+
RG
VEE
4pF
9
450k/27
10
450k/9
–
VDM
0V+
VCM
2.5V
VREF
RF
1
450k/9
2
450k/27
3
450k/81
Figure 2. Calculating CM Input Voltage Range
–
6
VOUT = 117 • VDM
+
4pF
450k
REF 5
LT1996
4
1996 F02
1996 F01
Figure 1. Difference Amplifier Cannot Produce 0V on a Single
Supply. Provide a Negative Supply, or Raise Pin 5, or Provide
400µV of VDM
These two voltages represent the high and low extremes
of the common mode input range, if the other limits have
not already been exceeded (1 and 3, above). In most cases,
the inverting inputs M9 through M81 can be taken further
than these two extremes because doing this does not
move the op amp input common mode. To calculate the
limit on this additional range, see Figure 3. Note that, with
RF
the output REF connected to ground. This is a great circuit,
but it does not support VDM = 0V at any common mode
because the output clips into ground while trying to
produce 0VOUT. It can be fixed simply by declaring the
valid input differential range not to extend below +0.4mV,
or by elevating the REF pin above 40mV, or by providing
a negative supply.
VINT = VEXT • (RF/(RF + RG)) + VREF • (RG/(RF + RG))
Or, solving for VEXT:
VEXT = VINT • (1 + RG/RF) – VREF • RG/RF
But valid VINT voltages are limited to VCC – 1.2V and VEE +
1V, so:
MAX VEXT = (VCC – 1.2) • (1 + RG/RF) – VREF • RG/RF
and:
MIN VEXT = (VEE + 1) • (1 + RG/RF) – VREF • RG/RF
–
VMORE
VEXT
MAX OR MIN
VINT
RG
+
VEE
VREF
RF
1996 F03
Figure 3. Calculating Additional Voltage Range of
Inverting Inputs
Calculating Input Voltage Range
Figure 2 shows the LT1996 in the generalized case of a
difference amplifier, with the inputs shorted for the common mode calculation. The values of RF and RG are
dictated by how the P inputs and REF pin are connected.
By superposition we can write:
VCC
RG
VMORE = 0, the op amp output is at VREF. From the max
VEXT (the high cm limit), as VMORE goes positive, the op
amp output will go more negative from VREF by the amount
VMORE • RF/RG, so:
VOUT = VREF – VMORE • RF/RG
Or:
VMORE = (VREF – VOUT) • RG/RF
The most negative that VOUT can go is VEE + 0.04V, so:
Max VMORE = (VREF – VEE – 0.04V) • RG/RF
(should be positive)
The situation where this function is negative, and therefore
problematic, when VREF = 0 and VEE = 0, has already been
dealt with in Figure 1. The strength of the equation is
demonstrated in that it provides the three solutions
1996f
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suggested in Figure 1: raise VREF, lower VEE, or provide
some negative VMORE.
The Classical Noninverting Amplifier: High Input Z
representation of the circuit on the top. The LT1996 is
shown on the bottom configured in a precision gain of 9.1.
One of the benefits of the noninverting op amp configuration is that the input impedance is extremely high. The
LT1996 maintains this benefit. Given the finite number of
available feedback resistors in the LT1996, the number of
gain configurations is also finite. The complete list of such
Hi-Z input noninverting gain configurations is shown in
Table 1. Many of these are also represented in Figure 5 in
schematic form. Note that the P-side resistor inputs have
been connected so as to match the source impedance
seen by the internal op amp inputs. Note also that gain and
noise gain are identical, for optimal precision.
Perhaps the most common op amp configuration is the
noninverting amplifier. Figure 4 shows the textbook
Table 1. Configuring the M Pins for Simple Noninverting Gains.
The P Inputs are driven as shown in the examples on the next
page
Likewise, from the lower common mode extreme, making
the negative input more negative will raise the output
voltage, limited by VCC – 0.04V.
MIN VMORE = (VREF – VCC + 0.04V) • RG/RF
(should be negative)
Again, the additional input range calculated here is only
available provided the other remaining constraint is not
violated, the maximum voltage allowed on the pin.
RF
RG
–
VOUT
VIN
+
VOUT = GAIN • VIN
GAIN = 1 + RF/RG
CLASSICAL NONINVERTING OP AMP CONFIGURATION.
YOU PROVIDE THE RESISTORS.
8
450k/81
9
450k/27
10
450k/9
450k
4pF
–
6
1
450k/9
2
450k/27
3
450k/81
VOUT
+
4pF
450k
LT1996
5
VIN
Gain
M81
M81, M27, M9 Connection
M27
M9
1
Output
Output
Output
1.08
Output
Output
Grounded
1.11
Output
Float
Grounded
1.30
Output
Grounded
Output
1.32
Float
Output
Grounded
1.33
Output
Grounded
Float
1.44
Output
Grounded
Grounded
3.19
Grounded
Output
Output
3.7
Float
Grounded
Output
3.89
Grounded
Output
Float
4.21
Grounded
Output
Grounded
9.1
Grounded
Float
Output
10
Float
Float
Grounded
11.8
Grounded
Grounded
Output
28
Float
Grounded
Float
37
Float
Grounded
Grounded
82
Grounded
Float
Float
91
Grounded
Float
Grounded
109
Grounded
Grounded
Float
118
Grounded
Grounded
Grounded
CLASSICAL NONINVERTING OP AMP CONFIGURATION
IMPLEMENTED WITH LT1991. RF = 45k, RG = 5.6k, GAIN = 9.1.
GAIN IS ACHIEVED BY GROUNDING, FLOATING OR FEEDING BACK
THE AVAILABLE RESISTORS TO ARRIVE AT DESIRED RF AND RG.
WE PROVIDE YOU WITH <0.1% RESISTORS.
1996 F04
Figure 4. The LT1996 as a Classical Noninverting Op Amp
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VS+
8
M81
9
M27
10
M9
VIN
VS+
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VIN
VS–
6
VOUT
VIN
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VS+
VS–
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
6
GAIN = 37
VS+
VIN
GAIN = 9.1
VS+
8
M81
9
M27
10
M9
7
VCC
6
VOUT
VIN
GAIN = 28
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VS–
VIN
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VS–
VIN
VOUT
GAIN = 3.893
VS+
8
M81
9
M27
10
M9
6
VS–
GAIN = 10
VS+
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VS–
GAIN = 1
8
M81
9
M27
10
M9
VS+
8
M81
9
M27
10
M9
VS+
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VIN
VS–
8
M81
9
M27
10
M9
6
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VS–
6
VOUT
VS–
VIN
GAIN = 11.8
GAIN = 82
GAIN = 91
VS+
8
M81
9
M27
10
M9
VIN
VS+
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VS–
6
VOUT
VS–
VIN
GAIN = 109
GAIN = 118
1996 F05
Figure 5. Some Implementations of Classical Noninverting
Gains Using the LT1996. High Input Z Is Maintained
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Attenuation Using the P Input Resistors
Attenuation happens as a matter of fact in difference
amplifier configurations, but it is also used for reducing
peak signal level or improving input common mode range
even in single ended systems. When signal conditioning
indicates a need for attenuation, the LT1996 resistors are
ready at hand. The four precision resistors can provide
several attenuation levels, and these are tabulated in
Table 2 as a design reference.
VIN
VIN
OKAY UP
TO ±60V
RA
VINT
RG
VINT = A • VIN
A = RG/(RA + RG)
1
450k/9
2
450k/27
3
450k/81
VINT
+
450k
LT1996
5
CLASSICAL ATTENUATOR
LT1991 ATTENUATING TO THE +INPUT BY
DRIVING AND GROUNDING AND FLOATING
INPUTS RA = 50k, RG = 50k/9, SO A = 0.1.
1996 F06
Figure 6. LT1996 Provides for Easy Attenuation to the Op Amp’s
+Input. The P9 Input Can Be Taken Well Outside of the Supplies
Because the attenuations and the noninverting gains are
set independently, they can be combined. This provides
high gain resolution, about 700 unique gains between
0.0085 and 118, as plotted in Figure 7. This is too large a
number to tabulate, but the designer can calculate achievable gain by taking the vector product of the gains and
attenuations in Tables 1 and 2, and seeking the best match.
Average gain resolution is 1.5%, with worst case steps of
about 50% as seen in Figure 7.
1000
100
GAIN
10
1
0.1
0.01
0.001
0
100
200
300 400
COUNT
500
600
700
1996 F07
Figure 7. Over 600 Unique Gain Settings Achievable with the
LT1996 by Combining Attenuation with Noninverting Gain
Table 2. Configuring the P Pins for Various Attenuations. Those
Shown in Bold Are Functional Even When the Input Drive
Exceeds the Supplies
A
0.0085
0.0092
0.0110
0.0122
0.0270
0.0357
0.0763
0.0769
0.0847
0.0989
0.1
0.110
0.229
0.231
0.237
0.243
0.248
0.25
0.25
0.257
0.270
0.305
0.308
0.314
0.686
0.692
0.695
0.730
0.743
0.75
0.752
0.757
0.763
0.769
0.771
0.890
0.9
0.901
0.915
0.923
0.924
0.964
0.973
0.988
0.989
0.991
0.992
P81
Grounded
Grounded
Grounded
Grounded
Float
Float
Grounded
Grounded
Grounded
Grounded
Grounded
Grounded
Grounded
Grounded
Grounded
Float
Grounded
Float
Grounded
Grounded
Float
Grounded
Grounded
Grounded
Driven
Driven
Driven
Float
Driven
Float
Driven
Float
Driven
Driven
Driven
Driven
Float
Driven
Driven
Driven
Driven
Float
Float
Driven
Driven
Driven
Driven
P81, P27, P9, REF Connection
P27
P9
REF
Grounded
Grounded
Driven
Grounded
Float
Driven
Float
Grounded
Driven
Float
Float
Driven
Grounded
Grounded
Driven
Grounded
Float
Driven
Grounded
Driven
Grounded
Grounded
Driven
Float
Grounded
Driven
Driven
Float
Driven
Grounded
Float
Driven
Float
Float
Driven
Driven
Driven
Grounded
Grounded
Driven
Grounded
Float
Driven
Grounded
Driven
Grounded
Driven
Grounded
Driven
Float
Grounded
Grounded
Driven
Float
Driven
Float
Float
Driven
Float
Driven
Grounded
Driven
Driven
Driven
Driven
Grounded
Driven
Driven
Float
Driven
Driven
Driven
Grounded
Grounded
Grounded
Grounded
Grounded
Float
Grounded
Grounded
Driven
Driven
Grounded
Grounded
Grounded
Float
Grounded
Driven
Grounded
Float
Grounded
Float
Driven
Driven
Grounded
Driven
Grounded
Driven
Grounded
Grounded
Driven
Float
Grounded
Driven
Driven
Float
Grounded
Grounded
Float
Driven
Grounded
Float
Grounded
Driven
Driven
Grounded
Grounded
Driven
Grounded
Float
Driven
Grounded
Driven
Driven
Float
Grounded
Driven
Driven
Grounded
Float
Float
Grounded
Float
Driven
Grounded
Driven
Float
Grounded
Driven
Driven
Grounded
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Inverting Configuration
Table 3. Configuring the M Pins for Simple Inverting Gains
The inverting amplifier, shown in Figure 8, is another
classical op amp configuration. The circuit is actually
identical to the noninverting amplifier of Figure 4, except
that VIN and GND have been swapped. The list of available
gains is shown in Table 3, and some of the circuits are
shown in Figure 9. Noise gain is 1+|Gain|, as is the usual
case for inverting amplifiers. Again, for the best DC performance, match the source impedance seen by the op amp
inputs.
RF
RG
VIN
–
VOUT
+
VOUT = GAIN • VIN
GAIN = – RF/RG
CLASSICAL INVERTING OP AMP CONFIGURATION.
YOU PROVIDE THE RESISTORS.
VIN
(DRIVE)
8
450k/81
9
450k/27
10
450k/9
1
450k/9
2
450k/27
3
450k/81
450k
Gain
M81
M81, M27, M9 Connection
M27
M9
–0.083
Output
Output
Drive
–0.110
Output
Float
Drive
–0.297
Output
Drive
Output
–0.321
Float
Output
Drive
–0.329
Output
Drive
Float
–0.439
Output
Drive
Drive
–2.19
Drive
Output
Output
–2.7
Float
Drive
Output
–2.89
Drive
Output
Float
–3.21
Drive
Output
Drive
–8.1
Drive
Float
Output
–9
Float
Float
Drive
–10.8
Drive
Drive
Output
–27
Float
Drive
Float
–36
Float
Drive
Drive
–81
Drive
Float
Float
–90
Drive
Float
Drive
–108
Drive
Drive
Float
–117
Drive
Drive
Drive
4pF
–
6
VOUT
+
4pF
450k
LT1996
5
CLASSICAL INVERTING OP AMP CONFIGURATION IMPLEMENTED
WITH LT1991. RF = 45k, RG = 5.55k, GAIN = –8.1.
GAIN IS ACHIEVED BY GROUNDING, FLOATING OR FEEDING BACK
THE AVAILABLE RESISTORS TO ARRIVE AT DESIRED RF AND RG.
WE PROVIDE YOU WITH <0.1% RESISTORS.
1996 F08
Figure 8. The LT1996 as a Classical Inverting Op Amp. Note the
Circuit Is Identical to the Noninverting Amplifier, Except that VIN
and Ground Have Been Swapped
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VS +
8
M81
9
M27
10
M9
VIN
VS +
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
8
M81
9
M27
10
M9
VIN
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VIN
VS+
VIN
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
6
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VS–
VS–
VS–
GAIN = –27
GAIN = –36
GAIN = –8.1
VS +
8
M81
9
M27
10
M9
VS +
8
M81
9
M27
10
M9
VIN
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VIN
7
VCC
6
8
M81
9
M27
10
M9
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VS–
VS–
GAIN = –10.8
GAIN = –81
GAIN = –90
VS +
VIN
6
VOUT
VS +
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
7
VCC
VS–
8
M81
9
M27
10
M9
6
VS+
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VOUT
GAIN = –2.89
VS +
8
M81
9
M27
10
M9
6
VS–
GAIN = –9
VS +
VIN
6
VS–
GAIN = –0.321
VIN
VIN
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VS–
8
M81
9
M27
10
M9
VS+
8
M81
9
M27
10
M9
6
VIN
VOUT
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VS–
VS–
GAIN = –108
GAIN = –117
6
VOUT
1996 F09
Figure 9. It Is Simple to Get Precision Inverting Gains with the LT1996.
Input Impedance Varies from 3.8kΩ (Gain = –117) to 50kΩ (Gain = –9)
1996f
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Difference Amplifiers
RF
The resistors in the LT1996 allow it to easily make difference amplifiers also. Figure 10 shows the basic 4-resistor
difference amplifier and the LT1996. A difference gain of
27 is shown, but notice the effect of the additional dashed
connections. By connecting the 50k resistors in parallel,
the gain is reduced by a factor of 10. Of course, with so
many resistors, there are many possible gains. Table 4
shows the difference gains and how they are achieved.
Note that, as for inverting amplifiers, the noise gain is 1
more than the signal gain.
VIN–
VIN+
VIN+
VIN–
Output
GND (REF)
0.083
P9
M9
M27, M81
P27, P81
0.110
P9
M9
M81
P81
0.297
P27
M27
M9, M81
P9, P81
0.321
P9
M9
M27
P27
0.329
P27
M27
M81
P81
0.439
P9, P27
M9, M27
M81
P81
2.189
P81
M81
M9, M27
P9, P27
2.700
P27
M27
M9
P9
2.893
P81
M81
M27
P27
3.214
P9, P81
M9, M81
M27
P27
8.1
P81
M81
M9
P9
9
P9
M9
10.8
P27, P81
M27, M81
27
P27
M27
36
P9, P27
M9, M27
81
P81
M81
90
P9, P81
M9, M81
108
P27, P81
M27, M81
117
M9
P9
RG
–
VOUT
+
RF
VOUT = GAIN • (VIN+ – VIN–)
GAIN = RF/RG
CLASSICAL DIFFERENCE AMPLIFIER USING THE LT1991
8
450k/81
9
450k/27
10
450k/9
1
450k/9
2
450k/27
4pF
3
450k/81
450k
Table 4. Connections Giving Difference Gains for the LT1996
Gain
RG
VIN–
PARALLEL
TO CHANGE
R F, R G
VIN+
450k
4pF
–
6
VOUT
+
5
LT1996
CLASSICAL DIFFERENCE AMPLIFIER IMPLEMENTED
WITH LT1991. RF = 450k, RG = 16.7k, GAIN = 3.
ADDING THE DASHED CONNECTIONS CONNECTS THE
TWO 450k RESISTOR IN PARALLEL, SO RF IS REDUCED
TO 45k. GAIN BECOMES 45k/16.7k = 2.7.
1996 F10
Figure 10. Difference Amplifier Using the LT1996. Gain Is Set
Simply by Connecting the Correct Resistors or Combinations of
Resistors. Gain of 27 Is Shown, with Dashed Lines Modifying It
to Gain of 2.7. Noise Gain Is Optimal
P9, P27, P81 M9, M27, M81
1996f
16
LT1996
U
U
W
U
APPLICATIO S I FOR ATIO
VS +
VIN
8
M81
9
M27
10
M9
–
7
VCC
LT1996
VIN
1
P9
2
P27 VEE
3
P81
4
+
VS +
OUT
REF
5
6
VIN
–
VIN
+
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
VS–
VIN+
7
VCC
VIN–
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VIN+
6
VOUT
VIN+
8
M81
9
M27
10
M9
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VOUT
VS–
VS+
7
VCC
6
VOUT
VIN+
8
M81
9
M27
10
M9
VIN+
VOUT
GAIN = 90
VS +
VS +
VIN–
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VS–
GAIN = 81
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VS–
GAIN = 10.8
VOUT
GAIN = 8.1
VIN–
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VIN+
6
VS–
VS +
7
VCC
VIN–
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
GAIN = 36
8
M81
9
M27
10
M9
VOUT
VS+
VIN–
VS–
VIN–
6
GAIN = 2.89
7
VCC
VS +
VIN+
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VS–
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VOUT
GAIN = 27
8
M81
9
M27
10
M9
VIN+
7
VCC
VS +
8
M81
9
M27
10
M9
VS–
VIN–
VOUT
GAIN = 9
VS +
VIN–
6
8
M81
9
M27
10
M9
VS–
GAIN = 0.321
8
M81
9
M27
10
M9
VS+
VIN–
6
VOUT
VIN+
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VS–
VS–
GAIN = 108
GAIN = 117
6
VOUT
1996 F11
Figure 11. Many Difference Gains Are Achievable Just by Strapping the Pins
1996f
17
LT1996
U
U
W
U
APPLICATIO S I FOR ATIO
VIN–
RF
VIN
–
VIN+
RG
RG
CROSSCOUPLING
–
VOUT
+
VIN+
+
RF
8
450k/81
450k
9
450k/27
10
450k/9
1
450k/9
2
450k/27
4pF
3
450k/81
450k
4pF
–
6
VOUT
+
–
VOUT = GAIN • (VIN – VIN )
GAIN = RF/RG
5
LT1996
CLASSICAL DIFFERENCE AMPLIFIER IMPLEMENTED
WITH LT1991. RF = 450k, RG = 16.7k, GAIN = 27.
GAIN CAN BE ADJUSTED BY "CROSS COUPLING." MAKING THE
DASHED CONNECTIONS REDUCE THE GAIN FROM 3 T0 2.
WHEN CROSS COUPLING, SEE WHAT IS CONNECTED TO THE
VIN+ VOLTAGE. CONNECTING P27 AND M9 GIVES 27 – 9 = 18.
CONNECTIONS TO VIN– ARE SYMMETRIC: M27 AND P9.
CLASSICAL DIFFERENCE AMPLIFIER
1996 F10
Figure 12. Another Method of Selecting Difference Gain Is “Cross-Coupling.”
The Additional Method Means the LT1996 Provides Extra Integer Gains
Difference Amplifier: Additional Integer Gains Using
Cross-Coupling
VS+
VIN–
Figure 12 shows the basic difference amplifier as well as
the LT1996 in a difference gain of 27. But notice the effect
of the additional dashed connections. This is referred to as
“cross-coupling” and has the effect of reducing the differential gain from 27 to 18. Using this method, additional
integer gains are achievable, as shown in Table 5 below.
Note that the equations can be written by inspection from
the VIN+ connections, and that the VIN– connections are
simply the opposite (swap P for M and M for P). The
method is the same as for the LT1991, except that the
LT1996 applies a multiplier of 9. Noise gain, bandwidth,
and input impedance specifications for the various cases
are also tabulated, as these are not obvious. Schematics
are provided in Figure 13.
VIN+
Gain
VIN+
VIN–
18
P27, M9
M27, P9
27 – 9
39
45 P81, M27, M9 M81, P27, P9 81 – 27 – 9 117
54
P81, M27
12
16
6
108
5
16
6
5
16
5
72
90
6
45
6
99 P81, P27, M9 M81, M27, P9 81 + 27 – 9 117
5
45
4
M81, P9
81 – 27
5
46
63 P81, P9, M27 M81, M9, P27 81 + 9 – 27 117
P81, M9
M81, P27
14
81 – 9
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VOUT
VIN+
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VIN+
8
M81
9
M27
10
M9
VS+
VIN–
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VOUT
VIN+
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VS–
GAIN = 63
VS+
VIN+
8
M81
9
M27
10
M9
6
VS–
GAIN = 45
VIN–
VOUT
GAIN = 54
VS+
VIN–
6
VS–
GAIN = 18
–
Gain Noise –3dB BW RIN
RIN
Equation Gain
kHz Typ kΩ Typ kΩ
VS+
VIN–
7
VS–
Table 5. Connections Using Cross-Coupling. Note That Equations
Can Be Written by Inspection of the VIN+ Column
+
8
M81
9
M27
10
M9
VS+
VIN–
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VS–
GAIN = 72
6
VOUT
VIN+
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VOUT
VS–
GAIN = 99
1996 F13
Figure 13. Integer Gain Difference
Amplifiers Using Cross-Coupling
1996f
18
LT1996
U
W
U
U
APPLICATIO S I FOR ATIO
High Voltage CM Difference Amplifiers
This class of difference amplifier remains to be discussed.
Figure 14 shows the basic circuit on the top. The effective
input voltage range of the circuit is extended by the fact
that resistors RT attenuate the common mode voltage
seen by the op amp inputs. For the LT1996, the most
useful resistors for RG are the M9 and P9 50kΩ resistors,
because they do not have diode clamps to the supplies and
therefore can be taken outside the supplies. As before, the
input CM of the op amp is the limiting factor and is set by
the voltage at the op amp +input, VINT. By superposition
we can write:
VINT = VEXT • (RF||RT)/(RG + RF||RT) + VREF • (RG||RT)/
(RF + RG||RT) + VTERM • (RF||RG)/(RT + RF||RG)
Solving for VEXT:
Table 6. HighV CM Connections Giving Difference Gains
for the LT1996
Gain
VIN+
VIN–
9
P9
M9
10
10/9 • VLIM - VREF/9
9
P9
M9
P27, M27
37
37/9 • VLIM – VREF/9 – 3 • VTERM
9
P9
M9
P81, M81
91
91/9 • VLIM – VREF/9 – 9 • VTERM
9
P9
M9
P27||P81 118 118/9 • VLIM – VREF/9 – 12 • VTERM
M27||M81
VCC
RG
VIN–
but this exceeds the 60V absolute maximum rating of the
P9, M9 pins, so –60V becomes the de facto negative
common mode limit. Several more examples of high CM
circuits are shown in Figures 15, 16, 17 for various
supplies.
RT
VOUT = GAIN • (VIN+ – VIN–)
VEE GAIN = RF/RG
RF
VREF
VTERM
HIGH CM VOLTAGE DIFFERENCE AMPLIFIER
INPUT CM TO OP AMP IS ATTENUATED BY
RESISTORS RT CONNECTED TO VTERM.
12V
7
10V
8
450k/81
450k
9
450k/27
10
450k/9
1
450k/9
2
450k/27
4pF
3
450k/81
450k
4pF
–
6
VIN+
VIN–
INPUT CM RANGE
= –60V TO 18.9V
= (10.11)(1) – 0.11(2.5) – 9(10) = –80.2V
VOUT
+
RT
= (10.11) • (10.8) – 0.11(2.5) – 9(10) =
18.9V
MIN VEXT = 91/9 • (VEE + 1V) – VREF/9 – 9 • VTERM
–
RG
VIN+
(= VEXT)
MAX VEXT = 91/9 • (VCC – 1.2V) – VREF/9 – 9 • VTERM
and:
RT
RF
VEXT = (1 + RG/(RF||RT)) • (VINT – VREF • (RG||RT)/
(RF + RG||RT) – VTERM • (RF||RG)/(RT + RF||RG))
Given the values of the resistors in the LT1996, this
equation has been simplified and evaluated, and the resulting equations provided in Table 6. As before, substituting VCC – 1.2 and VEE + 1 for VLIM will give the valid
upper and lower common mode extremes respectively.
Following are sample calculations for the case shown in
Figure 14, right-hand side. Note that P81 and M81 are
terminated so row 3 of Table 6 provides the equation:
Max, Min VEXT
(Substitute VCC – 1.2,
VEE + 1 for VLIM)
Noise
Gain
VOUT
+
REF 5
2.5V
LT1996
4
HIGH NEGATIVE CM VOLTAGE DIFFERENCE AMPLIFIER
IMPLEMENTED WITH LT1996.
RF = 450k, RG = 50k, RT 5.55k, GAIN = 9
VTERM = 10V = VCC = 12V, VREF = 2.5V, VEE = 0V.
1996 F14
Figure 14. Extending CM Input Range
1996f
19
LT1996
U
W
U
U
APPLICATIO S I FOR ATIO
3V
8
M81
9
M27
10
M9
VIN –
3V
7
VCC
6
VOUT
LT1996
OUT
1
REF
P9
2
5
P27 VEE
1.25V
3
P81
4
VIN +
VIN –
VIN +
VCM = 0.97V TO 1.86V
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VIN –
1
P9
2
P27 VEE
3
P81
4
VIN +
VOUT
6
VOUT
OUT
REF
5
1.25V
VIN –
8
M81
9
M27
10
M9
LT1996
VIN +
1
P9
2
P27 VEE
3
P81
4
6
VOUT
OUT
REF
5
1.25V
6
VOUT
VCM = –.78V TO 1.67V
VDM <–45mV
3V
7
VCC
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
3V
VIN +
3V
7
VCC
LT1996
6
VIN –
VCM = 1.11V TO 2V
VDM > 45mV
3V
8
M81
9
M27
10
M9
3V
8
M81
9
M27
10
M9
3V
8
M81
9
M27
10
M9
VIN –
7
VCC
6
LT1996
VOUT
OUT
1
REF
P9
2
5
P27 VEE
1.25V
3
P81
4
VIN +
1.25V
VCM = 4V TO 7.26V
VCM = 0.22V TO 3.5V
3V
8
M81
9
M27
10
M9
VIN –
7
VCC
6
LT1996
VOUT
OUT
1
REF
P9
2
5
P27 VEE
1.25V
3
P81
4
VIN +
VCM = –5V TO –1.74V
3V
3V
VIN –
VIN +
8
M81
9
M27
10
M9
7
VCC
6
LT1996
VOUT
OUT
1
REF
P9
2
5
P27 VEE
1.25V
3
P81
4
VIN –
3V
8
M81
9
M27
10
M9
7
VCC
6
LT1996
VOUT
OUT
1
REF
P9
2
5
P27 VEE
1.25V
3
P81
4
VIN +
1.25V
VCM = –1.28V TO 6.8V
VCM = 9.97V TO 18V
3V
VIN –
VIN +
8
M81
9
M27
10
M9
7
VCC
6
LT1996
VOUT
OUT
1
REF
P9
2
5
P27 VEE
1.25V
3
P81
4
VCM = –17V TO –8.9V
3V
3V
VIN –
VIN +
8
M81
9
M27
10
M9
7
VCC
6
LT1996
VOUT
OUT
1
REF
P9
2
5
P27 VEE
1.25V
3
P81
4
VIN –
VIN +
3V
8
M81
9
M27
10
M9
7
VCC
6
LT1996
VOUT
OUT
1
REF
P9
2
5
P27 VEE
1.25V
3
P81
4
1.25V
VCM = –2V TO 8.46V
VCM = 12.9V TO 23.4V
VCM = –23V TO –12.5V
1996 F15
Figure 15. Common Mode Ranges for Various LT1996 Difference Amp Configurations on VS = 3V, 0V, with Gain = 9
1996f
20
LT1996
U
U
W
U
APPLICATIO S I FOR ATIO
5V
8
M81
9
M27
10
M9
VIN –
5V
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
2.5V
3
P81
4
VIN +
6
VIN –
VOUT
VIN +
VCM = –0.83V TO 3.9V
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VIN –
1
P9
2
P27 VEE
3
P81
4
VIN +
VOUT
OUT
REF
5
2.5V
6
VIN –
VOUT
8
M81
9
M27
10
M9
VIN +
1
P9
2
P27 VEE
3
P81
4
6
VOUT
VCM = –0.56V TO 3.7V
VDM <–5mV
5V
7
VCC
LT1996
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
5V
VIN +
5V
7
VCC
LT1996
6
VIN –
VCM = 1.1V TO 4.2V
VDM > 5mV
5V
8
M81
9
M27
10
M9
5V
8
M81
9
M27
10
M9
OUT
REF
5
2.5V
6
VIN –
VOUT
5V
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
2.5V
3
P81
4
VIN +
6
VOUT
2.5V
VCM = 3.8V TO 15.3V
VCM = –3.7V TO 7.8V
5V
8
M81
9
M27
10
M9
VIN –
5V
5V
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
2.5V
3
P81
4
VIN +
VCM = –11.7V TO 0.3V
6
VIN –
VOUT
VIN +
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
2.5V
3
P81
4
6
VIN –
VOUT
5V
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
2.5V
3
P81
4
VIN +
6
VOUT
2.5V
VCM = –12.6V TO 15.6V
VCM = 9.8V TO 38.1V
5V
VIN –
8
M81
9
M27
10
M9
5V
5V
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
2.5V
3
P81
4
VIN +
VCM = –35.1V TO –6.8V
6
VIN –
VOUT
VIN +
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
2.5V
3
P81
4
6
VIN –
VOUT
VIN +
5V
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
2.5V
3
P81
4
6
VOUT
2.5V
VCM = –17.1V TO 19.5V
VCM = 12.8V TO 49.5V
VCM = –47.2V TO –10.5V
1996 F16
Figure 16. Common Mode Ranges for Various LT1996 Difference Amp Configurations on VS = 5V, 0V, with Gain = 9
1996f
21
LT1996
U
U
W
U
APPLICATIO S I FOR ATIO
5V
8
M81
9
M27
10
M9
VIN –
5V
7
VCC
LT1996
OUT
REF
5
1
P9
2
P27 VEE
3
P81
4
VIN +
6
8
M81
9
M27
10
M9
VIN –
VOUT
7
VCC
LT1996
1
P9
2
P27 VEE
3
P81
4
VIN +
–5V
5V
VIN –
7
VCC
OUT
REF
5
1
P9
2
P27 VEE
3
P81
4
2.5V
6
8
M81
9
M27
10
M9
VIN –
VOUT
1
P9
2
P27 VEE
3
P81
4
VIN +
1
P9
2
P27 VEE
3
P81
4
OUT
REF
5
6
VIN –
VOUT
LT1996
OUT
REF
5
–5V
VCM = –52.4V TO 49.8V
6
VIN –
VOUT
–5V
VOUT
VCM = –31.4V TO 0.6V
5V
OUT
REF
5
6
VIN –
VOUT
5V
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VIN +
6
VOUT
VCM = –60V TO –10.2V
5V
7
VCC
1
P9
2
P27 VEE
3
P81
4
6
–5V
LT1996
VIN +
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
VIN +
5V
7
VCC
1
P9
2
P27 VEE
3
P81
4
VIN –
VOUT
–5V
8
M81
9
M27
10
M9
VOUT
5V
8
M81
9
M27
10
M9
VCM = 4.6V TO 60V
5V
VIN +
6
7
VCC
1
P9
2
P27 VEE
3
P81
4
VIN +
6
–5V
LT1996
–5V
–5V
VIN –
OUT
REF
5
5V
8
M81
9
M27
10
M9
VCM = –40.4V TO 38.4V
8
M81
9
M27
10
M9
5V
VCM = –16.4V TO 15.6V
7
VCC
LT1996
VIN +
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
–5V
VIN +
–5V
–5V
7
VCC
–5V
VCM = –3.9V TO 4.8V
VDM <–5mV
7
VCC
5V
VIN –
VIN –
VOUT
5V
LT1996
VCM = –23.9V TO 8.1V
8
M81
9
M27
10
M9
6
5V
LT1996
VIN +
OUT
REF
5
–5V
VCM = –5V TO 3.7V
VDM > 5mV
VCM = –4.4V TO 4.2V
8
M81
9
M27
10
M9
5V
8
M81
9
M27
10
M9
OUT
REF
5
–5V
VCM = 7.6V TO 60V
6
VIN –
VOUT
VIN +
5V
8
M81
9
M27
10
M9
7
VCC
LT1996
OUT
1
REF
P9
2
5
P27 VEE
3
P81
4
6
VOUT
–5V
VCM = –60V TO –10.2V
1996 F17
Figure 17. Common Mode Ranges for Various LT1996 Difference Amp Configurations on VS = ±5V, with Gain = 9
1996f
22
LT1996
U
PACKAGE DESCRIPTIO
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699)
R = 0.115
TYP
6
0.38 ± 0.10
10
0.675 ±0.05
3.50 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
3.00 ±0.10
(4 SIDES)
PACKAGE
OUTLINE
1.65 ± 0.10
(2 SIDES)
PIN 1
TOP MARK
(SEE NOTE 6)
(DD10) DFN 1103
5
0.50
BSC
2.38 ±0.05
(2 SIDES)
2.38 ±0.10
(2 SIDES)
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
0.889 ± 0.127
(.035 ± .005)
5.23
(.206)
MIN
0.25 ± 0.05
0.50 BSC
0.75 ±0.05
0.200 REF
0.25 ± 0.05
1
10 9 8 7 6
3.20 – 3.45
(.126 – .136)
0.254
(.010)
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
DETAIL “A”
0.497 ± 0.076
(.0196 ± .003)
REF
0° – 6° TYP
GAUGE PLANE
0.50
0.305 ± 0.038
(.0197)
(.0120 ± .0015)
BSC
TYP
RECOMMENDED SOLDER PAD LAYOUT
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
1 2 3 4 5
0.86
(.034)
REF
1.10
(.043)
MAX
0.18
(.007)
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
BSC
0.127 ± 0.076
(.005 ± .003)
MSOP (MS) 0603
1996f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT1996
U
TYPICAL APPLICATIO
Micropower AV = 90 Instrumentation Amplifier
10
9
8
VOUT
6
450k
450k/81
+
VM
7
4pF
450k/27
1/2 LT6011
–
450k/9
–
450k/9
+
VP
+
LT1996
450k
450k/27
1/2 LT6011
450k/81
–
1
2
4pF
3
4
5
1996 TA02
Bidirectional Controlled Current Source
AC Coupled Amplifier
VS +
VIN –
VIN +
8
M81
9
M27
10
M9
VS +
VS +
8
M81
9
M27
10
M9
7
6
LT1996
5
R1
10k
0.1µF
VIN
4
ILOAD =
VS –
1
P9
2
P27
3
P81
9(VIN + – VIN –)
10kΩ
7
6
LT1996
VIN –
VOUT
5
4
VS –
VIN +
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
7
6
LT1996
10k
–
4
LT6010
VS –
GAIN = 117
BW = 4Hz TO 5kHz
VOUT+
VOCM
5
+
1
P9
2
P27
3
P81
Differential Input/Output G = 9 Amplifier
USE VOCM TO SET THE DESIRED
OUTPUT COMMON MODE LEVEL
10k
VOUT–
1996 TA03
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1990
High Voltage Difference Amplifier
±250V Input Common Mode, Micropower, Pin Selectable Gain = 1, 10
LT1991
Precision, 100µA Gain Selectable Amplifier
Gain Resistors of 450k, 150k, 50k
LT1995
30MHz, 1000V/µs Gain Selectable Amplifier
High Speed, Pin Selectable Gain = –7 to 8
LT6010/LT6011/LT6012
Single/Dual/Quad Precision Op Amp
Similar Performance as LT1996 Diff Amp, 135µA, 14nV√Hz,
Rail-to-Rail Out
LT6013/LT6014
Single/Dual Precision Op Amp
Lower Noise AV ≥ 5 Version of LT1991, 145µA, 8nV/√Hz,
Rail-to-Rail Out
LTC6910-X
Programmable Gain Amplifiers
3 Gain Configurations, Rail-to-Rail Input and Output
1996f
24
Linear Technology Corporation
LT/TP 0205 1K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2005