CIRF
Circuit Intégré Radio Fréquence
CIRF
Circuit Intégré Radio Fréquence
Lecture I
Lecture I
•Introduction
•Baseband Pulse Transmission
•Digital Passband Transmission
•Circuit Non-idealities Effect
•Introduction
•Baseband Pulse Transmission
•Digital Passband Transmission
•Circuit Non-idealities Effect
Hassan Aboushady
Université Paris VI
M.H. Perrott
Hassan Aboushady
Université Paris VI
MIT OCW
M.H. Perrott
MIT OCW
1
M.H. Perrott
MIT OCW
M.H. Perrott
MIT OCW
Multidisciplinarity of radio design
M.H. Perrott
MIT OCW
H. Aboushady
University of Paris VI
2
References
CIRF
Circuit Intégré Radio Fréquence
• S. Haykin, “Communication Systems”, Wiley 1994.
• B. Razavi, “RF Microelectronics”, Prentice Hall, 1997.
• M. Perrott, “High Speed Communication Circuits and
Systems”, M.I.T.OpenCourseWare, http://ocw.mit.edu/,
Lecture I
Massachusetts Institute of Technology, 2003.
• D. Yee, “ A Design methodology for highly-integrated
low-power receivers for wireless communications”,
http://bwrc.eecs.berkeley.edu/, Ph.D. University of
California at berkeley, 2001.
•Introduction
•Baseband Pulse Transmission
•Digital Passband Transmission
•Circuit Non-idealities Effect
Hassan Aboushady
University of Paris VI
H. Aboushady
University of Paris VI
Digital Baseband Transmission
Matched Filter
Linear Receiver Model
Major sources of errors in the detection of
transmitted digital data:
ISI : InterSymbol Interference
The result of data transmission over a non-ideal
channel is that each received pulse is affected
by adjacent pulses.
• g(t) : transmitted pulse signal, binary symbol ‘1’ or ‘0’.
• w(t) : channel noise, sample function of a white noise process
of zero mean and power spectral density N0/2.
x(t ) = g (t ) + w(t ) , 0 ≤ t ≤ T
Channel Noise
y (t ) = g 0 (t ) + n(t )
• Filter Requirements, h(t) :
• Make the instantaneous power in the output signal g0(t) ,
measured at time t=T, as large as possible compared with
the average power of the output noise, n(t).
Detecting a pulse transmitted over a
channel that is corrupted by additive noise.
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h (t )
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Matched Filter for Rectangular Pulse
x(t)
hopt (t ) = k g (T − t )
h(t) for a rectangular Pulse:
h(t) = g(T-t)
g(-t)
g(t)
T
Error Rate due to Noise
t
T
t
T
t
Filter Output g(t)*h(t):
0 ≤ t ≤ Tb , the received signal:
In the interval
 + A + w(t ) , symbol '1' was sent
x (t ) = 
− A + w(t ) , symbol '0' was sent
Tb is the bit duration, A is the transmitted pulse amplitude
g(t)
• The receiver has prior knowledge of the pulse shape but not
its polarity.
• There are two possible kinds of error to be considered:
(1) Symbol ‘1’ is chosen when a ‘0’ was transmitted.
(2) Symbol ‘0’ is chosen when a ‘1’ was transmitted.
Implementation:
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PDF: Probability Density Function
• Gaussian Distribution:
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BER in a PCM receiver
Normalized PDF
µY= 0 and σY=1
 ( y − µY ) 2 
1
exp −
fY ( y ) =

2σ Y2 
σ Y 2π

H. Aboushady
Pe = p0 Pe 0 + p1 Pe1
Pe 0 = Pe1
p0 = p1 =
• Symbol ‘0’ was sent:
µY = − A , σ Y2 =
Pe 0 = P ( y > λ symbol '0' was sent)
N0
2Tb
Pe = Pe 0 = Pe1
∞
=
∫
Pe =
fY ( y 0) dy
λ
• Symbol ‘1’ was sent:
µY = + A , σ Y2 =
Pe 0 = P ( y < λ symbol '1' was sent)
1
2
 Eb 
1

erfc
 N0 
2


N0
2Tb
λ
=
H. Aboushady
∫
−∞
f Y ( y 1) dy
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H. Aboushady
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Why Modulation?
CIRF
Circuit Intégré Radio Fréquence
• In wired systems, coaxial lines exhibit superior shielding
at higher frequencies
• In wireless systems, the antenna size should be a
Lecture I
significant fraction of the wavelength to achieve a
reasonable gain.
•Introduction
•Baseband Pulse Transmission
•Digital Bandpass Transmission
•Circuit Non-idealities Effect
• Communication must occur in a certain part of the
spectrum because of FCC regulations.
• Modulation allows simpler detection at the receive end in
the presence of non-idealities in the communication
channel.
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Université Paris VI
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Message Source
Transmitter
• mi : one symbol every T seconds
• Symbols belong to an alphabet of M symbols: m1, m2, …, mM
• Message output probability:
• Signal Transmission Encoder: produces a vector si made up of N
real elements, where N ≤ M .
• Modulator: constructs a distinct signal si(t) representing mi of
duration T .
P ( m1 ) = P ( m2 ) = ... = P ( mM )
pi
1
= P (mi ) =
M
T
• Energy of si(t) :
Ei =
∫ s (t )
2
i
dt , i = 1, 2, ..., M
0
• si(t) is real valued and transmitted every T seconds.
• Example: Quaternary PCM, 4 symbols: 00, 01, 10, 11
H. Aboushady
University of Paris VI
University of Paris VI
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Examples of Transmitted signals: si(t)
Communication Channel
• The modulator performs a step change in the amplitude, phase
or frequency of the sinusoidal carrier
• ASK:
Amplitude Shift Keying
• Two Assumptions:
•The channel is linear (no distortion).
• si(t) is perturbed by an Additive, zero-mean, stationnary,
White, Gaussian Noise process (AWGN).
• PSK:
Phase Shift Keying
• Received signal x(t) :
 0≤t ≤T
x (t ) = si (t ) + w(t ), 
i = 1, 2,..., M
• FSK:
Frequency Shift Keying
Special case: Symbol Duration T = Bit Duration, Tb
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Receiver
H. Aboushady
University of Paris VI
Coherent and Non-Coherent Detection
•TASK: observe received signal, x(t), for a duration T and make a
best estimate of transmitted symbol, mi .
•Detector: produces observation vector x .
•Signal Transmission Decoder: estimates m̂ using x, the
modulation format and P(mi) .
• The requirement is to design a receiver so as to minimize
the average probability of symbol error:
Pe =
• Coherent Detection:
- The receiver is time synchronized with the transmitter.
- The receiver knows the instants of time when the
modulator changes state.
- The receiver is phase-locked to the transmitter.
• Non-Coherent Detection:
- No phase synchronism between transmitter and receiver.
M
∑ P(mˆ ≠ m )P(m )
i
i
i =1
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University of Paris VI
H. Aboushady
University of Paris VI
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Coherent Binary PSK:
• M=2, N=1
s1 (t ) =
0 ≤ t ≤ Tb
Generation and Detection of Coherent Binary PSK
2 Eb
cos( 2πf c t )
Tb
s2 (t ) =
2 Eb
cos( 2πf c t + π )
Tb
• To ensure that each transmitted bit contains an integral number of
cycles of the carrier wave, fc =nc/Tb, for some fixed integer nc.
• One basis function: φ1 (t ) =
• Assuming white Gaussian Noise with
PSD = N0/2 ,
The Bit Error Rate for coherent binary
PSK is:
Pe =
 Eb 
1

erfc
 N0 
2


Binary PSK Transmitter
2
cos( 2πf ct ) , 0 ≤ t ≤ Tb
Tb
• Signal constellation consists of
two message points:
Tb
Coherent Binary PSK Receiver
∫
s11 = s1 (t ) φ1 (t ) dt = Eb
0
Tb
∫
s21 = s2 (t ) φ1 (t ) dt = − Eb
0
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University of Paris VI
Coherent QPSK:
• M=4, N=2:
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Constellation Diagram of Coherent QPSK System
• Assuming AWGN with PSD = N0/2 ,
The Bit Error Rate for coherent QPSK is:
 2E
π


cos 2πf ct + (2i − 1)  , 0 ≤ t ≤ T
si (t ) =  T
4

0
, elsewhere

π
2E
s1 (t ) =
cos( 2π f ct + )
T
4
π
2E
s 2 (t ) =
cos( 2π f c t + 3 )
T
4
π
2E
s3 (t ) =
cos( 2π f ct + 5 )
T
4
π
2E
s 4 (t ) =
cos( 2π f c t + 7 )
T
4
2
cos( 2π f ct ) , 0 ≤ t ≤ T
T
2
φ2 (t ) =
sin( 2π f c t ) , 0 ≤ t ≤ T
T
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Pe =
 E/2
1
erfc
 N0
2





with E = 2 Eb
Pe =
• Two basis function: φ1 (t ) =
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University of Paris VI
 Eb 
1

erfc
 N0 
2


Identical to BPSK
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University of Paris VI
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QPSK waveform: 01101000
Generation and Detection of Coherent QPSK Signals
QPSK Transmitter
Coherent QPSK Receiver
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University of Paris VI
H. Aboushady
University of Paris VI
Power Spectra of BPSK ,QPSK and M-ary PSK
• Power Spectral Density of
an M-ary PSK signal:
8-PSK
S B ( f ) = 2 E sinc 2 (Tf )
= 2 Eb log 2 M sinc 2 (Tb f log 2 M )
S B ( f ) = 4 Eb sinc 2 ( 2Tb f )
BPSK
Lecture I
•Introduction
•Baseband Pulse Transmission
•Digital Passband Transmission
•Circuit Non-idealities Effect
S B ( f ) = 6 Eb sinc 2 (3Tb f )
QPSK
H. Aboushady
CIRF
Circuit Intégré Radio Fréquence
T = Tb log 2 M
• Symbol Duration:
S B ( f ) = 2 Eb sinc 2 (Tb f )
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Université Paris VI
University of Paris VI
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QPSK Receiver
φ1 (t ) =
Receiver Circuit Non-Idealities
2
cos( 2π f c t )
T
Threshold = 0
T
T
Circuit Noise (Thermal, 1/f)
Decision
Device
∫ dt
path I
Gain Mismatch
0
011011
Binary Data
Multiplexer
Phase Mismatch
antenna
path Q
T
DC Offset
Decision
Device
∫ dt
0
Frequency Offset
T
φ1 (t ) =
2
sin(2π f c t )
T
Threshold = 0
Local Oscillator phase noise
01
Q
11
I
QPSK Constellation Diagram
00
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10
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Circuit Noise
H. Aboushady
University of Paris VI
Gain Mismatch
Q
A(1+α/2)
Circuit Noise:
• Thermal Noise
• Resistors
• Transistors
I
A(1-α/2)
2
Q
1.5
• Flicker (1/f) Noise
• MOS transistors
1
0.5
I
0
-0.5
-1
-1.5
-2
-2
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University of Paris VI
H. Aboushady
-1.5
-1
-0.5
0
0.5
1
1.5
2
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Phase Mismatch
DC Offset
offset
I
Q
2
offset
Q
2
Q
1.5
1.5
1
1
0.5
0.5
I
0
I
0
-0.5
-0.5
-1
-1
-1.5
-1.5
-2
-2
-1.5
-1
-0.5
0
0.5
1
H. Aboushady
1.5
2
-2
-2
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Frequency Offset
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-1.5
-1
-0.5
0
0.5
1
1.5
2
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Local Oscillator Phase Noise
2
Q
1.5
1
0.5
I
0
-0.5
-1
-1.5
-2
-2
H. Aboushady
-1.5
-1
-0.5
0
0.5
1
1.5
2
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Reciprocal Mixing
CIRF
Circuit Intégré Radio Fréquence
Lecture II
•Introduction
•Negative Resistance Oscillators
•Integrated Passive Components
•Phase Noise in Local Oscillators
Hassan Aboushady
University of Paris VI
H. Aboushady
University of Paris VI
References
CIRF
Circuit Intégré Radio Fréquence
• M. Perrott, “High Speed Communication Circuits and
Systems”, M.I.T. OpenCourseWare, http://ocw.mit.edu/,
Massachusetts Institute of Technology, 2003.
Lecture II
• T. Lee, “The Design of CMOS Radio-Frequency Integrated
Circuits”, Cambridge University Press, 2004.
•Introduction
•Negative Resistance Oscillators
•Integrated Passive Components
•Phase Noise in Local Oscillators
• B. Razavi, “Design of Analog CMOS Integrated Circuits”,
Mc Graw-Hill, 2001.
Hassan Aboushady
University of Paris VI
H. Aboushady
University of Paris VI
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QPSK Receiver
φ1 (t ) =
2
cos( 2π f c t )
T
Threshold = 0
T
T
Decision
Device
∫ dt
path I
0
011011
Binary Data
Multiplexer
antenna
path Q
T
Decision
Device
∫ dt
0
T
φ2 (t ) =
2
sin(2π f c t )
T
Threshold = 0
01
Q
11
I
QPSK Constellation Diagram
00
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10
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CIRF
Circuit Intégré Radio Fréquence
Lecture II
•Introduction
•Negative Resistance Oscillators
•Integrated Passive Components
•Phase Noise in Local Oscillators
Hassan Aboushady
University of Paris VI
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CIRF
Circuit Intégré Radio Fréquence
Lecture II
•Introduction
•Negative Resistance Oscillators
•Integrated Passive Components
•Phase Noise in Local Oscillators
Hassan Aboushady
University of Paris VI
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Vertical Metal Capacitors
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Lateral Metal Capacitors
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Vertical Mesh Metal Capacitors
K.T. Christensen,”Low Power RF Filtering for CMOS Transceivers”, Ph.D. Denmark Technical
University, 2001, http://phd.dtv.dk/2001/oersted/k_t_christensen.pdf
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CIRF
Circuit Intégré Radio Fréquence
s
Lecture II
•Introduction
•Negative Resistance Oscillators
•Integrated Passive Components
•Phase Noise in Local Oscillators
Hassan Aboushady
University of Paris VI
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QPSK Receiver
φ1 (t ) =
Local Oscillator Phase Noise
2
cos( 2π f c t )
T
Threshold = 0
T
T
Decision
Device
∫ dt
path I
0
011011
Binary Data
Multiplexer
antenna
path Q
T
Decision
Device
∫ dt
0
T
φ2 (t ) =
2
sin(2π f c t )
T
Threshold = 0
01
Q
11
I
QPSK Constellation Diagram
00
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Reciprocal Mixing
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Phase Noise in Oscillators
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