Theoretical and Experimental Investigations on
Snubber Circuits for High Voltage Valves of
FACTS-Equipment for Over-voltage Protection
Submitted to
The Faculty of Engineering at the Friedrich-Alexander University of
Erlangen-Nuremberg
to obtain the degree
DOKTOR-INGENIEUR
presented by
Jamal Alnasseir
Erlangen 2007
As dissertation approved by
The Faculty of Engineering Science
of the Friedrich-Alexander University of Erlangen-Nuremberg
Day of submission:
26.03.2007
Day of examination:
31.05.2007
Dean:
Prof. Dr.-Ing. A. Leipertz
Examiners:
Prof. Dr.-Ing. G. Herold
Prof. Dr.-Ing. J. Petzoldt
Theoretische und experimentelle Untersuchung
der Schutzbeschaltung von
Hochspannungsventilen für FACTS-Anlagen
Der Technischen Fakultät
der Friedrich-Alexander Universität Erlangen-Nürnberg
zur Erlangung des Grades
DOKTOR-INGENIEUR
vorgelegt von
Jamal Alnasseir
Erlangen 2007
Als Dissertation genehmigt von
der Technischen Fakultät
der Friedrich-Alexander-Universität Erlangen-Nürnberg
Tag der Einreichung:
26.03.2007
Tag der Promotion:
31.05.2007
Dekan:
Prof. Dr.-Ing. A. Leipertz
Berichterstatter:
Prof. Dr.-Ing. G. Herold
Prof. Dr.-Ing. J. Petzoldt
Preface
Preface
This dissertation follows from my research work done as a doctoral student at the Institute for
Electrical Power System in the Friedrich-Alexander University of Erlangen-Nuremberg, Germany.
I would like to express my deep thankfulness to my supervisor Univ.-Prof. Dr.-Ing. habil. Gerhard
Herold for giving me the chance to this interesting research area, for his invaluable friendship, for his
completely trusts with me, and for the excellent working conditions.
I would like to thank Prof. Dr.-Ing. habil J. Petzoldt, TU Ilmenau, for being the external examiner
of my dissertation. A warm thanks also to Prof. Dr.-Ing. J. Schlücker for being one of my examiners.
My sincere thanks go to Prof. Dr.-Ing A. Hamzeh for his help and support.
I want to thank all my colleagues at my institute, Dr.-Ing Mayer, Prof. Dr-Ing. Jäger, Mrs. Biegel,
Mrs Gambel, Mrs. Strößner, Dipl.-Ing. Braisch, Dr.-Ing. Gawlik, Dipl.-Ing. Ebner, Dipl.-Ing.
Rubenbauer, Dipl.-Ing. Keil, Dipl.-Ing. Ramold, Dipl.-Ing. Rasic, Dipl.-Ing .Weiland, Dipl.-Ing.
Mladenovic, and Mr. Domhardt, for the unforgettable friendship, for their support, help and consulting
in technology, society and in everyday life. They make me truly enjoy the research work and always
have the feeling as at home.
Warm and special thanks to my colleagues Dr.-Ing Weidl, M.Sc. Gamil, Mr. Leuschner, Mr.
Ruschig and Mr. Oschmann and M.Sc. M. Assem Alrabat. Their support, friendship, discussion and
time for me are much appreciated.
Finally, my sincere gratitude goes to my parents, my sisters and brothers for their ongoing
encouragement and trust. Without their great support, this work would not have been done.
Vorwort
Vorwort
Die vorliegende Dissertation entstand während meines Aufenthaltes als Promotionsstudent am
Lehrstuhl für Elektrische Energieversorgung der Friedrich-Alexander-Universität Erlangen-Nürnberg.
Meinem Doktorvater Herrn Prof. Dr.-Ing. habil G. Herold bin ich Ermöglichung dieser
hochinteressanten Arbeit, seine nette Unterstützung, für sein in mich gesetztes Vertrauen und die
ausgezeichneten Arbeitsbedingungen besonders zu Dank verpflichtet.
Herrn Prof. Dr.-Ing habil J. Petzoldt möchte ich für die Übernahme des Korreferates danken.
Herrn Prof. Dr.-Ing J. Schlücker danke ich herzlich für die Teilnahme an meinem Rigorosum als
fachfremder Prüfer.
Mein aufrichtiger Dank gehört Prof Dr.-Ing A. Hamzeh für seine Hilfe und Unterstützung.
Allen meinen Kolleginnen und Kollegen am Institut, besondere Dr.-Ing Mayer, Prof. Jäger, Fr..
Biegel, Fr. Gambel, Fr. Strößner, Dipl.-Ing. Braisch, Dr.-Ing. Gawlik, Dipl.-Ing. Ebner, Dipl.-Ing.
Rubenbauer, Dipl.-Ing. Keil, Dipl.-Ing. Ramold, Dipl.-Ing. Rasic, Dipl.-Ing. Weiland, Dipl.-Ing.
Mladenovic, and Mr. Domhardt, sei für ihren netten Hilfsstellung, die sehr gute Zusammenarbeit und
das angenehme Arbeitsklima besondere gedankt. Ich habe mich am Lehrstuhl stets wie zuhause
gefühlt.
Darüber hinaus möchte ich mich bei meinen Kollegen Dr.-Ing Weidl, M.Sc. Gamil, Mr.
Leuschner, Mr. Ruschig and Mr. Oschmann and M.Sc. M. Assem Alrabat, für ihre Unterstützung, die
zahlreichen Diskussionen und die aufgewandte Zeit bedanken.
Schließlich geht mein lieber Dank an meine Eltern und meine Geschwister für ihre fortwährende
Ermutigung sowie für ihr Vertrauen. Ohne das hätte ich die Arbeit nicht durchführen können.
Contents
Contents
1 Introduction............................................................................................................................................ 1
2 Einleitung ............................................................................................................................................... 3
3 Power Semiconductor Devices .............................................................................................................. 6
3.1. Introduction........................................................................................................................................ 6
3.2 Principal high power device characteristics and requirements ........................................................... 8
3.2.1 Voltage and current rating............................................................................................................ 8
3.2.2 Losses and speed of switching ..................................................................................................... 9
3.2.3 Parameter trade-off of the devices ............................................................................................... 9
3.3 Power device material....................................................................................................................... 10
3.4 Perspectives on power devices equipment........................................................................................ 10
3.5 Power diodes..................................................................................................................................... 10
3.5.1 Dynamic characteristics of power switching diodes.................................................................. 11
3.5.2 Reverse-recovery characteristic ................................................................................................. 12
3.5.3 Power diodes types..................................................................................................................... 12
3.6 Insulated Gate Bipolar Transistor (IGBT) ........................................................................................ 13
3.6.1 Switching characteristic of the IGBT ........................................................................................ 15
3.7 Types of high-power thyristor devices.............................................................................................. 15
3.7.1 Thyristors ................................................................................................................................... 16
3.7.1.1 Switching characteristic of thyristor (SCR) ........................................................................ 19
3.7.2 Gate turn-off thyristor (GTO) .................................................................................................... 20
3.7.2.1 Switching characteristics of GTO (turn-on and turn-off process) ...................................... 22
3.7.3 MOS Turn-Off Thyristor ........................................................................................................... 23
3.7.4 Emitter Turn-on Thyristor (ETO) .............................................................................................. 24
3.7.5 Integrated Gate-Commutated Thyristor (GCT and IGCT) ........................................................ 25
3.7.6 MOS-Controlled Thyristor (MCT) ............................................................................................ 26
3.8 Control characteristic of power devices............................................................................................ 27
4 FACTS concept and general system considerations ............................................................................ 29
4.1 Need of transmission interconnection............................................................................................... 29
4.2 Opportunities of FACTS................................................................................................................... 29
4.3 Basic types of FACTS controller ...................................................................................................... 30
4.3.1 Series Controller: ....................................................................................................................... 30
4.3.2 Shunt controller.......................................................................................................................... 31
4.3.3 Combined series-series controller.............................................................................................. 31
4.3.4 Combined series-shunt controller .............................................................................................. 31
4.4 Relative importance of different types of controllers ....................................................................... 31
4.5 Description and definition of FACTS Controllers............................................................................ 34
4.6 Shunt connected controllers .............................................................................................................. 35
4.6.1 Static Synchronous Compensator .............................................................................................. 35
4.6.2 Static Synchronous Generator (SSG)......................................................................................... 36
4.6.3 Battery Energy Storage System (BESS) .................................................................................... 36
4.6.4 Superconducting Magnetic Energy Storage (SMES)................................................................. 36
4.6.5 Static Var Compensator (SVC).................................................................................................. 36
4.6.6 Thyristor Controlled Reactor (TCR).......................................................................................... 37
4.6.7 Thyristor-Switched Reactor (TSR) ............................................................................................ 37
I
Contents
4.6.8 Thyristor Switched Capacitor (TSC) ......................................................................................... 37
4.6.9 Static Var Generator or Absorber (SVG)................................................................................... 37
4.6.10 Static Var System (SVS).......................................................................................................... 37
4.6.11 Thyristor controlled braking Resistor (TCBR) ........................................................................ 37
4.7 Series connected controllers.............................................................................................................. 38
4.7.1 Static Synchronous Series Compensator (SSSC) ...................................................................... 38
4.7.2 Interline Power Flow Controller (IPFC) .................................................................................... 38
4.7.3 Thyristor Controlled Series Capacitor (TCSC).......................................................................... 38
4.7.4 Thyristor-switched Series Capacitor (TSSC)............................................................................. 38
4.7.5 Thyristor-Controlled Series Reactor (TCSR) ............................................................................ 39
4.7.6 Thyristor-switched Series Reactor (TSSR)................................................................................ 40
4.8 Combined Shunt Series Connected Controllers................................................................................ 40
4.8.1 Unified Power Flow Controller (UPFC).................................................................................... 40
4.8.2 Thyristor-Controlled Phase Shifting Transformer (TCPST) ..................................................... 40
4.8.3 Inter-phase Power Controller (IPC) ........................................................................................... 40
4.8.4 Thyristor-Controlled Voltage Limiter (TCVL).......................................................................... 42
4.8.5 Thyristor-Controlled Voltage Regulator (TCVR)...................................................................... 42
4.9 Benefits of FACTS Controllers: ....................................................................................................... 42
5 Voltage-Sourced Converters ................................................................................................................ 43
5.1 Basic concept of Voltage-Sourced Converter (VSC) ....................................................................... 43
5.2 Single-Phase, Voltage Source Converter Circuits ............................................................................ 45
5.3 Output voltage control of single-phase converter ............................................................................. 46
5.3.1 Output voltage control via input voltage regulation .................................................................. 46
5.3.2 Phase control of the converter legs ............................................................................................ 47
5.3.3 Sinusoidal Pulse Width Modulation (SPWM) ........................................................................... 47
5.3.3.1 Full-bridge SPWM converter.............................................................................................. 47
5.3.4 Single-phase SPWM converter with Uni-polar Switching Scheme........................................... 48
5.4 Three-phase Voltage-Source Converters .......................................................................................... 49
5.4.1 Converter waveforms with 180° conduction angle.................................................................... 50
5.4.2 Converter waveforms with 120°conduction angle..................................................................... 50
5.5 Output voltage control of thee-phase converters .............................................................................. 50
5.6. Three-level Voltage-Sourced Converter .......................................................................................... 51
5.6.1. Pulse width modulation (PWM) for Three Level Converter .................................................... 52
6 Current-Sourced Converter and Self- and Line-Commutated ............................................................. 55
6.1 Introduction and basic concept of Current-Sourced Converter ........................................................ 55
6.2 Single-phase bridge rectifier ............................................................................................................. 57
6.3 Three-phase bridge rectifier .............................................................................................................. 57
6.4 Phase-controlled AC-DC converters................................................................................................. 58
6.4.1 Single-phase, fully-controlled bridge rectifier ........................................................................... 59
6.5 Three-phase fully controlled bridge converters ................................................................................ 59
6.6 Current-Sourced Converter with turn-off devices ............................................................................ 61
6.7 Current-Sourced versus Voltage-Sourced Converters ...................................................................... 63
7 Snubber Circuits................................................................................................................................... 64
7.1 Introduction....................................................................................................................................... 64
7.2 Function and types of snubber circuits ............................................................................................. 64
7.3 Diode snubber ................................................................................................................................... 65
7.3.1 Capacitive snubber..................................................................................................................... 65
7.3.2 Effect of adding a snubber resistor ............................................................................................ 66
II
Contents
7.4 Snubber circuits for Thyristors ......................................................................................................... 69
7.5 Need for snubber circuits for the transistor....................................................................................... 71
7.6 Turn-off snubber: .............................................................................................................................. 73
7.7 Over-voltage snubber........................................................................................................................ 77
7.8 Turn-on snubber................................................................................................................................ 79
7.9 GTO snubber circuit consideration ................................................................................................... 81
7.10 IGBT Snubber design ..................................................................................................................... 82
7.10.1 Over-voltage causes and their suppression .............................................................................. 82
7.10.2 Over-voltage suppression methods .......................................................................................... 83
7.10.3 Type of IGBT snubber circuits and their features.................................................................... 84
7.10.3.1 RC Snubber circuit ............................................................................................................ 84
7.10.3.2 Charge and discharge RCD Snubber circuit ..................................................................... 85
7.10.3.3 Discharge-suppressing RCD snubber circuit .................................................................... 85
7.10.3.4 C snubber circuit ............................................................................................................... 85
7.10.3.5 RCD snubber circuit.......................................................................................................... 86
7.10.4 Discharge-suppressing RCD snubber circuit design.................................................................... 87
7.10.4.1 A Study of applicability .................................................................................................... 87
7.10.4.2 Calculating the capacitance of the snubber circuit capacitor............................................ 88
7.10.4.3 Calculating snubber resistor.............................................................................................. 88
7.10.4.4 Snubber diode selection .................................................................................................... 88
7.10.4.5 Snubber circuit wiring precautions ................................................................................... 88
8 Simulation results of three-level VSC snubber circuits ....................................................................... 89
8.1 Introduction....................................................................................................................................... 89
8.2 Common snubber circuit for three level inverters ............................................................................ 89
8.3 Double snubber circuit for three level inverters................................................................................ 89
8.4 An optimized snubber design for three level inverters ..................................................................... 91
8.4.1 Performance of the optimized snubber design....................................................................... 97
8.5 Dual-use snubber circuit for three-level inverter .............................................................................. 97
8.6 Dual-inductive snubber circuit for three-level Inverter .................................................................... 98
8.7 Simulation and discussion of the results ........................................................................................... 99
8.7.1 Description of the PWM in Matlab®/SimulinkTM ................................................................... 100
8.7.2 Comparison of the proposed double snubber configuration and the common snubber circuit 102
8.7.2.1 Conclusion remarks........................................................................................................... 108
8.7.3 Comparing the common and the optimized snubber design .................................................... 109
8.7.4 Comparison of the common and the dual-use snubber circuits for different values for CS-D .. 116
8.7.5 Comparison of the dual-inductive- and common snubber circuit............................................ 122
9 Experimental investigation on the dual snubber circuit design ......................................................... 126
9.1 Introduction..................................................................................................................................... 126
9.2 Design Procedure ............................................................................................................................ 126
9.3 Driving the IGBT bridge circuit...................................................................................................... 128
9.4 The Logic of the Driving Circuit .................................................................................................... 129
9.5 Output Circuit ................................................................................................................................. 131
9.6 Experimental results........................................................................................................................ 131
9.6.1 Test of the dual-use snubber circuit ......................................................................................... 132
9.6.2 Test of the dual-inductive snubber circuit................................................................................ 133
10 Conclusion ....................................................................................................................................... 137
11 Zusammenfassung............................................................................................................................ 139
Appendix A1 Abbreviations and symbols ............................................................................................ 141
III
Contents
Appendix A2 List of Figures ................................................................................................................ 144
Appendix A3 List of Tables.................................................................................................................. 149
Appendix A4 References ...................................................................................................................... 150
IV
Inhaltverzeichnis
Inhaltsverzeichnis
1 Introduction............................................................................................................................................ 1
2 Einleitung ............................................................................................................................................... 3
3 Power Semiconductor Devices .............................................................................................................. 6
3.1. Introduction........................................................................................................................................ 6
3.2 Principal high power device characteristics and requirements ........................................................... 8
3.2.1 Voltage and current rating............................................................................................................ 8
3.2.2 Losses and speed of switching ..................................................................................................... 9
3.2.3 Parameter trade-off of the devices ............................................................................................... 9
3.3 Power device material....................................................................................................................... 10
3.4 Perspectives on power devices equipment........................................................................................ 10
3.5 Power diodes..................................................................................................................................... 10
3.5.1 Dynamic characteristics of power switching diodes.................................................................. 11
3.5.2 Reverse-recovery characteristic ................................................................................................. 12
3.5.3 Power diodes types..................................................................................................................... 12
3.6 Insulated Gate Bipolar Transistor (IGBT) ........................................................................................ 13
3.6.1 Switching characteristic of the IGBT ........................................................................................ 15
3.7 Types of high-power thyristor devices.............................................................................................. 15
3.7.1 Thyristors ................................................................................................................................... 16
3.7.1.1 Switching characteristic of thyristor (SCR) ........................................................................ 19
3.7.2 Gate turn-off thyristor (GTO) .................................................................................................... 20
3.7.2.1 Switching characteristics of GTO (turn-on and turn-off process) ...................................... 22
3.7.3 MOS Turn-Off Thyristor ........................................................................................................... 23
3.7.4 Emitter Turn-on Thyristor (ETO) .............................................................................................. 24
3.7.5 Integrated Gate-Commutated Thyristor (GCT and IGCT) ........................................................ 25
3.7.6 MOS-Controlled Thyristor (MCT) ............................................................................................ 26
3.8 Control characteristic of power devices............................................................................................ 27
4 FACTS concept and general system considerations ............................................................................ 29
4.1 Need of transmission interconnection............................................................................................... 29
4.2 Opportunities of FACTS................................................................................................................... 29
4.3 Basic types of FACTS controller ...................................................................................................... 30
4.3.1 Series Controller: ....................................................................................................................... 30
4.3.2 Shunt controller.......................................................................................................................... 31
4.3.3 Combined series-series controller.............................................................................................. 31
4.3.4 Combined series-shunt controller .............................................................................................. 31
4.4 Relative importance of different types of controllers ....................................................................... 31
4.5 Description and definition of FACTS Controllers............................................................................ 34
4.6 Shunt connected controllers .............................................................................................................. 35
4.6.1 Static Synchronous Compensator .............................................................................................. 35
4.6.2 Static Synchronous Generator (SSG)......................................................................................... 36
4.6.3 Battery Energy Storage System (BESS) .................................................................................... 36
4.6.4 Superconducting Magnetic Energy Storage (SMES)................................................................. 36
4.6.5 Static Var Compensator (SVC).................................................................................................. 36
4.6.6 Thyristor Controlled Reactor (TCR).......................................................................................... 37
V
Inhaltverzeichnis
4.6.7 Thyristor-Switched Reactor (TSR) ............................................................................................ 37
4.6.8 Thyristor Switched Capacitor (TSC) ......................................................................................... 37
4.6.9 Static Var Generator or Absorber (SVG)................................................................................... 37
4.6.10 Static Var System (SVS).......................................................................................................... 37
4.6.11 Thyristor controlled braking Resistor (TCBR) ........................................................................ 37
4.7 Series connected controllers.............................................................................................................. 38
4.7.1 Static Synchronous Series Compensator (SSSC) ...................................................................... 38
4.7.2 Interline Power Flow Controller (IPFC) .................................................................................... 38
4.7.3 Thyristor Controlled Series Capacitor (TCSC).......................................................................... 38
4.7.4 Thyristor-switched Series Capacitor (TSSC)............................................................................. 38
4.7.5 Thyristor-Controlled Series Reactor (TCSR) ............................................................................ 39
4.7.6 Thyristor-switched Series Reactor (TSSR)................................................................................ 40
4.8 Combined Shunt Series Connected Controllers................................................................................ 40
4.8.1 Unified Power Flow Controller (UPFC).................................................................................... 40
4.8.2 Thyristor-Controlled Phase Shifting Transformer (TCPST) ..................................................... 40
4.8.3 Inter-phase Power Controller (IPC) ........................................................................................... 40
4.8.4 Thyristor-Controlled Voltage Limiter (TCVL).......................................................................... 42
4.8.5 Thyristor-Controlled Voltage Regulator (TCVR)...................................................................... 42
4.9 Benefits of FACTS Controllers: ....................................................................................................... 42
5 Voltage-Sourced Converters ................................................................................................................ 43
5.1 Basic concept of Voltage-Sourced Converter (VSC) ....................................................................... 43
5.2 Single-Phase, Voltage Source Converter Circuits ............................................................................ 45
5.3 Output voltage control of single-phase converter ............................................................................. 46
5.3.1 Output voltage control via input voltage regulation .................................................................. 46
5.3.2 Phase control of the converter legs ............................................................................................ 47
5.3.3 Sinusoidal Pulse Width Modulation (SPWM) ........................................................................... 47
5.3.3.1 Full-bridge SPWM converter.............................................................................................. 47
5.3.4 Single-phase SPWM converter with Uni-polar Switching Scheme........................................... 48
5.4 Three-phase Voltage-Source Converters .......................................................................................... 49
5.4.1 Converter waveforms with 180° conduction angle.................................................................... 50
5.4.2 Converter waveforms with 120°conduction angle..................................................................... 50
5.5 Output voltage control of thee-phase converters .............................................................................. 50
5.6. Three-level Voltage-Sourced Converter .......................................................................................... 51
5.6.1. Pulse width modulation (PWM) for Three Level Converter .................................................... 52
6 Current-Sourced Converter and Self- and Line-Commutated ............................................................. 55
6.1 Introduction and basic concept of Current-Sourced Converter ........................................................ 55
6.2 Single-phase bridge rectifier ............................................................................................................. 57
6.3 Three-phase bridge rectifier .............................................................................................................. 57
6.4 Phase-controlled AC-DC converters................................................................................................. 58
6.4.1 Single-phase, fully-controlled bridge rectifier ........................................................................... 59
6.5 Three-phase fully controlled bridge converters ................................................................................ 59
6.6 Current-Sourced Converter with turn-off devices ............................................................................ 61
6.7 Current-Sourced versus Voltage-Sourced Converters ...................................................................... 63
7 Snubber Circuits................................................................................................................................... 64
7.1 Introduction....................................................................................................................................... 64
7.2 Function and types of snubber circuits ............................................................................................. 64
7.3 Diode snubber ................................................................................................................................... 65
7.3.1 Capacitive snubber..................................................................................................................... 65
VI
Inhaltverzeichnis
7.3.2 Effect of adding a snubber resistor ............................................................................................ 66
7.4 Snubber circuits for Thyristors ......................................................................................................... 69
7.5 Need for snubber circuits for the transistor....................................................................................... 71
7.6 Turn-off snubber: .............................................................................................................................. 73
7.7 Over-voltage snubber........................................................................................................................ 77
7.8 Turn-on snubber................................................................................................................................ 79
7.9 GTO snubber circuit consideration ................................................................................................... 81
7.10 IGBT Snubber design ..................................................................................................................... 82
7.10.1 Over-voltage causes and their suppression .............................................................................. 82
7.10.2 Over-voltage suppression methods .......................................................................................... 83
7.10.3 Type of IGBT snubber circuits and their features.................................................................... 84
7.10.3.1 RC Snubber circuit ............................................................................................................ 84
7.10.3.2 Charge and discharge RCD Snubber circuit ..................................................................... 85
7.10.3.3 Discharge-suppressing RCD snubber circuit .................................................................... 85
7.10.3.4 C snubber circuit ............................................................................................................... 85
7.10.3.5 RCD snubber circuit.......................................................................................................... 86
7.10.4 Discharge-suppressing RCD snubber circuit design.................................................................... 87
7.10.4.1 A Study of applicability .................................................................................................... 87
7.10.4.2 Calculating the capacitance of the snubber circuit capacitor............................................ 88
7.10.4.3 Calculating snubber resistor.............................................................................................. 88
7.10.4.4 Snubber diode selection .................................................................................................... 88
7.10.4.5 Snubber circuit wiring precautions ................................................................................... 88
8 Simulation results of three-level VSC snubber circuits ....................................................................... 89
8.1 Introduction....................................................................................................................................... 89
8.2 Common snubber circuit for three level inverters ............................................................................ 89
8.3 Double snubber circuit for three level inverters................................................................................ 89
8.4 An optimized snubber design for three level inverters ..................................................................... 91
8.4.1 Performance of the optimized snubber design....................................................................... 97
8.5 Dual-use snubber circuit for three-level inverter .............................................................................. 97
8.6 Dual-inductive snubber circuit for three-level Inverter .................................................................... 98
8.7 Simulation and discussion of the results ........................................................................................... 99
8.7.1 Description of the PWM in Matlab®/SimulinkTM ................................................................... 100
8.7.2 Comparison of the proposed double snubber configuration and the common snubber circuit 102
8.7.2.1 Conclusion remarks........................................................................................................... 108
8.7.3 Comparing the common and the optimized snubber design .................................................... 109
8.7.4 Comparison of the common and the dual-use snubber circuits for different values for CS-D .. 116
8.7.5 Comparison of the dual-inductive- and common snubber circuit............................................ 122
9 Experimental investigation on the dual snubber circuit design ......................................................... 126
9.1 Introduction..................................................................................................................................... 126
9.2 Design Procedure ............................................................................................................................ 126
9.3 Driving the IGBT bridge circuit...................................................................................................... 128
9.4 The Logic of the Driving Circuit .................................................................................................... 129
9.5 Output Circuit ................................................................................................................................. 131
9.6 Experimental results........................................................................................................................ 131
9.6.1 Test of the dual-use snubber circuit ......................................................................................... 132
9.6.2 Test of the dual-inductive snubber circuit................................................................................ 133
10 Conclusion ....................................................................................................................................... 137
11 Zusammenfassung............................................................................................................................ 139
VII
Inhaltverzeichnis
Appendix A1 Abbreviations and symbols ............................................................................................ 141
Appendix A2 List of Figures ................................................................................................................ 144
Appendix A3 List of Tables.................................................................................................................. 149
Appendix A4 References ...................................................................................................................... 150
VIII
Chapter 1
Introduction
1 Introduction
Recently the multi-level voltage-sourced inverters and converters have drawn tremendous interest
for high power applications. The typical Applications are modern HVDC systems and FACTS
(Flexible AC Transmission Systems). These inverters typically have ratings of 300 MVA and above
and they e.g. are used to increase the power transmission capacity of existing lines or to improve the
power system stability. Therefore high-power semiconductor devices like high-power GTOs, IGCTs
and IGBTs are the best suitable devices for these high rating converters. The power semiconductor
devices need protection systems to overcome the electrical stresses which are placed on the device
during the switching process (turn-off and turn-on) to safe levels within the electrical range of the
device. These protection systems are called snubber circuit. The conventional snubber circuits are RCD
and RLD. Snubber circuits would be used to protect the power semiconductor devices (all Thyristor and
transistor types) and reduce the electrical stresses brought to the device during the switching operations
under normal operation conditions and under several fault conditions. This means that the rate of the
anode-cathode voltage growth, dv/dt, and the rate of current increasement, di/dt, for e.g., GTOs, must
be limited below certain levels to prevent the destruction of the power semiconductor device caused by
the current crowding and the failure in turning-off, respectively. There are different types of snubber
circuit proposed by W. McMurray and T. Undeland. The snubber circuits can be divided into
unpolarized (RC) and polarized snubber circuits (RCD, and RLD) or turn-off and turn-on and overvoltage snubber circuits. The turn-on snubber consists of an inductor with a parallel resistor and a diode
in series to the power semiconductor device to limit the changing rate of the current di/dt. While the
turn-off snubbers consist of a capacitor in series with a parallel-connected resistor and diode.
In FACTS systems, the magnitude of the AC output voltage would be wanted to vary without
having to change the magnitude of the DC voltage, the three-level converter is a common converter
system. The power switching semiconductor device must be protected in this converter. Therefore
snubber circuit should be used. Normally, conventional RCD and RLD will be suitable as a protection
system.
A new circuit designs for the protection of three-level converters will be presented. Firstly, the socalled ‘Double Snubber Circuit’ optimises the behaviour of conventional RCD snubber circuits
especially in the direction of the over voltage protection and allows a minimizing of the total losses in
the entire circuit including the power semiconductors. The proposed circuit overcomes hereby the
limitations of many of the existing designs, because the losses and the over voltage can be controlled
using only a handful of additional passive elements. The second proposed design ‘optimized snubber
design’ still comprises most of the positive features as a low number of components, improved
efficiency due to the low number of snubber elements and power semiconductor losses, reduced overvoltage across the semiconductor devices and no balancing problems. The third design “Dual-use
snubber Circuit” has almost the same advantages of the second design and it has an additional
advantage while the turn-off resistor is more effective in the new location. The fourth proposed design
is “dual-inductive snubber circuit”, which is the same as the third design but with a new turn-on
snubber circuit. The turn-on snubber circuit has an extra inductor which is connected to the snubber
circuit resistor. The over-voltage across the switching devices is strongly suppressed and the current
peaks are limited much more. With these advantages, the new proposed snubber circuits can be used
for high power inverters as well as the Flexible AC Transmission Systems (FACTS). The presented
snubber circuits have been analyzed and confronted with different existing converter designs using a
simulation environment. The simulation results are compared with a standardized three level inverter
system to verify the opportunities of the new snubber design.
1
Chapter 1
Introduction
Chapter 2: is the German version of Chapter 1.
Chapter 3: “Power Semiconductor Devices”, in this chapter, the new generations of high power
semiconductor device will be discussed. The power semiconductor devices like GTOs, IGBTs, and
IGCTs are the most important elements in all power conversion applications. A review of the basic
characteristics of these power devices will be presented.
Chapter 4: “FACTS concept and general system considerations” introduce the main ideas of the
FACTS system technology which opens up new opportunities for controlling power and enhancing the
usable capacity of the present lines. FACTS controllers can be divided to four categories: series
controller, shunt controller, combined series-series controller, and series-shunt controller. The
mentioned types will briefly be described.
Chapter 5: “Basic concept of Voltage-Sourced Converter (VSC)” has always one polarity for the
direct voltage. The VSC is the building block of the most FACTS controller. The basic functioning of
the VSC and the internal topology of the converter valves, single-phase half-bridge and full-bridge, the
output voltage control, full-bridge SPWM and three-phase voltage-Sourced Converter were explained.
At last, the three-level voltage-sourced converter with the needed pulse width modulation (PWM)
technology will be discussed.
Chapter 6: “Current-Sourced Converter and Self and Line-Commutated” in which the direct
current has always has one polarity. The power flow reverses with the reversal of the DC current. The
three principle types of Current-Sourced Converter were diode converter, Line-commutated converter,
and Self-commutated converter will be discussed.
Chapter 7: “Snubber Circuits” is used to protect the power semiconductor device during the turnon (RLD) and turn-off (RCD) operations from the over- voltages and currents. A separate snubber
circuit unit for each power semiconductor device is usually used, which is composed of turn-off
capacitor to limit the dv/dt, turn-on inductor to limit di/dt, resistors and diodes. The common snubber
circuit which are used to keep the high rated power semiconductor devices like GTOs and IGBTs,
which have the turn-off capability, working in the Safe Operation Area (SOA) were studied in details.
Chapter 8: “Simulation results of Three-Level VSC snubber circuit”. The three-level converter,
which will be investigated here, is suitable for high-voltage applications (HVDC and FACTS) since it
guarantees equal voltage sharing between serially connected power devices. The Simulations were
performed by utilizing Matlab®/SimulinkTM software tools. The analyzed model comprises of a single
phase of a MV three-level converter. The results clarify the advantages of the new three-level converter
with the proposed snubber circuit designs, especially for the operation close to the SOA, the protection
from the over-voltages and the total losses in the converter system.
Chapter 9: “Experimental investigation on the dual snubber circuit design” The dual-use and
dual-inductive snubber circuit in three-level IGBT inverter system will be tested. For that, the needed
drive circuit will be achieved, this consists of three steps: the first step is to generate the required PWM
of the IGBTs, then build the essential generating logic using the ispDesignExpert software from Lattice
to get the necessary JEDEC file. The last step is to transfer the JEDEC file to the CPLD chip. The
resulted PWM will transmit to the IGBTs with optocoupler systems. Then the inverter system will be
loaded with inductive load to show the different voltages and currents in the load and IGBTs. The
results will be discussed and commented.
Chapter 10: “Conclusion” gives out an overview and main results of the dissertation.
Chapter 11: German version of Chapter 10.
Appendix A1 is a list of symbols and abbreviations. Appendix A2 is list of Figures. Appendix A3
list of Tables. Then the list of references finishes the dissertation.
2
Chapter 2
Einleitung
2 Einleitung
Insbesondere in den letzten Jahren haben die Mehrpunktumrichter großes Interesse u.a. für
Hochleistungsanwendungen erlangt. Die typischen Anwendungen sind moderne HVDC -Systeme und
FACTS-Analgen. Nennwerte von mehr als 300MVA sind typisch für diese Stromrichter.
Beispielsweise werden sie genutzt, um die Übertragungsfähigkeit von Übertragungsleitungen zu
steigern oder die Stabilität des Versorgungssystems zu verbessern. Die eingesetzten
Hochleistungshalbleiter wie z.B. GTO, IGCT und IGBT bilden die Basis für diese Stromrichter im
Höchstleistungsbereich. Diese Leistungshalbleiterschalter benötigen Schutzsysteme, um u.a. die
Spannungsbeanspruchungen zu überstehen, die auf die Schalter während der Schaltvorgänge
einwirken. Diese Schutzsysteme werden „Schutzbeschaltungen“ oder auch „Snubber circuits“ genannt.
Die konventionellen Schutzbeschaltungen bezeichnet man als RCD- und RLD-Snubber. Sie
werden genutzt, um die Leistungshalbleiter (alle Thyristor- und Transistorenarten) zu schützen und die
elektrischen Beanspruchungen zu verringern, welche während des Schaltbetriebs unter Normalbetrieb
und in verschiedenen Fehlerfällen einwirken. Das heißt, der Anstieg der Anoden-Kathoden-Spannung,
und der Stromanstieg müssen auf definierte Pegel begrenzt werden, um eine Zerstörung der
Leistungshalbleiter insbesondere im Fehlerfall zu vermeiden. Es gibt verschiedene Typen der von W.
McMurray und T. Undeland vorgeschlagenen Schutzbeschaltungen. Die Schutzbeschaltungen könnten
in symmetrische RC und unsymmetrische RCD und RLD Schutzbeschaltungen oder Einschalt-,
Ausschalt-,
und
Überspannungsschutzbeschaltungen
untergliedert
werden.
Die
Einschaltschutzbeschaltung besteht aus einer Drossel mit Nebenschlusswiderstand und
Reihenschlussdiode, um die Änderungsrate des Stromes zu begrenzen. Dagegen besteht die
Ausschaltschutzbeschaltung in der Regel aus einem Kondensator in Reihe mit einer Parallelschaltung
aus Widerstand und Diode.
In FACTS Systemen ist teilweise eine Anpassung der AC-Ausgangsspannung wünschenswert,
ohne dass die Höhe der DC-Spannung verändert werden muss. Der Dreipunktumrichter ist in gewissen
Grenzen hierzu in der Lage. Die Leistungshalbleiter müssen hierbei besonders geschützt werden.
Normalerweise werden hierzu RCD- und RLD-Snubber als Schutzsystem zweckmäßig sein.
In Rahmen der vorliegenden Arbeit werden neue Entwürfe für den Schutz der
Dreipunktumrichter vorgestellt. Zuerst optimiert die so genannte „Doppelschutzbeschaltung“ oder
„Double snubber circuit“ das Verhalten der konventionellen RCD-Schutzbeschaltungen, insbesondere
in die Richtung des Überspannungsschutzes. Zusätzlich erlaubt diese eine Minimierung der
Gesamtverluste. Die vorgeschlagene Schaltung besticht gegenüber vielen existierenden Entwürfen, da
die Verluste und die Überspannungen nur anhand einiger weniger zusätzlicher passiver Elemente
kontrolliert
werden
können.
Der
zweite
vorgeschlagene
Entwurf
„Optimierter
Schutzbeschaltungsentwurf“ oder auch „ optimized snubber circuit design“ beinhaltet viele dieser
positiven Eigenschaften wie z.B. niedrige Zahl der Komponenten, verbesserter Wirkungsgrad wegen
den wenigen Schutzbeschaltungselementen, niedrige Leistungshalbleiterverluste, reduzierte
Überspannungen über den Halbleitern und keine Probleme bzgl. einer unsymmetrischen
Spannungsaufteilung. Der Dritte Vorschlag „zweifache Schutzbeschaltung“ oder „Dual-use snubber
design“ hat fast identische Vorteile und dazu noch einen Vorteil, dass der Ausschaltwiderstand weit
effektiver an der neuen Position eingesetzt wird. Der vierte vorgeschlagene Entwurf ist "die Doppelinduktive Schutzbeschaltung“, oder auch „dual-inductive snubber circuit“, die bis auf eine veränderte
Einschaltschutzbeschaltung dem des dritten Entwurfs entspricht. Die Einschaltschutzbeschaltung
3
Chapter 2
Einleitung
umfasst eine zusätzliche Drosselspule, die mit dem Widerstand der Einschaltschutzbeschaltung parallel
geschaltet ist. Die Überspannung über dem Leistungshalbleiter wird noch stärker unterdrückt und der
Stromspitzen werden noch besser begrenzt. Mit diesen Vorteilen können die neuen
Schutzbeschaltungen für Hochleistungsstromrichter als auch in FACTS eingesetzt werden. Die
vorgestellte Schutzbeschaltung wurde eingehend analysiert und mit verschiedenen existierenden
Stromrichterentwürfen anhand einer Simulationsumgebung verglichen. Die Simulationsergebnisse
werden
mit einem üblichen Dreipunktumrichter verglichen, um die Vorteile der neuen
Schutzbeschaltung zu verifizieren.
Kapitel 2 ist die deutsche Version des Kapitels 1
Kapitel 3: „Leistungshalbleiter“, in diesem Kapitel werden die neueren Generationen von der
Hochleistungshalbleitern vorgestellt. Die Leistungshalbleiter wie GTO, IGBT und IGCT sind die
wesentlichen Betriebsmittel in allen Leistungsumwandlungsanwendungen. Zusätzlich wird das
Grundverhalten der Leistungshalbleiter kurz dargestellt.
Kapitel 4: „FACTS-Anlagen“ stellt die Grundprinzipien der FACTS Technologie vor, die neuen
Möglichkeiten zur Leistungsregelung und die zusätzlichen Nutzungsmöglichkeiten in den Systemen
der elektrischen Energievorsorgung. FACTS-Regler lassen sich prinzipiell in vier Kategorien einteilen:
Reihenschaltung,
Parallelschaltung,
kombinierte
Reihen-Reihenschaltung
und
ReihenParallelschaltung. Die oben genannten Typen werden kurz dargestellt.
Kapitel 5: „Stromrichter mit Spannungszwischenkreis“. Der VSC ist das am meisten verbreitete
Betriebsmittel innerhalb der FACTS. Die Grundfunktion des VSC und die Topologie der
Stromrichterventile, der einphasigen Halbbrücke und der Vollbrücke, die Vollbrücke mit SPWM und
der dreiphasige VSC werden vorgestellt. Schließlich wird der Dreipunktumrichter mit der zugehörigen
Pulsbreitenmodulation (PWM) diskutieret.
Kapitel 6: „Stromrichter mit Stromzwischenkreis (selbst- und netzgeführt)“, der Gleichstrom
fließt hierbei stets in einer Richtung. Der Leistungsfluss kehrt sich mit der Umkehrung der Spannung
um. Die drei Prinziptypen des CSC: Diodenstromrichter, netz- und selbstgeführter Stromrichter werden
diskutiert.
Kapitel 7: „Schutzbeschaltungen“. Diese sind zum Schutz der Leistungshalbleitergeräte vor
Überspannung und Überströmen während der Einschaltung und Ausschaltung vorzusehen. Eine
separate Schutzbeschaltung für jedes Leistungshalbleitergerät wird üblicherweise verwendet. Die
allgemeinen Schutzbeschaltungen, die zum Einhalten der Leistungshalbleiter wie GTO und IGBT
innerhalb eines sicheren Arbeitsbereiches (SOA) dienen, werden detailliert vorgestellt.
Kapitel
8:
„Simulationsergebnisse
der
Schutzbeschaltungen
eines
DreipunktSpannungszwischenkreis-Stromrichters“
Die
hier
untersuchten
Schaltungen
sind
für
Hochspannungsanwendungen (HGÜ und FACTS) geeignet, da eine symmetrische
Spannungsaufteilung zwischen mehreren in Reihen geschalteten Leistungshalbleiter gewährleistet
wird. Die zugehörigen Simulationen werden mit Matlab®/SimulinkTM durchgeführt. Das analytische
Model umfasst einen einphasiger Mittelspannungs-Dreipunktumrichter. Die Ergebnisse zeigen die
Vorteile des neuen Dreipunktumrichters mit den vorgeschlagenen Schutzbeschaltungsentwürfen,
besonders für den Betrieb in der Nähe der SOA, den Schutz vor Überspannungen und die reduzierten
Gesamtverluste im Stromrichtersystem.
4
Chapter 2
Einleitung
Kapitel 9: "Experimentelle Untersuchung der Mehrzweckschutzbeschaltung“ Hier werden IGBTs
in Dreipunktumrichtern geprüft. Eingangs wird der erforderliche Pulsbereiten-Modulator (PWM)
vorgestellt, dann die wesentlichen Komponenten der Steuerungslogik, erzeugt mittels
der
ispDesignExpert Software vom Lattice. Die resultierende Pulsbereiten-Modulation wird an die IGBTs
mittels Optokoppler übertragen. Im Anschluss wird ein Dreipunktumrichter in einer Modellumgebung
mit induktiver Last betrieben. Die Messergebnisse (Spannungen und Ströme) des Hardware-Aufbaus
werden vorgestellt.
Kapitel 10: "Zusammenfassung" gibt einen Überblick über die vorliegende Dissertation.
Kapitel 11: Deutsche Version von Kapitel 10.
Appendix A1 ist eine Liste von Notationen und Formelzeichen. Appendix A2 ist die Liste der
Abbildungen. Appendix A3 ist die Liste der Tabelle. Das Literaturverzeichnis schließt die Dissertation
ab.
5
Chapter 3
Power Semiconductor Devices
3 Power Semiconductor Devices
3.1. Introduction
The modern age of power electronics began with the introduction of Thyristors in the late
1950s. Now there are several types of power devices available for high-power and high-frequency
applications. The most notable ones are gate turn-off Thyristors, power Darlington transistors,
power MOSFETs, and insulated-gate bipolar transistors (IGBTs). Power semiconductor devices are
the most important functional elements in all power conversion applications. The power devices are
mainly used as switches to convert power from one form to another. They are used in motor control
systems, uninterrupted power supplies, high-voltage DC transmission, FACTS-Systems (Flexible
AC Transmission Systems), power supplies, induction heating and in many other power conversion
applications. A review of the basic characteristics of these power devices is presented in this section
[1].
A power semiconductor switch (power semiconductor device) is a component that is controlled
to either conduct a current when it is commanded ON or block a voltage when it is commanded
OFF. This change of conductivity is made possible in a semiconductor by specially arranged device
structures that control the carrier transportation. The time that it takes to change the conductivity is
also reduced to the microsecond level as compared to the millisecond level of a mechanical switch.
By employing this kind of switches, a designed electrical system can control the flow of electric
energy and shaping the electricity into desired forms. On the other hand, if a power semiconductor
device can block forward voltage as well as the reverse voltage during the OFF state, it is defined as
a symmetrical device. On the other hand, a power semiconductor device that can only block the
forward voltage during the OFF state is defined as an asymmetrical device. Most of the
semiconductor devices can only conduct forward current during the ON state [1], [2]. Therefore, the
symmetrical device has three operational states:
• Forward conduction mode.
• Forward blocking mode.
• Reverse blocking mode.
Fig.3-1 shows the operational modes for the both the symmetrical and the asymmetrical devices
respectively, for a symmetrical device, only two operation modes exist: forward conduction mode
and forward blocking mode.
The intent of this section is to give only general information about the most important power
semiconductor devices which are suitable for FACTS Controllers. Sufficient information is
provided for power system engineers to understand the option and their relevance to FACTS
applications. Generally, FACTS applications represent a three-phase power rating from tens to
hundreds of megawatts. Basically, FACTS Controllers based on an assembly of AC/DC or /and
DC/AC converters and/or high power AC switches. A converter is an assembly of valves (without
other equipment). Each valve in turn is an assembly of power devices along with snubber circuits
(damping circuits) as needed and turn-on/turn-off gate drive circuits. Similarly, each AC switch is
an assembly of back-to-back connected power devices along with their snubber circuits and turnon/turn-off gate drive circuits.
6
Chapter 3
Power Semiconductor Devices
I
Forward
conduction
INom
VD-Nom
V
Forward blocking
(a)
(b)
Fig.3-1 Device operational states for (a) symmetrical device and (b) asymmetrical device.
Nominal rating of large power devices is in range of 1-5 kA and 5-10 kV per device and their
useable circuit rating may be 25 to 50 % of their nominal rating. This conveys that the converters
and the AC switches would be an assembly of a large number of power devices. The converters, AC
switches, and devices are connected in series and/or in parallel in order to achieve the FACTS
Controllers rating and performance. Controllers in some cases may also be separated into singlephase assemblies. These considerations provide an interesting possibility and indeed a necessity for
the supplier to adapt modularity for an effective use of power devices. If properly utilized
modularity, cannot only reduce the cost through standardization of modules and sub-modules but it
can also an asset from the user perspective in terms or reliability, redundancy, and staged
investment [3].
The device rating and characteristics and their exploitations have a significant leverage on the
cost, performance, size, weight, and losses of FACTS controllers. The leverage includes the cost of
all that surrounds the devices including snubber circuits, gate-driver circuits, transformers, and
other magnetic equipment such as filters, cooling equipment, losses, operating performance and
maintenance requirements. For example, faster switching capability leads to fewer snubber
component, lower snubber losses and adaptation of concepts that produce less harmonics and faster
FACTS Controller response. This is also important for successful implementation of particular
concepts of FACTS Controllers, such as active filters.
There are many advanced circuit concepts used in low power industrial applications, mostly
driven by basic cost, the economic application at high power level is largely a function of advances
in devices. These concepts include pulse width modulation (PWM), soft switching, resonant
converters, choppers, and others. Therefore, the design of FACTS Controllers equipment would
usually be based on the devices with best available characteristic, even at high prices. Although the
cost of devices is basically important factor, it would be correct to say that availability of devices
with better characteristics provides an important leverage for the FACTS option. The availability of
devices is considered now a competitive edge for suppliers of FACTS technology to meet certain
specified performance at lowest evaluated cost. Thus, cost, performance, and market success of
7
Chapter 3
Power Semiconductor Devices
FACTS Controllers is very much tied to the progress in power semiconductors devices and their
packaging.
In general, high-power electronic devices are fast switches based on high-purity single-crystal
silicon wafers, designed for variety of switching characteristics. In their forward-conducting
direction, the devices may have control to turn on and to off the current flow when ordered to do so
by means of gate control. Some power devices are designed without the capability to block in
reverse direction, in which case they are provided with another reverse blocking device (diode) in
series or they are bypassed in reverse direction by another parallel device (diode). Basically, power
semiconductor devices consist of a variety of diodes, transistors, and thyristors [3].
3.2 Principal high power device characteristics and requirements
3.2.1 Voltage and current rating
Device cells for high power are usually single crystal silicon wafers which are about 75-125mm
in diameter, and now pushing towards 150mm in diameter. The same diameter device can be made
for high voltage with lower current and vice versa.
Potentially, silicon crystal has very high voltage breakdown strength of 200kV/cm and
resistivety somewhere in between metals and insulators. Doping with impurities can alter its
conduction characteristic. With doping, the number of carriers is increased and as a result, its
withstand voltage decreases and its current capability increases. Lower doping means higher
voltage capability, but it means also higher forward voltage drop and lower current capability. To
some extent current and voltage capabilities are interchangeable as mentioned above. A larger
diameter naturally means higher current capability. A 125mm device has a current-carrying
capability of 3000-4000A and a voltage-withstand capability in the range of 6000-10,000 volts.
With higher device rating, the total number of devices as well as the cost of all the surrounding
components decreases. The highest blocking capability along with other desirable characteristics is
somewhere in the range of 8-10kV for thyristors, 5-8kV for GTO’s, and 3-5kV for IGBTs. After
making various allowances for over-voltages and redundancy in a circuit, the usable device voltage
will be about half the blocking voltage capability. More often than not, it will be necessary to
connect devices in series for high-voltage valves. Ensuring equal sharing of voltage during turn-on,
turn-off and dynamic voltage changes becomes a major exercise for a valve designer in considering
trade-offs among various means to do and deciding on the best mix. One of these means is the
matching of device, especially the device-switching characteristics.
Large power devices can be designed to handle several hundreds Amperes of load current,
which generally makes it unnecessary to connect devices in parallel. However, it is often the shortcircuit current duty that determines the required current capacity in which case connecting two
matched devices directly in parallel on the same heat sink is a good solution. Devices are usually
required to ride through to blocked state after one cycle of offset fault current in an application
circuit. While it is a common practice to use fuses in industrial power electronics, the usage of fuses
is undesirable in high-voltage applications such as FACTS Controllers. The device selection must
therefore consider all possible fault protection scenarios to decide on the current and voltage
8
Chapter 3
Power Semiconductor Devices
margins as well as redundancy. The thyristor can carry a large overload current for a short periods
and a very large single-cycle current without failures. The thyristor and the diode family of devices
fail in a short circuit with low-voltage drop. So the circuit may continue to operate if the remaining
devices in the circuit can perform the needed function [3].
3.2.2 Losses and speed of switching
Apart from the voltage withstand and current-carrying capabilities, there are many
characteristics that are important to the device. The most important among these are:
• Forward-voltage drop and consequent losses during full conducting state (on-state losses).
• Speed of switching.
• Switching losses.
• The gate-driver.
Serious attention to losses is important for two reasons
• Since they are cost liability to the user, losses are invariably evaluated by utilities and often
by industrial customers on a lifetime present worth basis. When the losses equal 2% for
example and the cost of an FACTS converter is $100 per kilowatt; that means (0.02kW
losses per kW rating), the value of losses for an evaluated value of $2000 kW will be $40.
Therefore, the efficiency of a complete FACTS Controller of several hundred MW rating
needs to be better than the converter valve losses that have to be less than 1%.
• The device losses have to be efficiently removed from inside the wafer to outside the sealed,
high-voltage, insulating package and on to the external cooling medium. For this reason,
packaging and cooling of the device is a formidable challenge to ensure that its wafer
temperature does not exceed the safe operating level, which is about 100°C, with safe
switching characteristics and adequate margin for the overload and short circuit currents.
More often than not, fault current determines the normal useable rating of the devises.
Higher losses mean higher cost of packaging, further losses and cost in disposing the
thermal losses to water or air, as well as the size and weight of the complete equipment [3].
3.2.3 Parameter trade-off of the devices
The cost of the devices is also related to the production yield of good device, which are then
graded into various rating. This therefore calls for good quality control all the way from starting
material to the finished product and including the quality of electric power supply in the production
plant. All power devices of high-power controllers are individually tested, as is the practice with
HVCD converters, and their record is kept for future replacement service. Apart from the trade-off
between the voltage and current capability, other trade-off parameters include:
• Power requirements for the gate.
• di / dt capability.
• dv / dt capability.
• Turn-on time and turn-off time.
• Turn-on and turn-off capability (so-called Safe Operating Area (SOA)).
• Uniformity of characteristics.
• Quality of starting silicon wafers.
• Class of clean environment for manufacturing of devices, etc.
9
Chapter 3
Power Semiconductor Devices
Advanced design and processing methods have been developed and continue to be developed. It
is common for device manufacturers to make the devices for individual large customers and even
individual large project orders, such as HVDC and FACTS projects. The switching speed, the
switching losses, the size, and the cost of snubber circuits and the associated losses, usually
attributed to the power semiconductor devices, largely result from the fact that the devices are sold
separately from gate-driver circuits and from the snubber circuit [4].
3.3 Power device material
Power semiconductor devices are based on high-purity, single-crystal silicon. Single crystal
several meters long and with required diameter (up to 150mm) are grown in the so-called Float
Zone Furnaces. Then, this huge crystal is sliced into wafers to be turned into power devices through
numerous process steps. Pure silicon atoms have four electron bonds per atom with adjacent atoms
in the lattice. It has high resistivety and very high dielectric strength (Over 200kV/cm). Its
resistivety and charge carriers available for conduction can be changed, shaped in layers, and
graded by implementation of specific impurities (doping). With different impurities, levels and
shapes of doping, along with the high technology of photolithography, laser cutting, etching,
insulation, and packaging, large finished devices are produced [4].
3.4 Perspectives on power devices equipment
In the following, some details about power semiconductor devices which are suitable for
FACTS controllers. Generally, FACTS systems are used for the dynamic control of the voltage,
impedance and phase angle of high voltage AC lines. Basically, FACTS controllers depend on an
assembly of AC/DC and/or DC/AC converters and/or high power switches. Those systems depend
on fast and highly reliable power electronic devices (thyristor valves). Using those valves in
FACTS systems and HVDC applications proved their effectiveness in HV transmission systems to
reduce energy transfer limitations. Further development in semiconductors (GTO and IGCT)
allowed new power electronic configurations to be introduced to the tasks of power transmission
and load flow control. The mentioned power semiconductor devices would be connected in series
and /or parallel to achieve its function in FACTS systems. The most common elements, which can
be used in FACTS systems will be discussed in [4].
3.5 Power diodes
The diodes are a family of two-layer devices with unidirectional conduction (see Fig.3-2). A
diode conducts in a forward (conducting) direction from anode to cathode, when its anode is
positive with respect to the cathode. It does not have a gate to control conduction in its forward
direction. The diode blocks conduction in the reverse direction, when its cathode is made positive
with the respect to its anode, (shown in Fig.3-3). The importance of diodes for FACTS Controllers
comes from the possibility that:
• A diode converter can be used as a simple low cost and efficient converter, to supply active
power in a FACTS Controller.
• A diode is connected across each turn-off thyristor in voltage-sourced converters; it is also
connected in intermediate levels in multilevel voltage-sourced converters.
• A diode may be connected in series with each turn-off thyristor for reverse blocking of
voltage.
• Diodes are used in snubber and gate-driver circuits.
10
Chapter 3
Power Semiconductor Devices
Power diodes are made of silicon pn junction with two terminals, anode and cathode. The pn
junction is formed by alloying, diffusion, and epitaxial growth as shown in Fig.3-2(b, c). Modern
techniques in diffusion and epitaxial processes permit the desired device characteristics. The diodes
have the following advantages:
• High mechanical and thermal reliability.
• High peak inverse voltage.
• Low reverse current and low forward voltage drop.
• High efficiency.
• Compactness.
Cathode
Cathode
Cathode
n+
n
n-
P
P+
Anode
Anode
(a)
Anode
(b)
(c)
Fig.3-2 Diode: (a) diode symbol, (b) diode structure, and (c) more detailed diode structure.
A conducting diode will have a small voltage drop across it. A diode is reverse biased when the
cathode is made positive with respect to the anode. When the diode reverse biased, a small reverse
current known as leakage current flows. This leakage current increases with the increase in the
magnitude of a reverse voltage until the avalanche voltage is reached (the breakdown voltage), (see
Fig.3-3) [3], [4].
3.5.1 Dynamic characteristics of power switching diodes
At low frequency and low current, the diode may be assumed to act as a perfect switch and the
dynamic characteristics (turn on and turn off characteristics) are not very important. But at high
frequency and high current, the dynamic characteristics plays an important role because it increases
the power loss and gives a large voltage spikes which may damage the device if proper protection is
not given to the device as shown in Fig.3-4 [4].
I
A
T2
Revserse leakage
Current
K
R
+
T1
V
V _
T1
T2
(a)
(b)
Fig.3-3 (a) Diode forward- and (b) reverse-recovery-biased.
11
Chapter 3
Power Semiconductor Devices
3.5.2 Reverse-recovery characteristic
The reverse recovery characteristic is much more important than forward recovery
characteristics because it adds recovery losses to the forward loss. When the diode is forward
biased, the current is due to the net effects of majority and minority carriers (see Fig.3-4). When the
diode is in the forward conduction mode and then its forward current is reduced to zero (by
applying reverse voltage) the diode continues to conduct due to minority carriers which remains
stored in the pn junction and in the bulk of semi-conductor material. The minority carriers take
some time to recombine with opposite charges to be neutralized. This time is called the reverse
recovery time. The reverse recovery time, trr, is measured from the initial zero crossing of the diode
current to 25% of maximum reverse current Irr. Where, trr has 2 components, t1 and t2. The time t1 is
as a result of charge storage in the depletion region of the junction. It is the time between the zero
crossing and the peak reverse current IRR. t2 is as a result of charge storage in the bulk semiconductor material: trr = t2 + t1 , I RR = t1.(di / dt ) . The reverse recovery time depends on the junction
temperature, rate of fall of forward current and the magnitude of forward current prior to
commutation (turning off). When the diode is in reverse biased condition the flow of leakage
current is due to minority carriers. The application of the forward voltage would force the diode to
carry current in the forward direction. But a certain time known as forward recovery time (turn-on
time) is required before all the majority carriers over the whole junction can contribute to current
flow. Normally the forward recovery time is less than the reverse recovery time. The forward
recovery time limits the rate of rise of the forward current and the switching speed [4].
3.5.3 Power diodes types
Power diodes can be classified as the following:
• General purpose diodes.
• High speed (fast recovery) diodes.
• Schottky diode.
Table 1.1 compares between the main properties of the diode’s type, like the rated voltage and
current, reverse-recovery time, and turn off time [4].
Diode type
General Purpose Diodes
Fast Recovery Diodes
Schottky Diodes
Rated voltage [V].
5000
3000
100
Rated current [A].
3500
1000
300
25
0.1-5
few nanoseconds
Turn off time.
Long
Short
Extremely short
Switching frequency.
Low
High
Very high.
0.7 to 1.2
0.8 to 1.5
0.4 to 0.6
Reverse-recovery time trr [µs].
Forward voltage VF [V].
Table 1.1 the available diodes information’s.
12
Chapter 3
Power Semiconductor Devices
VF
Vi
0
(a)
t1
IF
t
-VR
t1
(b)
Pn-Pno
at Junction 0
IF ≈
IR ≈
t2
t
IRR/4
t
VF I
RL
IO
0
(c)
trr
IRR
t
VR
RL
(e)
V
(d)
0
t2
t
Minority
Carrier
storage, ts
Transition
interval, tt
t1
-VR
Forward
bias
Fig.3-4 Diode characteristics.
(a).Input waveform applied to the diode in Fig.3-3(a), (b) The excess-carrier density at the
junction, (c) The diode current, (d) The diode voltage, (e) Diode reverse-recovery
characteristics.
3.6 Insulated Gate Bipolar Transistor (IGBT)
A modern power transistor is the Insulated Gate Bipolar Transistor (IGBT), It operates as a
transistor with high-voltage and high-current capability and moderate forward voltage drop during
the conduction state. The IGBT has progressed to become a choice in a wide range of low and
medium power applications going up to several megawatts and even tens megawatts. Thus IGBT is
of some importance to FACTS controllers [3], [4].
The IGBT is a voltage controlled device. It has high impedance like a MOSFET and low onstate conduction like BJT. Fig.3-5 shows the basic silicon cross-section of an IGBT. Its construction
is same as power MOSFET except that the n+ layer at the drain in a power MOSFET is replaced by
P+ substrate called collector. The IGBT has three terminals gate (G), collector (C), and emitter (E).
With the collector and the gate voltage positive with respect to the emitter the device is in forward
blocking mode. When the gate to emitter voltage becomes greater than the threshold voltage of
IGBT, an n- channel is formed in the P-region. Now the device is in forward conduction state. In
this state p+ substrate injects holes into the epitaxial n- layer. Increasing in the collator to emitter
13
Chapter 3
Power Semiconductor Devices
voltage will result in increasing of injected holes concentration and finally a forward current
established.
Collector
C
C
p+
n+
n+
n- epi
p
G
n+
Gate
G
Gate
i2
i1
Emitter
E
E
(a)
(b)
(c)
Fig.3-5 IGBT transistor: (a) IGBT transistor structure and the location of the equivalent
circuit, (b) the equivalent circuit, (c) IGBT symbol.
The advantage of the IGBT is its fast turn-on and turn-off because it is more like a majority
carrier (electrons) devices. It can be therefore used in pulse width modulator (PWM) converters
operating at high frequency. On the other hand being a transistor device, it has higher forward drop
voltage compared to thyristor type devices such as GTOs. Nevertheless, the IGBT has become a
workhorse for industrial applications and has reached sizes capable of application in the range of
10MW or more. The transistor devices, such as MOSFETs and IGBTs, potentially have currentlimiting capabilities by controlling the gate voltage. During this current-limiting action, the device
losses are very high, and in high-power applications, current-limiting action can only be used for
very short periods of a microsecond. Yet, this time can be enough to allow other protective actions
to be taken for safe turn-off of the devices. This feature is extremely valuable in voltage-sourced
converters, in which fault current can rapidly rise to high levels due to the presence of a large DC
capacitor across the converter. On the other hand, with fast sensing, combined with the fast turn-off
of the advanced GTOs, an effective turn-off can be achieved within 2-3µs. This method will also
spare the devices from high-power dissipation and sacrifice their useable capacity. The turn-off time
of the conventional GTOs is too long for high-speed protective turn-off. IGBT is coming from a
low-power end, has been pushing out conventional GTO’s rating go up (as available packaged
parallel-IGBTs). This is because the conventional GTO’s have serious disadvantages which are
basically related to the large gate-drive requirements, the slow-switching and the high-switching
losses. The IGBT has its own generic limitations, including: high forward voltage drop,
complexities with providing double-side cooling, the nature of the repetitive MOS on the chip limits
that can be achieved in increased blocking voltage. Also, IGBT production needs much clearer
production facility. A major advantage for IGBT for high-power applications is its low-switching
losses, fast switching, and current-limiting capability. However, with the advanced GTOs, and
MCT’s, which will be discussed later, there is a prospect for major advances for devices suitable for
a wide range of FACTS Controllers. On the other hand, future outcome often depends on the market
forces of volume production, and this is a favor of the IGBTs continuing to push its application to
14
Chapter 3
Power Semiconductor Devices
higher power levels. Fig.3-6 shows the output characteristic of the collector current IC versus
collector to emitter voltage VCE for given value of gate to emitter voltage VGE [3], [4].
IC
VGE
VCE
VCE(sat)
Fig.3-6 IGBT current-voltage (IC-VGE) characteristics for a given value of VCE.
3.6.1 Switching characteristic of the IGBT
The switching process may be seen as a combination of switching performance of a MOSFET
and BJT which did not discussed here. Fig.3-7 shows the switching characteristic of an IGBT. The
turn-on time consists of a delay time td(on) and a rise time tr. The turn on delay time is the time
required by the leakage current ICE to rise to 0.1IC, where IC is the final value of the collector
current.
The rise time is the time required for the collector current to rise from 0.1IC to its final value IC.
After turn-on, the collector-emitter voltage VCE will be very small during the steady state
conduction of the device [3]. The turn-off time consists of the delay off time td(off) and fall time tf.
The off time delay is the time during which the collector current falls from IC to 0.9IC and VGE falls
to threshold voltage VGET. During the fall time tf the collector current falls from 0.90IC to 0.1IC.
During the turn-off time the interval collector-emitter voltage rises to its final value VCE. IGBT’s are
voltage controlled power transistors. They are faster than BJT’s, but still not quite as fast as
MOSFET’s. The IGBT’s offer superior drive and output characteristics compared to BJT’s. IGBT’s
are suitable for high voltage; high currents and frequencies up to 20KHz. IGBTs are available up to
1400V, 600Amps and 1200V, 1000Amps [4].
3.7 Types of high-power thyristor devices
Technically, the terms “thyristor” and “Silicon Controlled Rectifier (SCR)” are applied to a
basic family of four-layer controlled semiconductors devices in which turn-on and turn-off depends
on pnpn regenerative feedback. The name Silicon Controlled Rectifier (SCR) was given by
inventors and commercially pioneered by GE (General Electric). In the text of a device which has a
turn-on but no turn-off capability, the term SCR was later changed by others to thyristors. With the
emergence of a device with both turn-on and turn-off capability, named Gate Turn-off Thyristors
referred to as GTO, the device with just the turn-on capability began to be referred to as
“conventional thyristor” or just “thyristor”. Other members of the thyristor or SCR family have
acquired other names based on acronyms. In use of the term thyristor is generally meant to be the
conventional thyristor.
15
Chapter 3
Power Semiconductor Devices
VGE
VGET
td(on)
tr
td(off)
tf
t
VCE
0.9 VCE
t( on ) = td ( on ) + tr
toff = td ( off ) + t f
0.1 VCE
t
IC
0.9 ICE
0.1 ICE
t
td(off)
tf
Fig.3-7 Switching characteristic of an IGBT.
The thyristor starts to conduct in forward direction when a trigger current pulse is passed from
gate to cathode, and rapidly latches into full conduction with a low forward voltage drop (1.5V to
3V depending on the type of the thyristor and the current). As mentioned previously, the
conventional thyristor cannot force its current back to zero. Instead, it relies on the current itself for
the current comes to zero. When the circuit current comes to zero, the thyristor recovers in a few
microseconds of reverse backing voltage, following which it can block the forward voltage until the
next turn-on pulse is applied. Because of their low cost, high frequency, ruggedness, and high
voltage and high current capability, conventional thyristors are extensively used when circuit
configuration and cost-effective application do not call for turn-off capability. Often the turn-off
capability does not offer sufficient benefits to justify higher cost and losses of the devices. The
conventional thyristor has been the device of choice for most HVDC projects, some FACTS
controllers, and a large percentage of industrial applications. It is often referred to as the workhorse
of the power electronics business. The several versions of thyristors with turn-on capability among
these and relevant to the FACTS technology are presented in the following sections [3].
3.7.1 Thyristors
The thyristor are a family of four-layer devices. A thyristor latches into full conduction in its
forward direction when one of its electrodes (anode) is positive with the respect to its other
16
Chapter 3
Power Semiconductor Devices
electrodes (cathode) and turn-on voltage or current signal (pulse) is applied to the third electrode
(gate) (see Fig.3-8(a)). Latched conduction is a key to low on-state conduction losses, called base.
Most thyristors are designed without gate-controlled turn-off capability, in which case the thyristor
recovers from its latched conducting state to a non-conducting state only when the current brought
to zero by other means. Other thyristors are designed to have both gate-controlled turn-on and turnoff capability.
The thyristor may be designed to block in both the forward and reverse direction (referred as a
symmetrical device) or it may be designed to block only in the forward direction (referred to as
asymmetrical device). Thyristors are the most important devices for FACTS Controllers. Compared
to thyristors, transistors generally have superior switching performance, in terms of faster switching
and lower switching losses. On the other hand, thyristors have lower on-state conduction losses and
higher power handling capability than transistors. Advances are continuously being made to achieve
the best of both, i.e., low on-state losses, while increasing their power handling capability.The
Thyristor (SCR Silicon Controlled Rrectifier), which is shown in Fig.3-8, is a three-junction, fourlayer device. The thyristor is a unidirectional switch, which once turned on by a trigger pulse,
latches into conduction with the lowest forward voltage drop of 1.5V to 3V at its continuous rated
current. It does not have the capability to turn off the current, so that it recovers its turned-off state
only when the external circuit causes the current to come to zero. The thyristor is referred to as the
workhorse of power electronics. In a many applications, turn-off capabilityy is not necessary.
Without turn-off capability, the resulting device can have higher voltage and/or rating, cost less than
one-half, require a simple device control circuit, has lower losses, etc, compared to device with
turn-off capability. Therefore, the choice in favor of a more expensive and higher loss device with
turn-off capability will occur when there is a decisive application advantage [3], [4].
As shown in Fig.3-8(c), the thyristor is equivalent to the integration of two transistors, pnp
and npn. When a positive gate trigger is applied to the p gate of the upper npn transistor with
respect to the n+ emitter (cathode), it starts conducting. The current through the npn transistor
becomes the gate current of the pnp transistor as shown by the arrows, causing it conducts as well.
The current through this pnp transistor in turn becomes the gate current of the npn transistor giving
a regenerative effect to latched conduction with low forward voltage drop with the current flow
essentially limiting the external circuit. What is important is that due to the internal regenerative
action into saturation, once the thyristor is turned on, the internal n+ and n layers become saturated
with electrons and holes and act like a short circuit in the forward direction.
The whole device behaves like a single pn junction device (a diode). Thus, its forward on-state
voltage drop corresponds to only one junction (even though it has three junction) compared to two
junction in transistor devices such as MOSFET and IGBT. The turn-off time, which can be a few
tens of microseconds, depends on the reverse voltage after zero current and has to be carefully
considered for specific applications. This turn-off time must elapse before any positive voltage can
safely be applied [3], [4].
In large thyristor wafers, the gate structure is brought out through the cathode side at the top.
Several amplifying stages are provided in concentric circles at the centre in order to decrease the
required external gate pulse current. It is essential to rapidly spread out turn-on current over the
17
Chapter 3
Power Semiconductor Devices
whole device. It is also appropriate to consider adding another high-voltage, very low current
external pilot thyristor in order to increase the gain and reduce the gate turn-on power at the
thyristor level. Such a device would be expensive because of the very low current rating. A thyristor
can also be turned off by hitting the gate region with light of appropriate bandwidth. The direct light
trigger thyristor allows the triggering of the thyristor directly from the control circuit via optical
fiber. As an alternative, the external pilot thyristor, which is mentioned above, may be a light
triggered thyristor with the main thyristor as an electrically triggered thyristor.
Cathode
Cathode
Cathode
Gate
Gate
n
Turn-on
P
Cathode
Gate
n+
n+
P
P
P
n
Gate
n
P
n
n
n
P
P
P
Anode
Anode
Anode
Anode
(a)
(b)
(c)
(d)
Fig.3-8 Thyristor (a) Thyristor symbol, (b) Thyristor structure, (d) two-transistor structure,
and (d) Thyristor equivalent circuit.
The application of a positive anode to the cathode voltage with high rate of rise (dv/dt) can also
turn on the device. This happens because the capacitive coupling of the cathode to gate and the high
dv/dt causes just enough current to turn the device on. This is not a safe way to turn on a thyristor.
Turning-on a thyristor in such away can occur at weak spot, which does not spread rapidly and may
damage the device. Unsafe turn-on will also occur if the forward voltage is too high, which creates
charge carriers in weak spot through acceleration of internal charge carriers. This also suggests that
the device can be made with deliberately designed weak spot from where safe turn-on can be
designed into the device. Such devices with self-protection and optional triggering have been
introduced in recent HVDC projects [3], [4], and [5].
Another important aspect is that when a turn-on pulse is applied, there has to be enough anodeto-cathode forward voltage, or rate of rise of voltage, to cause a rapid turn-on. Insufficient voltage
can lead to soft turn-on with device voltage falling slowly while the current is rising. This can lead
to a high turn-on loss in certain areas of the device and cause a possible damage. Depending on the
application, the device has to be designed for the specific minimum turn-on voltage and the turn-on
pulse is blocked if the forward voltage is inadequate. At high temperature, the thyristor has a
negative temperature coefficient. Thus, it has to be designed to ensure a uniform internal turn-on
and turn-off. Being a high voltage device, it includes doping-based carriers as well as a large
number of intrinsic carriers. With higher temperature, the number of thermal carriers and hence
total carriers increases and this leads to a lower forward voltage drop. Once a thyristor is turned on
there is a need to sustain a minimum anode-cathode current for the device to stay turned on. This
18
Chapter 3
Power Semiconductor Devices
minimum current is usually a percentage of the device current. The gate-drive is usually arranged to
send another turn-on pulse as needed. Generally thyristors have a large overload capability. They
have two times normal over-current capability for several seconds, ten times for several cycles, and
50 times fully short-circuit for one cycle [5], [6].
3.7.1.1 Switching characteristic of thyristor (SCR)
The Switching operation of an SCR is shown in Fig.3-9. Some of its important features are:
• Initially when forward voltage is applied across the device, the off state or static dv/dt has to
be limited so that the device does not turn on.
• When gate current IG is applied (with anode in forward blocking state), there is a finite delay
time before the anode current starts building up. This delay time, td, is usually a fraction of
microseconds.
• After the delay time, the device conducts and the anode current builds up to the full value IT.
The rate of rise of the anode current during this time depends upon the external load current.
If during turn-on, the anode current builds up too fast which may be damage the device. The
initial turn-on of the device occurs near the gate cathode periphery and then the turn-on area
of device spreads across the entire junction with a finite velocity. If IT rises at a rate faster
than the spreading velocity, then the entire current IT is confined to a small area of the device
eventually overheats the junction and may destroy the device. Therefore, it is necessary to
limit the turn-on di/dt of the circuit to less than the safe di/dt that can be tolerated by the
device.
IG
t
Commutating di/dt
IT
Turn on
di/dt
IRM
VAK
Off state dv/dt
t
VDRM
tq
Reapplied dv/dt
t
td
tr
VAK
VRRM
VR
Fig.3-9 Switching characteristics of the thyristor.
• During conduction, the middle junction is heavily saturated with minority carries and the gate
has no further control on the device. The device drop voltage under this condition is typically
about 1V.
19
Chapter 3
Power Semiconductor Devices
• From the conducting state, the SCR can be turned off by temporarily applying a negative
voltage across the device from external circuit. When reverse voltage is applied, the forward
current first goes to zero and then the current builds up in the reverse direction with the
commutation di/dt. The commutation di/dt depends on the external commutating circuit. The
reverse current flow across the device to sweep the minority carries across the junction. At
maximum reverse recovery current IRM, the junction begins to block causing decay of reverse
current. The fast decay of the recovery current causes a voltage overshoot VRRM across the
device due to the leakage inductance effect. At zero current, the middle junction is still
forward biased and the minority carries in the vicinity must be given time for recombination.
The reapplied dv/dt has to be limited so that no spurious turn-on occurs. The device turn-off
time, tq, is a function of Tj (the temperature of the junction), IT, VR, VDRM, di/dt, dv/dt and VG.
Thyristors are available for power electronics system application with voltage ratings up to
6000V and current ratings about 6000Amps [4].
3.7.2 Gate turn-off thyristor (GTO)
Invented at GE, is now referred to as a GTO thyristor or simply a GTO. Like a conventional
thyristor, it turns on in a fully conducting mode (latched mode) with a low forward voltage drop
when a turn-on current pulse is applied to its gate with respect to its cathode. Like a conventional
thyristor, the GTO will turn off when the current naturally comes to zero, but the GTO also has
Turn-off capability when a turn-off pulse is applied to the gate in reverse direction. With an
adequate turn-off pulse, The GTO rapidly turns off and recovers to withstand the forward voltage
and be ready for the next turn-on pulse. The GTO is a widely used device for FACTS Controllers.
However, because of its bulky gate drivers, and slow turn-off and costly snubbers, it is likely to be
replaced in the coming years by more advanced GTOs Thyristors. These Advanced GTOs, which in
turn are part of the thyristor family, are explained later on. Basically, the gate turn-off (GTO)
thyristor is similar to the conventional thyristor and essentially most of the aspects discussed in the
last section apply to GTOs as well. The GTO, which is shown in Fig.3-10, like the thyristor is a
latched-on device, but it is also a latched-off device. Discussion of the GTO in this section refers to
conventional GTO without the recent advances made in devices made under different acronyms
which are discussed in later sections. Considering the equivalent circuit shown in Fig.3-10(d) which
is the same as the circuit shown in Fig.3-8(c) of the thyristor except that the turn-off has been added
between the gate and the cathode in parallel with the gate turn-on (shown only by arrows in the
equivalent circuit). If large pulse current is passed from the cathode to the gate to take away
sufficient charge carriers from the cathode, i.e., from the emitter of the upper pnp transistor, the npn
transistor will be drawn out of the regenerative action.
As the upper transistor turns off, the lower transistor is left with an open gate, and the device
returns to non-conducting state. However, the required gate current for turn-off is quite large.
Whereas the gate current pulse required for turn-on may be 3-5 %, i.e., 30 A for only 10 µsec for a
1000Amps devices, the gate current required for turn-off would be more like 30-50%, i.e., 300 A or
larger for 20-50µsec [3], [4]. The voltage required to drive the high current pulse is low (about 1020V) and being a pulse of 20-50µsec duration and the energy required for turn-off is very large. Yet
the losses are large enough to be a significant economic liability in terms of losses and cooling
requirements, when considering the number of valve turn-off events in a converter. Turn-off energy
20
Chapter 3
Power Semiconductor Devices
required is 10 to 20 times that is required for GTO turn-on, and the GTO turn-on required energy is
10 to 20 times that required for a thyristor.
Gate
Cathode
Cathode
Turn-off
Turn-on
n
Cathode
n+
Gate
P
P
Gate
n
n
n+
P
P
n
P
Anode
Anode
(a)
(b)
Anode
(c)
Fig.3-10 Gate turn-off thyristor (a) GTO -symbol, (b) -structure and (c) -equivalent circuit.
The cost and size of the turn-off circuits for GTO are comparable to the device costs itself.
Another consideration is that the turn-off has to be uniformly effective over the entire device.
Whereas in a thyristor, there is one cathode with a single gate structure spread out across the device,
a successful GTO turn-off requires dividing up the cathode into several thousand islands with a
common gate-line which surrounds each and every cathode islands in Fig.3-11. Thus, a GTO
consists of a large number of thyristor cathodes with a common gate, a drift region, and an anode.
Given the complex structure, state-of-the-art GTOs do not have built-in amplifying gates.
Consequently, the total available area on the device for the cathode decreases to about 50%
compared to a thyristor. Therefore, GTOs forward voltage drop is about 50% higher than that of a
thyristor but still 50% lower than that of a (IGBT) of the same rating [5].
(a)
(b)
Fig.3-11 (a) A picture of GTO surface (b) A picture of GTO wafer including definition of radii
rv.
The general process of making GTOs is about the same as that for thyristors, although due to
complications of the cathode and the grid distribution, its process requires a cleaner room, yield
21
Chapter 3
Power Semiconductor Devices
may be less, and cost perhaps twice that of thyristor for the same converter ratings. As for a
thyristor, there are trade-offs between voltage, current, di/dt, dv/dt, switching times, forward losses,
switching losses, etc., for the GTO design. Large sector of the market of GTOs is for voltagesourced converters in which a fast recovery diode is connected in reverse across each GTO which
means that GTOs do not need reverse voltage capability. This also provides beneficial tread-offs for
other parameters, particularly the voltage drop and higher voltage and current ratings. This is
achieved by the so-called buffer layer, a heavily doped n+ layer at the end of the n- layer. Such
GTOs are known as asymmetric GTOs. Like a thyristor, the continuous operating junction
temperature limit is about 100oC, after making allowance for the fault current requirements. Like a
thyristor, a GTO is capable of surviving a high, short-time over current (10 times for one offset
cycle) as long as it is not required to turn off that current. Failure mechanisms are also similar and
the edge requires appropriate contouring to reduce voltage stress and passivation to avoid a
flashover around the edge [7], [8]. In a thyristor, the current zero is brought about by the external
system. The voltage across the device automatically becomes negative immediately after the current
zero. On the other hand, the GTO is turned off while the circuit is driving in the forward direction.
Therefore, for successful turn-off it is necessary to reduce the rate-of-rise of forward voltage with
the help of a damping circuit. In a GTO, the anode side pn- junction is lightly doped and designed
to support almost all of the blocking voltage, essentially on the n-side. On the other hand, the
cathode side pn junction is heavily doped on both sides and the breakdown voltage may be about
20V [5], [6].
3.7.2.1 Switching characteristics of GTO (turn-on and turn-off process)
Apart from the gate driver power, GTOs also have high switching losses and it is important to
appreciate the turn-on and turn-off process with associated device stresses and losses. Fig.3-12
shows simplified waveforms for the turn-on and the turn-off processes. For turn-on, a 10µs current
pulse of greater than 5% of the load current with a fast rise time limited largely by the gate circuit
inductance is applied from the gate to cathode. However, there is a delay of microseconds, before
the anode-cathode current begins to rise and the voltage begins to fall.
The current rises at the rate limited by the circuit as required for safe turn-on of the device such
that all cathode islands turn on evenly. Also, given the circuit topology of the Voltage-Sourced
Converter, the GTO turn-on is accompanied by turning off the reverse-conducting diode in another
valve of the same phase. Therefore the GTO has to turn-on the main circuit current pulse a large
reverse leakage current of the diode. During this process of rising current, the anode-cathode
voltage falls slowly in accordance to the plasma spreading time, ultimately to its on-state lowvoltage level. Following the full turn-on, it is necessary to maintain some gate current of about
0.5% to ensure that the gate does not unlatch; this current is known as back -porch. GTO turn-on
losses result from simultaneous existence of voltage and current, made more difficult by current
overshoot corresponding to the reverse current of a diode, mentioned above.
The turn-off process in the GTO is initiated by a negative gate current. Due to the high
conductivity of the p-base, holes arriving from the anode partially flow to the negatively polarized
gate contact. During the storage time, the current progressively filaments towards the middle of the
cathode segments, until the filaments are finally “pinched off”. At this instant, the anode current
falls rapidly, and both the pn+ and pn junctions are again able to sustain voltage.
22
Chapter 3
Power Semiconductor Devices
Turn off gate pulse
Turn on
gate pulse
v
i
v
i
Turn off
Turn on
(a)
(b)
Fig.3-12 GTO turn-on and turn-off process: (a) turn-on and (b) turn-off.
The filamentation of the current, towards the cathode centers, reduces the active silicon area
during the critical turn-off phase. This would be a minor limitation where it is not compounded by
the filamentating current tending to commutate to those cathode areas remotely located from the
extinguishing gate current. This re-distribution of the cathode current continues during the storage
time (tens of microseconds) and culminates with a rising anode voltage and a falling cathode
current. It is this phase which requires the presence of a snubber (i.e. a capacitance) across the
device to limit the reapplied dv/dt to between 500 and 1000 V/ms. Fig.3-13 illustrates this short
phase in which both anode voltage and cathode current coexist with the danger of re-triggering that
this represents [4], [5], [6], and [8]. The GTOs principle handicap compared to IGBT discussed
before has been its large gate turn-off drive requirements. This in turn results in long turn-off time,
lower di/dt and dv/dt capability, and therefore costly turn-on and turn-off snubber circuits adding to
the cost and losses. Because of its slow turn-off, the GTO can be operated in PWM converters at a
relatively low frequency (up to a few hundred Hz), which is, however, sufficient for high-power
converters. On the other hand, it has lower forward voltage drop and is available in larger rating
than IGBT. GTO has been utilized in FACTS controller of several hundreds of MWs [5], [6], and
[7].
3.7.3 MOS Turn-Off Thyristor
The Silicon Power Corporation (SPCO) has developed the MTO Thyristor, which is a
combination of a GTO and MOSFETs. Together they overcome the limitation of GTO regarding its
gate-driver power, snubber circuit, and dv/dt limitation. Unlike the IGBT (discussed before), the
MTO structure is not implanted on the entire of the device surface, but instead the MOSFETs are
located on the silicon all around the GTO to eliminate the need for high-current GTO turn-off pulse.
The GTO structure is essentially retained for advantages of high voltage (up to 10kV), high current
(up to 4kA), and lower forward conduction losses than IGBTs.
23
Chapter 3
Power Semiconductor Devices
Conducting thyristor
GTO zone
Blocking transistor
Fig.3-13 Conduction and conventional Turn-off of a GTO.
With the help of these MOSFETs and tight packaging which minimize the stray inductance in
the gate-cathode loop, the MTO becomes significantly more efficient than the conventional GTO,
requiring drastically smaller gate drivers while reducing the charge storage time on turn-off,
providing improved performance and reduction of system costs. As before, the GTO is still
provided with double-sided cooling and lends itself to thin packaging technology for even more
efficient removal of heat from the GTO. Fig.3-14 shows the symbol, structure, and equivalent
circuit of MOS turn-on thyristor. The turn-off in MTO can be much faster; 1-2µs and the losses
corresponding to the storage time are almost eliminated, this also means high dv/dt, and much
smaller snubber capacitors and elimination of the snubber resistor [3], [4].
3.7.4 Emitter Turn-on Thyristor (ETO)
Similarly, MTO, ETO is another variation of exploiting the virtues of both the thyristor and the
transistor, i.e., GTO and MOSFET. ETO was invented at Virginia Power Electronics Centre in
collaboration with SPCO. The EMO symbol and equivalent circuit are shown in Fig.3-15. As
shown in Fig.3-15(b), a MOSFET T1 is connected in series with the GTO and a second MOSFET T2
is connected across this series MOSFET and the GTO gate. Actually T1 consists of several NMOSFETs and T2 consists of several P-MOSFETs packaged around the GTO in order to minimize
the inductance between the MOSFETs and the gate cathode of the GTO. N- and p-MOSFETs and
GTOs were commercially available devices made in large quantities [3].
The ETO has two gates: one is the GTOs own gate used for turn-on and the other is in series
MOSFET gate used for turn-off. When the turn-off voltage signal is applied to the n-MOSFET, it
turns off the transfer of all the current away from the cathode (n emitter of the upper transistor of
the GTO) into the base via MOSFET T2, thus stopping the regenerative latched state and a fast turn24
Chapter 3
Power Semiconductor Devices
off. It is important to note that the MOSFETs see high voltage, no matter how high the ETO
voltage. T2 is connected with its gate shorted to its drain and hence voltage across it’s clamped at a
value slightly higher than its threshold voltage and the maximum voltage across T1 cannot exceed
that across T2 [3], [7], and [8].
Cathode
Cathode
Turn-on
gate
Cathode
Turn-off
gate FET
Turn-on
gate
Cathode
FET
Turn-off
FET
Turn-off
n
n+
Turn-off
gate
Turn-on
P
Turn-on
P
n
n
P
n+
P
n
Anode
P
Anode
Anode
Anode
(a)
(b)
(c)
(d)
Fig.3-14 MOS Turn-off (MTO) thyristor (a) MTO symbol, (b) MTO structure; (c) MTO
equivalent circuit, and (d) more detailed equivalent circuit.
Cathode
Cathode
N-MOSFET
Turn-off
T1
P-MOSFET
Turn-off
T2
Turn-on
Turn-on
Anode
Anode
(a)
(b)
(c)
Fig.3-15 Emitter Turn-off (ETO) thyristor: (a) ETO symbol, (b) ETO equivalent circuit, and (c)
ETO structure.
3.7.5 Integrated Gate-Commutated Thyristor (GCT and IGCT)
The gate-commutated thyristor (GCT) is a hard-switched GTO involving very fast and large
current pulse, as large as the full rated current, which draws out all the current from the cathode into
the gate in 1µsec to ensure a fast turn-off. Its structure and its equivalent circuit are the same as that
of a GTO. The IGCT is a device with added value on GCT; including a multilayered printed circuit
25
Chapter 3
Power Semiconductor Devices
board gate drive supplied with the main device, and may also include a reverse diode, as shown in
the structured diagram in Fig.3-16. In order to apply a fast-rising and high-gate current, GCT
(IGCT) incorporates a special effort to reduce the inductance of the gate circuit (gate-driver-gatecathode loop) to the lowest possible value, as required also for MTO and ETO to extent possible.
Essentially, the key to GCT (IGCT) is to a very fast gate drive and this is achieved by a coaxial
cathode-gate feed through and multilayered gate-driver circuit boards, which enable the gate current
to rise at 4kA/µs with a gate-cathode voltage of 20V. 1µs, the GTOs upper transistor is totally
turned off and the lower npn transistor is effectively left with an open base turn-off. Being a very
short duration pulse, the gate-drive energy is greatly reduced. Also, by avoiding the gate overdrive,
the gate energy consumption is minimized [3].
Cathode
Gate
A(Anode)
n+
Ia
P
P
+
GTO
Diode
Vak
Ig
n-
-
n
P+
n+
Anode
K(Cathode)
(a)
(b)
Fig.3-16 IGCT thyristor (a) IGCT symbol (b) IGCT structure with a Gate-Commutated
Thyristor and reverse diode.
3.7.6 MOS-Controlled Thyristor (MCT)
An MOS controlled thyristor (MCT) incorporates a MOSFET-like structure in the device for
both the turn-on and turn-off. Fig.3-17 shows an n-type MCT. The equivalent circuit for the n-MCT
shows that for turn-on there is an n-type MOSFET (shown as n-FET) connected across the cathode
side transistor, similar to that for an IGBT. Another p-type MOSFET (shown as p-FET) is
connected across the gate cathode of the cathode side npn transistor for turn-off, similar to that for
an MTO. An n-FET is turned on with the application of positive voltage to the gate in respect to the
cathode, the current flows from the anode to the base of the lower transistor, which turns on and
leads to latched turn-on of the thyristor. As shown in Fig.3-17(c), the same gate voltage is also
applied to the base of the p-FET, which ensures that the p-FET stays off. When the gate voltage is
made negative, it turns off the n-FET and turns on the p-FET. The p-FET thereby bypasses the gate
cathode and thus unlatching the thyristor.
The MOS structure is spread across the entire surface of the device giving a fast turn-on and
turn-off with low-switching losses. The power/energy required for turn-on and turn-off is very
small, and so is the delay time (storage time). Furthermore, being a latching device, it has a low onstate voltage drop as for a thyristor. Its processing technology is essentially the same as the IGBT.
The key advantage for the MCT compared to other turn-off thyristor is that it brings distributed
26
Chapter 3
Power Semiconductor Devices
MOS gates for both turn-on and turn-off, very close to the distributed cathodes, resulting in fastswitching and low switching losses for a thyristor device. Therefore, the MCT represents the nearultimate turn-off thyristor with low on-state and switching losses, and the fast-switching device
needed for high-power advanced converters with active filtering capability [3].
Gate
Cathode
Cathode
Gate
P+
P+
n n+
n
P
n
SiO
Gate
SiO
Cathode
P-MOSFET
N-MOSFET
P+
P
n
Anode
Anode
P
Anode
(a)
(b)
(c)
Fig.3-17 MOS-Controlled Thyristor (MCT) (a) MCT symbol, (b) MCT structure (c) MCT
equivalent circuit.
3.8 Control characteristic of power devices
The power semiconductor switching devices can be operated as switches, e.g., by applying
control signals to the gate terminal of thyristors (or to the base of bi-polar transistors). The required
output is obtained by varying the conduction time of the switching devices. Fig.3-18 below shows
the output voltages and the control characteristics of commonly used power semiconductor
switching devices. Once a thyristor is in a conduction mode, the gate signal of either positive or
negative magnitude has no effect. When a power semiconductor switching device is in the normal
conduction mode, there is a small drop across the device. In the output voltage waveforms shown,
these voltage drops are considered negligible. According to that, the power semiconductor
switching devices can be classified based in the following:
•
•
•
Uncontrolled turn-on and turn-off (e.g. diode).
Controlled turn-on and uncontrolled turn-off (e.g. SCR).
Controlled turn-on and off characteristics (e.g. BJT, MOSFET, GTO, SITH (Static Induction
Thyristor), IGBT, SIT, and MCT).
• Continuous gate signal requirements (e.g. BJT, MOSFET, IGBT and SIT (Static Induction
Transistor)).
• Pulse gate requirement (e.g. SCR, GTO and MCT).
• Bipolar voltage withstanding capability (e.g. SCR and GTO).
• Uni-polar voltage withstanding capability (e.g. BIT, MOSFET, GTO, IGBT, and MCT).
• Bidirectional current capability (e.g. Triac and RCT).
27
Chapter 3
•
Power Semiconductor Devices
Unidirectional current capability (e.g. SCR, GTO, BJT, MOSFET, MCT, IGBT, SITH, SIT,
and diode) [4].
(a) Thyristor.
(b) GTO/MTO/ETO/IGCT/MCT/SITH (for MCT, the polarity of vG is reversed as shown).
(c) Transistor.
(D) MOSFET/IGBT switch.
Fig.3-18 Control characteristics of power switching devices.
28
F ATCS concept and general system considerations
Chapter 4
4 FACTS concept and general system considerations
Most of the world electric power supply systems are widely interconnected; involving
connections inside utilities own territories which extend to inter-utility interconnections and then to
inter-regional and international connections. This is done for economical reasons, to reduce the cost
of electricity and improve the reliability of the power system.
4.1 Need of transmission interconnection
Interconnections are needed because they are part from delivery. The purpose of the
transmission network is to pool power plants and load centers in order to minimize the necessary
total power generation capacity and fuel costs. Transmission interconnections enable taking
advantage of diversity of load, availability of source, and fuel price in order to supply electricity to
the loads at minimum cost with a required reliability. In general, if the power delivery system was
made up of radial lines form individual local generators without being part of a grid system, many
more generation resource would be needed to serve the load with the same reliability and the cost of
the electricity would be much higher. With that perspective, transmission is often an alternative to
new generation resource. The power systems of today are mostly mechanically controlled. There is
widespread use of microelectronics, computers and high speed communication for control and
protection of present transmission systems. However, when operating signals are sent to power
circuits, where the final power control action taken, the switching devices are mechanical and there
is little high-speed control. Another problem with mechanical devices is that control cannot be
initiated frequently, because these mechanical devices tend to wear out very quickly compared to
static devices. As a result from the point of view of both dynamic and steady-state operation, the
system is really uncontrollable. Power system planners, operators, and engineers have learned to
live with this limitation by using a variety of ingenious techniques to make the system work
effectively, at the price of providing greater operating margins and redundancies. These represent
an asset that can be effectively utilized with prudent use of FACTS technology (Flexible AC
Transmission Systems) selectivity. In the recent years, greater demands have been placed on the
transmission network and these demands will continue to increase because of the increasing number
of nonutility generators and heightened competition among the utilities themselves. Increased
demands on transmission, absence of long-term planning, and the need to provide open access for
generating companies and customers, all together have created tendencies toward less security and
reduced quality of supply. The FACTS technology is essential to alleviate perhaps some but not all
of these difficulties are enabling utilities to get the most service from their transmission facilities
and to enhance the grid reliability. It is worth mentioning, however, that for many of the capacity
expansion, building new lines or upgrading current and voltage capability of existing lines and
corridors will be necessary [3].
4.2 Opportunities of FACTS
The most interesting idea for transmission planners is that FACTS technology opens up the
opportunities for controlling power and enhancing the usable capacity of present, as well as new
and upgrade lines. The possibility that the current through a line can be controlled at reasonable
costs enables a large potential of increasing the capacity of existing lines with larger conductors,
29
F ATCS concept and general system considerations
Chapter 4
and use of one of the FACTS Controllers to enable corresponding power to flow through such lines
under normal and contingency conditions. These opportunities arise through the ability of the
FACTS Control to control the interrelated parameters that govern the operation of transmission
systems including series impedance, shunt impedance, current, voltage, phase angle, and the
damping of oscillations at various frequencies below the rated frequency. These constraints cannot
be overcome while maintaining the required system reliability, by mechanical means without
lowering the useable transmission capacity. By proving added flexibility, FACTS controllers can
enable a line to carry power closer to its thermal rating. Mechanical switching needs to be
supplemented by rapid-response power electronics. It must be emphasized that FACTS is an
enabling technology and not one-one-one substitute for mechanical switches. The FACTS
technology is not a single high-power controller, but rather a collection of controllers which can be
applied individually or in coordination with others to control one or more of the interrelated system
parameters mentioned previously. A well-chosen FATCS Controller can overcome the specific
limitations of a designated transmission line or corridor. Because all FACTS Controller represent
applications of the same basic technology, their production can eventually take advantage of
technologies of scale. Just as the transistor is the basic element for a whole variety of
microelectronics chips and circuit, the thyristor or high power transistor is the basic element of a
variety of high-power electrical controllers.
FACTS Controller also lent itself to extend usable transmission limits in a step-by-step manner
with incremental investment as and when required. A planner could foresee a progressive scenario
of mechanical switching means and enabling FACTS Controller such that the transmission lines
will involve a combination of mechanical and FACTS Controller to achieve the objective in an
appropriate, staged investment scenario. The unique aspect of the FACTS technology is that this
umbrella concept revealed the large potential opportunity for the power electronics technology to
greatly enhance the value of the power system, and thereby unleashed an array of new and
advanced ideas to make it a reality [3], [9].
4.3 Basic types of FACTS controller
In general, FACTS Controller can be divided into the categories:
• Series Controller.
• Shunt Controller.
• Combined series-series Controllers.
• Combined series-shunt Controllers.
Fig.4-1(a) shows the general symbol for a FACTS Controller: a thyristor arrow inside a box.
4.3.1 Series Controller:
The series controller could have variable impedance, such as a capacitor, a reactor… etc or a
power electronics based variable source of main frequency, subsynchronous and harmonic
frequencies (or a combination) to serve the desired need. In principle, all series controller inject a
voltage in series with the line. Even variable impedance multiplied by current flowing through it,
represents an injected series voltage in the line. As long as the voltage is in phase quadrature with
the line current, the series controller supplies or consumes variable reactive power. Any other phase
relationship will involve handling of real power as shown in Fig.4-1(b).
30
F ATCS concept and general system considerations
Chapter 4
4.3.2 Shunt controller
As in the case of series controllers, shunt controllers may have variable impedance, variable
source, or a combination of these. In principle, all shunt controllers inject current into the system at
the point or connection (see Fig.4-1 (c)). Even variable shunt impedances connected to the line
voltage causes a variable current flow and hence represents injection of current into the line. As
long as the injected current is in phase quadrature with the line voltage, the shunt controllers only
supply or consume variable reactive power. Any other phase relationship will involve handling of
real power as well [3].
4.3.3 Combined series-series controller
This could be a combination of a separate series controller which is controlled in a coordinated
manner in a multi-line transmission system. It could also be a unified controller as shown in Fig.4-1
(d), in which the series controller provides independent series reactive compensation for each line
but also transfers real power among the lines via the power link. The real power transfer capability
of the unified series-series controller, referred to as Interline Power Flow Controller (IPFC), makes
it possible to balance both the real and active power flow in the lines and thereby maximizes the
utilization of the transmission system. The term unified means here that the DC terminals of all
controller converters are all connected together for real power transfer.
4.3.4 Combined series-shunt controller
Combined series-shunt controller could be a combination of a separate shunt and series
controller as shown in Fig.4-1(e), which are connected in a coordinated manner (see Fig.4-1 (e)), or
a Unified Power Flow Controller (UPFC) with series and shunt elements as shown in Fig.4-1 (f). In
principle, combined shunt and series controllers inject current into the system with the shunt part of
the controller and voltage in series in the line with the series part of the controller. However, when
shunt and series controllers are unified, there can be a real power exchange between the series and
the shunt controllers via the power link [3], [9].
4.4 Relative importance of different types of controllers
It is important to appreciate that the series-connected controller impacts the driving voltage and
hence the current and the power flow directly. Therefore, if the purpose of the application is to
control the current/power flow and damp oscillations, the series controller for a given MVA size is
several times more powerful than shunt controller. As mentioned previously, the shunt controller
acts like a current source, which draws or injects the current into the line. The shunt controller is
therefore a good way to control the voltage around the point of connection through injection of
reactive current (leading or lagging), alone or a combination of active and reactive current for more
effective voltage control and damping of voltage oscillation. This is not to say that the series
controller cannot be used to keep the line voltage within the specified voltage range. After all, the
voltage fluctuations are largely a consequence of the voltage drop in the series impedance of lines,
transformers, and generators. Therefore, adding or subtracting the FACTS controller voltage in
series (main frequency, sub-synchronous or harmonic voltage and a combination thereof) can be the
most cost-effective way to improve the voltage profile.
31
F ATCS concept and general system considerations
Chapter 4
Line
θ
i
Line
(a)
(b)
(c)
θ
line
i
Ac line
dc
power
link
Coordinated
Control
(d)
(e)
Ac line
θ
line
i
dc power link
dc power link
(f)
(g)
Line
Line
Line
dc power link
Storage
Storage
Storage
(h)
(i)
(j)
Fig.4-1 Basic types of FACTS Controllers (a) general symbol for FACTS controller; (b)
series controller; (c) shunt controller; (d) unified series-series controller; (e) coordinated series
and shunt controller; (f) unified series-shunt controller; (g) unified controller for multiple lines;
(h) series controller with storage; (i) shunt controller with storage; (j) unified series-shunt
controller with storage.
32
F ATCS concept and general system considerations
Chapter 4
Nevertheless, the shunt controller is much more effective in maintaining a required voltage
profile at a substation bus. One important advantage of the shunt controller is that it serves the bus
node independent of the individual lines connected to the bus. The series controller solution may
require, but not necessarily, a separate series controller for several lines connected to the substation,
particularly if the application reason for contingency outage of any one line. However, this should
not be a decisive reason of choosing a shunt-connected controller, because the required MVA size
of the series controller is small compared to the shunt controller, and, in any case, the shunt
controller doses not provide control over the power flow in the lines. On the other hand, seriesconnected controllers have to be designed to ride through contingency and dynamic overloads, and
ride through or bypass short circuit currents. They can be protected by metal-oxide arrestors or
temporarily bypassed by solid-state devices when the fault current is too high, but they have to be
rated to handle dynamic and contingency overload.
The above arguments suggest that a combination of the series and shunt controller (see (Fig.41(e) and (Fig.4-1(f)) can provide the best of both, i.e., an effective power/current flow and line
voltage control. For the combination of series and shunt controllers, the shunt controller can be a
single unit serving in coordination with individual line controllers (see Fig.4-1(g)). The arrangement
can provide additional benefits (reactive power flow control) with a unified power flow controller.
FACTS controllers may be based on thyristor devices with no gate turn-off (only with gate turn-on),
or with power devices with gate turn-off capability. In general, the basic controllers with gate turnoff devices are based on DC to AC converters which can exchange active and/or reactive power
with the AC system. When the exchange involves reactive power only, they are provided with a
minimal storage on the DC side. However, if the generated AC voltage or current is required to
deviate form 90 degrees with respect to the line current or voltage, the converter DC storage can
augmented beyond the minimum required for the converter operation as a source of reactive power
only. This can be done at the converter level to cater to short-term (a few tens of main frequency
cycle) storage needs. In addition, another storage source such as a battery, a superconducting
magnet, or any other source of energy can be added in parallel through an electrical interface to
replenish the converter's DC storage. Any of the converter-based, series, shunt, or combined shuntseries controllers can generally accommodate storage, such as a capacitor, batteries, and
superconducting magnets, which bring an added dimension to FACTS technology (see Fig.4-1(h),
(i), and (j). The benefits of an added storage system (such as large DC capacitors, batteries, and
superconducting magnets) to the controller are significant.
A controller with storage is much more effective for controlling the system dynamics than the
corresponding controller without the storage. This has to do with the dynamic pumping of real
power in or out of the system as against only influencing the transfer of real power within the
system as in the case with controller lacking storage. A converter-based controller can also be
designed with so-called high pulse order or with pulse width modulation (PWM) to reduce the low
order harmonic generation to a very low level. A converter can in fact be designed to generate the
correct waveform in order to act as an active filter. It can also be controlled and operated in a way
that it balances the unbalance voltages, involving the transfer of energy between phases. It can do
all of these beneficial things simultaneously if the converter is designed [3], [9].
33
F ATCS concept and general system considerations
Chapter 4
4.5 Description and definition of FACTS Controllers
The purpose of this section is to briefly describe and define various, shunt, series, and combine
controllers. Before going into a very brief description of a variety of specific FACTS Controllers, it
is worth mentioning here that for the converter-based controller, there are two principal types of
converters with gate turn-off devices. These are the so-called the Voltage-Source Converters and
the Current-Source Converters, As shown in Fig.4-2(a), the Voltage-Source Converter is presented
in symbolic form by a box with the gate turn-off device paralleled by a reverse diode, and a DC
capacitor as its voltage source. As shown in Fig.4-2(a), the Current-Source Converter is presented
by a box with the gate turn-off device with a series diode, and a DC reactor as its current source.
Details of variety of Voltage-Sourced Converters and Current-Sourced Converter suitable for
high power applications will be discussed in next two chapters. It would suffice to say that the
Voltage-Sourced Converter, unidirectional DC voltage or a DC capacitor is presented to the AC
side through a sequential switching of the devices. Through appropriate converter topology, it is
possible to vary the AC output voltage in its magnitude and also in any phase relationship to the AC
system voltage. The power reversal involves reversal currents, not the voltage. When the storage
capacity of the DC capacitor is small, and there is no other power source connected to it, the
converter cannot supply or absorb real power for much more than a cycle. The AC output voltage is
maintained at 90 degrees with reference to the AC current, leading or lagging, and the converter is
used to absorb or supply reactive power only. For the Current-Sourced Converter, the DC current is
presented to the AC side through the sequential switching of devices, as AC current, variable in
amplitude and also in any phase relationship to the AC system voltage. The power reversal involves
reversal of the voltage and not the current. The Current-Sourced Converter is represented
symbolically by a box with a power device, and a DC indicator as its current source.
From the overall cost point of view, the voltage-sourced converters seem to be preferred, and
will be the basis for the presentations of most converter-based FACTS Controller. One of the facts
of life is that those involved with FACTS will have to get used to large number of new acronyms
designated by manufactures for their specific products, and by other various papers on new
controllers or variations of known controllers, some of these acronyms are:
Flexibility of electrical power transmission: The ability to accommodate changes in the electric
transmission system or operating conditions while maintaining sufficient steady-state and transient
margins [3].
Flexibility AC transmission systems (FACTS): Alternating current transmission systems
incorporating power electronic-based and other static controllers to enhance controllability and
increase power transfer capability.
FACTS controllers: A power electronic-based system and other static equipment that provide
control of one or more AC transmission system parameters [3].
34
F ATCS concept and general system considerations
Chapter 4
Line
Line
Line
+
Interface
-
+
Stroage
(a)
(b)
Line
Line
TCR
TSR
TSC
Filter
(c)
(d)
Fig.4- 2 Shunt-connected Controllers (a) Static Synchronous Series Compensator
(STATCOM) based on voltage-sourced and current-sourced converter; (b) STATCOM with
storage, i.e., Battery Energy Storage System (BESS), Superconducting Magnet Energy System
and large capacitor; (c) Static VAR Compensator (SVC); (d) Static VAR Generator (SVG),
Static VAR System, Thyristor-Controlled Reactor (TCR), Thyristor-Switched Capacitor
(TSC), and Thyristor-Switched Reactor (TSC); (d) Thyristor-Controlled Braking Resistor.
4.6 Shunt connected controllers
4.6.1 Static Synchronous Compensator
A static synchronous generator operated as a shunt–connected static var compensator whose
capacitive or inductive output current can be controlled independently by the AC system voltage.
The Static Synchronous Compensator (STATCOM) is one of the key FACTS controllers. It can be
based on Voltage-Sourced or Current-Sourced Converters. Fig.4-2(a) shows a simple one-line
diagram of the STATCOM based on Voltage-Sourced or Current-Sourced Converter. As mentioned
35
F ATCS concept and general system considerations
Chapter 4
before, from an overall cost point of view, the Voltage Sourced Converters seem to be preferred,
and will be the basis for presentation of most converter-based FACTS controllers [3].
4.6.2 Static Synchronous Generator (SSG)
A static synchronous generator is a static self-commutated switching power converter is
supplied from an appropriate electric energy source and operated to produce a set of adjustable
multiphase output voltage, which may be coupled to an AC power system for the purpose of
exchanging independently controllable real and reactive power. SSG can be seen a combination of a
STATCOM and any energy source to supply or absorb power (see Fig-4-2(b)). The term, SSG
generalizes connecting any source of energy including a battery, flywheel, superconducting magnet,
large DC storage capacitor, another rectifier/inverter…etc. An electronic interface known as a
chopper is generally needed between the energy source and the converter.[3].
4.6.3 Battery Energy Storage System (BESS)
A battery energy storage system is chemical based energy storage system using a shuntcontroller, Voltage-Sourced Converters are capable of rapidly adjusting the amount of energy which
is supplied to or absorbed form an AC system. Fig.4-2(b) shows a simple one-line diagram in
which storage means is connected to a STATCOM. For transmission applications, BESS storage
unit size would tend to be a small (a few tens of MWHs). If the short-time converter rating was
large enough, it could deliver MWs with a high MW/MWH ratio for transient stability. The
converter can also simultaneously absorb or delver reactive power within the converter MVA
capacity. [3]:
4.6.4 Superconducting Magnetic Energy Storage (SMES)
A superconducting electromagnetic energy storage device is containing electrical converters that
rapidly injects and/or absorbs real and/or reactive power or dynamically controls power flow in AC
system. Since the DC current in the magnet does not change rapidly, the power input or output of
the magnet is changed by controlling the voltage across the magnet with a suitable electronics
interface for connection to a STATCOM.
4.6.5 Static Var Compensator (SVC)
A shunt-connected static Var generator or absorber whose output is adjusted to exchange
capacitive or inductive current, this will maintain or control specific parameters of the electrical
power system (typically the bus voltage). This is the general term for a thyristor-controlled or
thyristor-switched reactor, and/or thyristor-switched capacitor or combination (see Fig.4-2(c)).The
SVC is based on thyristors without the gate turn-off capability. It includes separate equipment for
leading and lagging vars; the thyristor-controlled or thyristor-switched reactor for absorbing
reactive power and the thyristor-switched capacitor for supplying the reactive power. The SVC is
considered by some applications as a lower cost alternative to the STATCOM, although this may
not be the case if comparison is made based on the required performance and not just the MVA size
[3], [9].
36
F ATCS concept and general system considerations
Chapter 4
4.6.6 Thyristor Controlled Reactor (TCR)
A thyristor controller reactor is a shunt-connected, thyristor-controlled inductor whose effective
reactance is varied in a continuous manner by partial-conduction control of the thyristor valve. A
TCR is a subset of SVC in which the conduction time and hence, the current in a shunt reactor is
controlled by a thyristor-based AC switch with firing angle control (see Fig.4-2(c)) [3]
4.6.7 Thyristor-Switched Reactor (TSR)
A thyristor- switched reactor is a shunt-connected, thyristor-switched inductor whose effective
reactance is varied in a stepwise manner by full- or zero-conduction operation of the thyristor valve.
A TSR (Fig.4-2(c)) is another a subset of the SVC, the TSR is made up of several shunt-connected
inductors which are switched in and out by thyristor switches without any firing angle control in
order to achieve the required step changes in the reactive power consumed from the system. [3].
4.6.8 Thyristor Switched Capacitor (TSC)
A thyristor switched capacitor is a shunt-connected thyristor-switched capacitor whose effective
reactance is varied in a stepwise manner by a full- or zero-conductions of the thyristor valve. A TSC
(see Fig.4-2(c)) is also a subset of the SVC in which thyristor based AC switches are used to switch
in and out (without firing angle control) shunt capacitors units, in order to achieve the required step
change in the reactive power supplied to the system [3].
4.6.9 Static Var Generator or Absorber (SVG)
A static var generator or absorber is a static electrical device, equipment, or system that is
capable of drawing controlled capacitive and/or inductive current from an electrical power system
and thereby generating or absorbing reactive power. Generally, it would be considered to be
consisted of shunt-connected, thyristor-controlled reactor(s) and/or thyristor-switched capacitors.
The SVG, as broadly defined by IEEE, is simply a reactive power (var) source that, with
appropriate controls, can be converted into any specific or multipurpose reactive shunt compensator
[3].
4.6.10 Static Var System (SVS)
A static var system is a combination of different static and mechanically-switched var
compensators whose output is coordinated.
4.6.11 Thyristor controlled braking Resistor (TCBR)
A thyristor controlled braking resistor is a shunt-connected thyristor-switched resistor is
controlled to aid stabilization of a power system or to minimize the power acceleration of a
generating unit during a disturbance. A TCBR involves cycle-by-cycle switching of a resistor
(usually a liner resistor) with a thyristor-based AC switch with firing angle control (see Fig.4-2(d)).
For lower cost, a TCBR may be thyristor switched, i.e., without firing angle control. However, with
firing control, half-cycle firing by half-cycle firing control can be utilized to selectively damp lowfrequency oscillations [3].
37
F ATCS concept and general system considerations
Chapter 4
4.7 Series connected controllers
4.7.1 Static Synchronous Series Compensator (SSSC)
A static synchronous series compensator is a static synchronous generator operated without an
external electric energy source as a series compensator whose output voltage is in quadrature with
and controllable independent of the line current for the purpose of increasing or decreasing the
overall reactive voltage drop across the line and thereby controlling the transmitted electric power.
The SSSC may include a transiently rated energy storage or an absorbing devices to enhance the
dynamic behavior of the power system by an additional temporal real power compensation to
increase or decrease momentarily the overall real (resistive) voltage drop across the line.
A SSSC is one of the most important FACTS controllers. It is similar to STATCOM except that
the output AC voltage is in series with the line. It can be based on a Voltage-Sourced Converter (see
Fig.4-3(a)), or a Current-Sourced Converter. Usually the injected voltage in series would be quite
small compared to the line voltage, and the insulation to the ground would be quite high. With an
appropriate insulation between the primary and the secondary side of the transformer, the converter
equipment is located at the ground potential unless the entire converter equipment is located on a
platform duly insulated to the ground. The transformer ratio is tailored to the most economical
converter design. (See Fig.4-3(b)) [3], [9].
4.7.2 Interline Power Flow Controller (IPFC)
A interline power flow controller is a recently introduced Controller and thus has no IEEE
definition yet. A possible definition is: The combination of two or more Static Synchronous Series
Compensators which are coupled via a common DC link to facilitate bi-directional flow of real
power between the AC terminals of the SSSC’s, and are controlled to provide independently
reactive compensation for adjustment of real power flow in each line and maintain the desired
distribution of reactive power flow among the lines. [3], [9].
4.7.3 Thyristor Controlled Series Capacitor (TCSC)
A thyristor controller series capacitor is a capacitive reactance compensator which consists of a
series capacitor bank shunted by a thyristor-controlled reactor in order to provide a smoothly
variable series capacitive reactance. The TCSC which is seen in Fig.4-3(c) is based on a thyristor
without the gate turn-off capability. It is an alternative to the SSSC above and link an SSSC. It is a
very important FACTS controller. A variable reactor such as a Thyristor-Controlled Reactor (TCR)
is connected across a series capacitor. When the TCR firing angle is 180 degrees, the reactor
becomes non-conducting and the series capacitor has its normal impedance. As the firing angle is
advanced from 180 degrees to less than 180 degrees the capacitive impedance increases. At the
other end, when the TCR angle is 90 degrees, the reactor becomes fully conducting and the total
impedance becomes inductive, because the reactor impedance is designed to be much lower than
the series capacitor impedance. [3].
4.7.4 Thyristor-switched Series Capacitor (TSSC)
A thyristor-switched series capacitor is a capacitive reactance compensator which consists of a
series capacitor bank. It would be shunted by a thyristor-switched reactor to provide a stepwise
38
F ATCS concept and general system considerations
Chapter 4
control of the series capacitive reactance. Instead of the continuous control of the capacitive
impedance, the approach of switching inductors at firing angle 90 degrees or 180 degrees but
without firing angle control could reduce cost and losses of the Controller (see Fig.4-3(c)). It is
reasonable to arrange one of the modules to have thyristor control, while others can be thyristor
switched [3].
Line
Line
+
Interface
Stroage
-
+
(a)
(b)
Line
Line
(c)
(d)
Fig.4-3 (a) Static Synchronous Series Compensator (SSSC) (b) SSSC with storage; (c)
Thyristor-Controlled Series Capacitor (TCSC) and Thyristor Switched Series Capacitor
(TSSC), and (d) Thyristor-Controlled Series Reactance (TCSR) and Thyristor-Switched
Series Reactance.
4.7.5 Thyristor-Controlled Series Reactor (TCSR)
A thyristor-controlled series reactor is an inductive reactance compensator which consists of a
series reactor shunted by a thyristor controlled reactor in order to provide smoothly variable series
inductive reactance. When the firing angle of the thyristor controlled reactor is 180 degrees, it stops
conducting, and the uncontrolled reactor acts as a fault current limiter (see Fig.4-3(d)). As the angle
decreases less than 180 degrees, the net inductance decreases until the firing angle of 90 degrees,
when the net inductance is the parallel combination of the two reactors. As for the TCSC, the TCSR
may be a single unit or several smaller series units.
39
F ATCS concept and general system considerations
Chapter 4
4.7.6 Thyristor-switched Series Reactor (TSSR)
An inductive reactance compensator which consists of a series reactor shunted by a thyristor
controlled switched reactor in order to provide stepwise control of series inductive reactance. This
is a complement of TCSR, but with thyristor switches fully on or off (without firing angle control)
to achieve a combination of stepped series inductance (Fig.4-3(d)).
4.8 Combined Shunt Series Connected Controllers
4.8.1 Unified Power Flow Controller (UPFC)
A unified power flow controller is a combination of static synchronous compensator
(STATCOM) and static series compensator (SSSC) which are coupled via a common DC link. This
is to allow bidirectional flow of real power between the series output terminals of the SSSC and the
shunt output terminals of the STATCOM. UPFC are controlled to provide concurrent real and
reactive series compensation without an external electric energy source. The UPFC, by means of
angularly unconstrained series voltage injection, is able to control, concurrently or selectively, the
transmission line voltage, impedance, and angle or, alternatively, the real and reactive power flow
in the line. The UPFC may also provide independent controllable shunt reactive compensation.
In UPFC (Fig.4-4(a)), which combines a STATCOM (shown in Fig.4-2(a)), and an SSSC (see
Fig.4-3(a)), the active power for the series unit (SSSC) is obtained from the line itself via the shunt
unit STATCOM; the latter is also used for voltage control with control of its reactive power. This is
a complete controlled for controlling active and reactive power control through the line, as well as
line voltage control. Additional storage such as superconducting magnet connected to the DC link
via an electronic interface would provide the means further enhancing the effectiveness of the
UPFC. As mentioned before, the controlled exchange of real power with an external source, such as
storage, is much more effective in controlling system dynamics than modulation of the power
transfer without a system [3], [9].
4.8.2 Thyristor-Controlled Phase Shifting Transformer (TCPST)
A thyristor-controlled phase shifting transformer is a phase-Shifting transformer adjusted by
thyristor switches to provide a rapidly variable phase angle. In general, phase shifting is obtained by
adding a perpendicular voltage vector in series with a phase. This vector is derived from the other
two phases via shunt connected transformers (see Fig.4-4(a)). The perpendicular series voltage is
made variable with a variety of power electronics topologies. A circuit concept that can handle a
voltage reversal can provide a phase shift in either direction. This controller is also referred to as the
Thyristor-Controlled Phase Angle Regulator (TCPAR) [3].
4.8.3 Inter-phase Power Controller (IPC)
An inter-phase power controller is a series-connected controller of active and reactive power,
consisting in each phase, of inductive and capacitive branches subjected to separately phase-shifted
voltages. The active and reactive power can be set independently by adjusting the phase shifts
and/or the branch impedance, using mechanical or electrical switches. In the particular case where
the inductive and capacitive impedance from a conjugate pair, each terminal of the IPC is a passive
current source dependent on the voltage at the other terminal. This is the broad-based concept of a
series controller which can be designed to provide the control of active and reactive power [3].
40
F ATCS concept and general system considerations
Chapter 4
3-phase line
Line
SSSC
dc link
STATCOM
(a)
(b)
Fig.4-4 (a) Thyristor-Controlled Phase-Shifting Transformer (TCPST) or ThyristorControlled Phase Angle Regulator (TCPR); (b) Unified Power Flow Controller (UPFC).
Line
(a)
(b)
Line
(c)
Fig.4- 5 Various other controllers (a) Thyristor-Controlled Voltage Limiter (TCVL), (b)
Thyristor-Controlled Voltage Regulator (TCVR) based on tap changer, (c) ThyristorControlled Voltage Regulator (TCVR) based on voltage injection.
41
F ATCS concept and general system considerations
Chapter 4
4.8.4 Thyristor-Controlled Voltage Limiter (TCVL)
A thyristor-controlled voltage limiter is a thyristor-switched metal-oxide varistor (MOV) used
to limit the voltage across its terminals during transient conditions. The thyristor switch can be
connected in series with a gapless arrestor (as shown in Fig.4-5(a), or part of the gapless arrestor
(10-20%) can be bypassed by a thyristor switch in order to dynamically lower the voltage limiting
level. In general, the MOV would have to be significantly more powerful than the normal gapless
arrestor, in order that a TCVL can suppress dynamic over-voltages which otherwise last for tens of
cycles [3].
4.8.5 Thyristor-Controlled Voltage Regulator (TCVR)
A thyristor-controlled voltage regulator is a thyristor-controlled transformer can provide
variable in-phase voltage with continuous control. For practical purposes, this may be a regular
transformer with a thyristor-controlled tap changer (see Fig.4-5(b)) or with a thyristor-controlled
AC to AC converter for the injection of variable AC voltage of the same phase in series with the
line (Fig.4-5(c)). Such a relatively low cost controller can be very effective in controlling the flow
of reactive power between two systems [3].
4.9 Benefits of FACTS Controllers:
FACTS Controllers have a lot of benefits in power systems some of them are mentioned in the
following:
• The control of the power flow as ordered. The control of the power flow may be to follow a
contract, meet the utilities own needs, ensure optimum power flow, ride through emergency
conditions, or a combination therefore.
• Increase the loading capability of lines to their thermal capabilities, including short term and
seasonal. This can be accomplished by overcoming other limitations, and the sharing of
power among lines according to their capability. It is also important to note that the thermal
capability of a line varies by a very large margin based on environmental conditions and
loading history.
• Increase the system security through raising the transient stability limit, limiting short-circuit
currents and overloads, managing cascading blackouts and damping electromagnetical
oscillations of power systems and machines.
• Provide secure tie line connection to neighboring utilities and regions thereby decreasing
overall generations reverse requirements on both sides.
• Provide greater flexibility in siting new generation.
• Upgrade online.
• Reduce reactive power flow which allows the lines to carry more active power.
• Reduce loop flows.
• Increase utilization of lowest cost generation. One of the reasons for transmission
interconnections is to utilize lowest cost generation. When this cannot be done, it follows
that there is not enough cost-effective transmission capacity. Cost-effective enhancement of
capacity will therefore increase use of the lowest generation [3], [9].
42
Chapter 5
Voltage-Sourced Converters
5 Voltage-Sourced Converters
5.1 Basic concept of Voltage-Sourced Converter (VSC)
The concept of FACTS Controller conveys that the Voltage-Sourced Converter is the basic
block in STATCOM, SSSC, UPFC, IPFC, and some other controller. Therefore, this chapter will
discuss this converter. As already explained, the conventional thyristor device has only the turn-on
control; its turn-off depends on the current coming to zero as per circuit and system condition.
Devices such as the Gate Turn-off Thyristor (GTO), Integrated Gate Bipolar Transistor (IGBT),
MOS Turn-off Thyristor (MTO), integrated Gate-Commutated Thyristor (IGCT), and other similar
devices have turn-on and turn-off capability. These devices (referred to as turn-off devices) are
more expensive and/or have higher losses than the thyristor without turn-off capability. However,
turn-off devices enable converter concept that can have significant overall system cost and
performance advantages. In principle these advantages result from the converters, which are selfcommutating against the line-commutating converters. Compared to the self-commutating
converter, the line-commutating converter must have an AC source connected to the converter,
which consumes reactive power, and suffers from occasional commutation failures in the mode
converter of operation. Therefore, unless a converter is required to operate in the two laggingcurrent quadrants, only (consuming reactive power while converting active power), converters
applicable to FACTS controllers would be of the self-commutating type. There are two basic
categories of self-commutating converters:
• Current-Sourced Converter in which the direct current always has one polarity and the power
reversal takes place through several DC voltage polarities (will be discussed in Chapter6).
• Voltage-Sourced Converter in which the direct voltage always has one polarity and the
power reversal takes place through several DC current polarities.
Conventional thyristor-based converters, being without turn-off capability, can only be CurrentSourced Converters, whereas turn-off device-based converters can be of either type. For economic
and performance reasons, Voltage-Sourced Converters are often preferred over Current-Sourced
Converters for FACTS applications. Here Voltage-Sourced Converters will be discussed, which
form the basis idea for several FCATS controller. Since the direct current in a Voltage-Sourced
Converter flows in either direction, reverse, the turn-off devices don’t need reverse voltage
capability; such turn-off devices are know as asymmetric turn-off devices. Thus, a Voltage-Sourced
Converter valve is made up of an asymmetric turn-off device such as a GTO, which is shown in
Fig.5-1(a), with parallel diode connected in reverse. Some turn-off devices, such as the IGBTs and
IGCT, may have a parallel reverse diode built in as part of a complete integrated device suitable for
Voltage-Sourced Converters. However, for high power converter, the provision of separate diodes
is advantageous. In reality, there would be several turn-off device-diode units in series for highvoltage application. In general, the symbol of one turn-off device and with one parallel diode, as
shown in Fig.5-1(a), will present a valve of appropriate voltage and current rating required for the
converter. Within the category of voltage sourced-converter, there are also a wide variety of
converter concepts. The ones relevant to FACTS controllers are described here [3].
The basic functioning of Voltage Sourced-Converter is shown in Fig.5-1(b). The internal
topology of the converter valves is represented in a box with a symbol inside. On the DC side, the
43
Chapter 5
Voltage-Sourced Converters
voltage is uni-polar and supported by a capacitor. This capacitor is large enough to handle at least a
sustained charge/discharge current that accompanies the switching sequence of the converter valves
and shifts in phase angle of switching valves without significant change in the DC voltage. In this
chapter, the DC capacitor voltage will be assumed constant. It is also shown on the DC side that the
DC current can flow in another direction. It can exchange DC power with connected DC system in
the either direction.
Turn-off
device
Diode
(a)
dc side
Active dc
power
id
ac side
Active and
reactive ac power
Vd
Va
(b)
id
Vd
1'
ia
A
1
Va
(c).
Fig.5-1 Basic principle of Voltage-Sourced Converter: (a) Valve for a Voltage-Sourced
Converter; (b) Voltage-Sourced Converter concept; (c) Single-valve operation.
Shown on the AC side is the generated AC voltage connected to AC system via an inductor.
Being an AC voltage source with internal impedance, a series inductive interface with the AC
system (usually through a series inductor and/or a transformer) is essential to ensure that the DC
capacitor is not short-circuited and discharge rapidly into a capacitive load such as a transmission
line. Also, an AC filter may be necessary (not shown in Fig.5-1) following the series inductive
interface to limit the consequent current harmonics entering the AC system. Basically, a Voltage
Sourced-Converter generates an AC voltage from a DC voltage. For historical reasons, it is often
referred to as a converter, even though it has the capability to transfer power in both directions.
With a Voltage Sourced-Converter, the magnitude, the phase angle and the frequency of output
voltage can be controlled. In order to further explain the principles, Fig.5-1(c) shows a diagram of a
single-valve operation. The DC voltage, the Vd is assumed to be constant, supported by a large
44
Chapter 5
Voltage-Sourced Converters
capacitor, with the positive polarity side connected to the anode side of the turn-off device. When
the turn-off device 1 is turned on, the positive DC terminal is connected to the AC terminal A, and
the AC voltage will jump to +Vd. If the current happens to flow from +Vd to A (through the device
1), the power would flow from the DC side to AC side (converter action). However, if the current
flows from A to +Vd it will flow through diode 1‘ even if the device 1 is called turned on, and the
power would flow from the AC side to the DC side (rectifier action). Thus, a valve with
combination of turn-off device and diode can handle the power flow in either direction, with the
turn-off device handling converter action, and with the diode handling rectifier action. This valve
combination and its capability to act as a rectifier or as a converter with the instantaneous current
flow in positive (AC to DC side) or negative direction, respectively, is a basic issue in the VoltageSourced Converter concepts [3].
5.2 Single-Phase, Voltage Source Converter Circuits
The basic single-phase converter circuit is shown in Fig.5-2 for single-phase, half-bridge
converter. It can be seen that these circuits have their duals in the AC-DC converter circuits in
which the DC and AC terminals (i.e., the source and load terminals) are interchanged and the
polarities of the switches are reversed. In this converter, the capacitor is split into two seriesconnected halves with the neutral point of the AC side connected to the mid-point N of the DC
capacitor. With the two turn-off devices alternately closing/opening, the AC voltage waveform is
square wave with peak voltage VS/2 (Vd/2). Normally, FACTS controlled will generally utilize
three-phase converters (will be discussed later), a single-phase and single-phase full-wave bridge
converter may also be used in some design. Fig.5-3 presents a single-phase, full-bridge converter. It
is important to understand the operation of a single-phase to further understand the principle of
Voltage-Sourced Converter. The converter which is presented in Fig.5-3, consists of four valves T1
to T4, a DC capacitor to provide stiff DC voltage, and two AC connection points, A and B [3], [10].
C
+
VS/2
T1
L
R
VS
D1
N
Load
C
+
VS/2
T2
D2
Fig.5-2 Single-phase half-bridge converter.
The output load voltage alternates between +Vd when T1 and T2 are on and -Vd when T3 and T4
are on, irrespective of the direction of the current flow. It is assumed that the load current does not
become discontinuous at any time (load constant time is bigger than the switching time). When the
switches T1 and T2 are on, the load current increases exponentially according to the Equation:
di
Vd = L + R i
(5-1)
dt
45
Chapter 5
Voltage-Sourced Converters
Fehler! Es ist nicht möglich, durch die Bearbeitung von Feldfunktionen Objekte zu
erstellen.
Fig.5-3 Single-phase full-bridge converter with RL-load
At the end of the on-period for T1 and T2, io is at maximum, and diodes D3 and D4, which are
necessary to allow a path for the current to flow when the transistor is turned off and protect the
transistor against the over-voltage that would be created by a sudden turn-off of the current through
the inductance load, start conducting. T3 and T4, though on, are now reversing biased by these
diodes. It should be expected that the mean or DC currents in the switches T1 and T2 should add up
to the supply DC current Id which is proportional to the load power. The mean diode currents,
however, do not represent the reactive component of the load power [10].
5.3 Output voltage control of single-phase converter
Converters with fixed output voltage and frequency are normally used for fixed AC power
supplies, such as uninterruptible power supplies for computer installations. These circuits also have
voltage control circuits to overcome variations in the input DC voltage and load, by closed loop
control. The output voltage of these converter circuits are normally regulated very tightly. For other
applications such as in variable speed drives, variable output voltage at variable frequency are often
required. While frequency control is readily obtained, voltage control requires more elaborate
techniques. Methods employed for the voltage control are:
• Input voltage regulation (rarely used for voltage source converters).
• Phase control of converter groups.
• Sinusoidal PWM (SPWM)-bilpolar switched.
• Modified SPWM.
• Uni-polar switched SPWM [10].
5.3.1 Output voltage control via input voltage regulation
In this scheme, the input DC voltage to the converter is adjustable by means of a phasecontrolled or a DC-DC converter which is terminated by a filter capacitor (see Fig.5-4 and Fig.5-5).
The input DC voltage to the converter is thus variable. The converter can be switched with the
square-wave switching signals of fixed frequency, as described previously. The variation of the
input DC voltage to the converter leads to a variable output AC voltage to the load. It should be
appreciated that the usually used filter capacitors in the AC-DC or DC-DC converters means that
the DC input voltage to the converter can only be varied slowly. Thus highly dynamic control of the
converter output voltage can not be achieved [10].
46
Chapter 5
Voltage-Sourced Converters
Vd
~
~
Vd
INV
~
~ ~
α
DC/DC
Converter
INV
~
D
Fig.5-4 The regulation of the output
voltage by means of a phase-controlled.
Fig.5-5 The regulation of the output voltage by
means of a DC-DC converter.
5.3.2 Phase control of the converter legs
The switching signals of the two converter legs of a single-phase bridge converter can be given
as a variable phase shift angle δ. This has the effect of varying the duty cycle of the output voltage
in each half cycle, producing a quasi-square output AC voltage waveform. The phase angle δ is
directly proportional to the duration of the non zero output voltage, so that the phase angle can be
controlled to vary the output voltage. The nth harmonic spectrum of the quasi-square output voltage
is given by following Equation [10]:
Von =
4Vd
nδ
sin ( )
2
2nπ
(5-2)
5.3.3 Sinusoidal Pulse Width Modulation (SPWM)
In this scheme a sinusoidal modulating voltage eC of the desired output frequency, fo, is
compared with a higher frequency tri-angular or saw-tooth carrier waveform to generate the
switching signals for the converter. The amplitude of eC also determines the amplitude (or rms
value) of the fundamental output voltage. The converter can be the half- or the full-bridge circuits
of Fig.5-3. The resulting switching pulses have widths approximately proportional to the sine of the
angular position at the centre of the pulses. The widths of these pulses are also proportional to the
amplitude of the modulating signal eC, relative to the amplitude of the carrier [10].
5.3.3.1 Full-bridge SPWM converter
The full bridge converter shown in Fig.5-3 can be either bipolar or uni-polar switched. In the
bipolar switching scheme, transistors T1 and T2 are switched on together, when eC >vtri, as are T3
and T4, when eC <vtri, as presented in Fig.5-6. The control voltage, eC, is sinusoidal of the frequency
equal to the desired output frequency and amplitude determined by the required rms output voltage.
The carrier frequency is generally much higher than the frequency of the modulating waveform
(eC). Regardless of the direction of current flow in the load, the load voltage waveform is
determined by the state of the switches. The amplitude of each SPWM voltage pulse across the load
is now ±Vd. This switching scheme is called bipolar, as opposed to uni-polar in which both switches
in a diagonal pair may not be switched on or off simultaneously. Two switches in the same leg of
the converter are never turned on together because that causes the constitution of a short circuit
47
Chapter 5
Voltage-Sourced Converters
across the DC source. The bipolar scheme is obtained by a comparator based on the following rule:
When eC >vtri, T1 and T2 are on T3 and T4 are off, When eC <vtri, T3 andT4 are on and T1 and T2 are
off. If the PWM switching or carrier frequency is far higher the frequency of the modulating
waveform, it can be assumed that the modulating wave changes a little over a switching period.
eC
Vtri
Vtri , eC [p.u.]
1,0
T1,2 ON
T3,4ON
+Vd
0,5
0,0
-0,5
-1,0
0,00
0,01
0,02
0,03
t [s]
0,04
1,0
0,5
0,0
0,00
0,01
0,02
0,03
t [s]
0,04
0,02
0,03
t [s]
0,04
0,02
0,03
1,0
0,5
0,0
0,00
0,01
0,00
0,01
V01
200
100
0
-100
-Vd
-200
t [s]
0,04
Fig.5-6 The waveform of the output voltage of Full-bridge converter.
The average output voltage over each switching period is then equal to the depth of modulation
(or the effective duty cycle over the switching period) times the supply voltage, Vd. It should be
expected that the fundamental output voltage waveform should be given by the average voltage
during each switching period. This is given by the dotted sinusoidal of Fig.5-6 which can be
formulated as the following [10]:
Vo1max = mVd for 0 ≤ m ≤ 1
(5-3)
5.3.4 Single-phase SPWM converter with Uni-polar Switching Scheme
In this scheme, the switches T1 and T2 or T3 and T4 are not switched on together in Fig.5-3
(single-phase full-wave converter). Instead, the load current is allowed to circulate through a diode
and the remaining switch whenever the control voltage is lower than the saw-tooth carrier. When
this happens, zero voltage is applied to the load, resulting in three-level output voltage. The local
circulation of the load current, without going through the DC source, means that the load current
ripple is smaller for the same switching frequency. The switching signals for the uni-polar singlephase converter can be obtained from the operation mode as in Table 3.1. The output of each pulsewidth modulator drives one leg of the converter only in a complementary manner. Whenever two
upper or lower switches in the two legs of the converter are simultaneously on, the output voltage
across the load is zero. Unlike the bipolar scheme, the output voltage now has three states, namely:
+Vd, -Vd and zero.
48
Chapter 5
Voltage-Sourced Converters
eC > vtri
T1
On
T4
Off
T3
-
T2
-
eC < vtri
Off
On
-
-
− eC > vtri
-
-
On
Off
− eC < vtri
-
-
Off
On
Table 3.1 Switching signals for uni-polar single-phase converter.
5.4 Three-phase Voltage-Source Converters
Three phase bridge converters can be viewed as extensions of the single-phase bridge circuit, as
shown in Fig.5-7. The switching signals for each converter leg are displaced by 120° with respect
to the adjacent legs. The output line-line voltages are determined by the potential differences
between the output terminals of each leg. Symmetrical three phase voltages across a three-phase
load can be produced by switching the devices on for either 180° or 120° of the output voltage
waveform. With 180° conduction, the switching sequence is T1T2T3 – T2T3T4 – T3T4T5 – T4T5T6 –
T5T6T1 – T6T1T2 – T1T2T3 -… for the positive A-B-C phase sequence and the other way a round for
the negative A-C-B phase sequence. With 120° conduction, the switching pattern is T1T2 – T2T3 –
T3T4 – T4T5 – T5T6 – T6T1 – T1T2-… for the positive A-B-C sequence and the other way a round for
the negative A-C-B phase sequence [10].
id
Vd
+Vd/2
T1
T4
T3
A iA
T
D3 5
B iB
D5
C iC
D4 T
2
D2 T6
D6
D1
-Vd/2
Phase A
A
R
Phase B
Phase C
R
N
R
Fig.5-7 Three-phase full-wave bridge converter.
Whenever an upper switch in a converter leg is connected with the positive DC rail is turned
on, the output terminal of the leg goes to the potential +Vd/2 with respect to the center-tap of the
DC supply. Whenever a lower switch in a converter leg connected with the negative DC rail is
turned on, the output terminal of that leg goes to potential -Vd/2 with respect to the center-tap of
the DC supply. The center-tap of the DC supply Vd has been created by connecting two equal
valued capacitors across it. The center-tap is assumed to be at zero potential or grounded. However,
this contraption is artificial and really not essential. It is assumed that the three-phase load
connected to the output terminals of the converter is balanced [11].
49
Chapter 5
Voltage-Sourced Converters
5.4.1 Converter waveforms with 180° conduction angle
In this case, each switch is turned on for 180°. Switches T1 and T4 which belong to the leftmost converter leg produces the output voltage for phase A. The switching signals for T1 and T4
are complementary, as for T3 and T6 or T5 and T2. The switching signals for switches T3 and T6,
which are for phase B, are delayed by 120° from those for T1 and T4, respectively, for the ABC
phase sequence. Similarly, for the same phase sequence, the switching signals for switches T5 and
T2 are delayed from the switching signals for T3 and T6 by 120° (see Fig.5-8). The phase terminal
voltages at A, B and C (sometimes called respective pole voltages) are determined by the states of
the switches connected at each pole. With 180° conduction (i.e., complementary switching), each
pole voltage can only have two values (or discrete states), namely: +Vd/2 or -Vd/2. Considering
3
that there are three poles, the number possible output voltage states from the converter is 2 = 8 [11]
T1
1
0
T2
1
0
T3
1
0
T4
1
0
T5
1
0
T6
t [s]
0,00
0,01
0,02
0,03
0,00
0,01
0,02
0,03
0,00
0,01
0,02
0,03
t [s]
0,04
0,00
0,01
0,02
0,03
t [s]
0,04
0,00
0,01
0,02
0,03
t [s]
0,04
0,00
0,01
0,02
0,03
t [s]
0,04
t [s]
0,04
0,04
1
0
Fig.5-8 The switching scheme of three-phase Voltage-Sourced Converter.
5.4.2 Converter waveforms with 120°conduction angle
In this switching scheme, switches T1–T6 are each turned on for 120° (instead of 180°for the
previous scheme), as shown in Fig.5-9. Switching signals for each phase leg is displaced from the
switching signals for the adjacent legs by 120°. As a result, the switching signals for each phase leg
have 60° of non overlap. Because of this, switches of a phase leg do not need any dead-time (which
is the time each switch waits before the other completely turns off). Therefore two switches
conduct at any time, in contrast to three of the previous scheme [11].
5.5 Output voltage control of thee-phase converters
The available methods are:
• Input DC voltage regulation which is not favoured for Voltage Source Converters
except for slow adjustment of output voltage.
• SPWM: this method is now favoured for low to medium power applications.
Unlike the case of a single-phase converter, variable phase displacement between converter
legs can not be used as a means for output voltage variation. This is due to the restriction that a
phase displacement of 2π/3 between the phases must be maintained in order to obtain a balance of
50
Chapter 5
Voltage-Sourced Converters
three phase output voltage [11].
T1 1
T2
0
0,00
1
0
0,00
t [s]
0,01
0,02
0,03
0,01
0,02
0,03
0,01
0,02
0,03
t [s]
0,04
T3 1
0
0,00
120°
120°
120°
T5
T6
t [s]
0,04
120°
T4 1
0
0,00
1
0
0,00
1
0
0,00
0,04
0,01
0,02
0,03
t [s]
0,04
0,01
0,02
0,03
t [s]
0,04
0,01
0,02
0,03
t [s]
0,04
Fig.5-9 The switching scheme for three-phase VSC 120° conducting stead of 180°.
5.6. Three-level Voltage-Sourced Converter
The three-level Voltage-Sourced Converter is needed to vary the magnitude of AC output
voltage without having to change the magnitude of the DC voltage. One phase-leg of three-level
converter is shown in Fig.5-10. The other two phase-legs would connect across the same DC busbar
and the clamping diodes are connected to the same mid-point N of the DC capacitor. It is seen that
each half of the phase leg is split into series connected valves, i.e., 1-1’ is split into 1-1’ and 1A1’A. The mid-point of the split valves is connected by diodes D1 and D4 to the mid-point N as
shown. On the face of it, this may be seen like doubling the number of valves from two to four per
phase-leg in addition to providing two extra diode valves. However, doubling the number of valves
with the same voltage rating would double the DC voltage and hence the power capacity of the
converter.
If the converter is a high voltage converter with devices in series, then the number of main
devices would be almost the same. A diode clamp at the mid-point may also help ensure more
decisive voltage sharing between the two valve-halves. On the other hand, requirement some extra
devices may be required if the converter has to continue safe operation with one failed phase in a
string of series connected devices [3].
The output voltage corresponding to one three-level phase-leg is shown in Fig.5-11. The first
waveform shown is full 180 degrees square wave obtained by the closing of devices 1 and 1A to
give +Vd/2 for 180 degrees, and the closing of valves 4 and 4A for 180 degrees to give -Vd/2 for 180
degrees. Consider now that the second voltage waveform in Fig.5-11 in which the upper device,
device 1, is turned off and device 4A is turned on an angle α earlier than they were due in the 180
degrees square wave operation. This leaves only device 1A and 4A on, which in combination with
diodes D1 and D2, clamp the phase voltage Va to zero with respect to the DC mid-point N regardless
of which way the current is flowing. This continues for a period 2α until device 1A is turned off
and device 4 is turned on, and the voltage jumps to -Vd/2 with both the lower device 4 and 4A
51
Chapter 5
Voltage-Sourced Converters
turned on and both upper devices 1 and 1A are turned off and so on. Of course, angle α is variable,
and the output voltage Va is made up of σ ° = 180° − 2α ° square waves. This variable period, σ, per
half-cycle potentially allows the voltage Va to be independently variable with potentially a fast
response. It is seen that the devices 1A and 4A are turned on for 180 degrees during each cycle
devices 1 and 4 are turned on for σ ° = 180° − 2 α ° during each cycle, while D1 and D2 conduct for
2 α ° = 180° − σ each cycle. The converter is referred to as three-level because the DC voltage has
three levels, i.e. −Vd / 2, 0, + Vd / 2 [3].
+Vd/2
1′
1
D1
1′A
1A
ia
N
4A
4′A
D4
4
4′
-Vd/2
Fig.5-10 One phase-leg of a three-level converter.
5.6.1. Pulse width modulation (PWM) for Three Level Converter
In some two-level or multilevel converters, there is only one turn-on turn-off per device per
cycle. In these converters, the AC output voltage can be controlled by varying the width of the
voltage pulses and/or the amplitude of the DC bus voltage. Another approach is to have multipulses
per half-cycle and to vary the width of the pulses to vary the amplitude of the AC voltage. The
principle reason for doing so is to be able to vary the AC output voltage and to reduce the low-order
harmonics as will be explain here briefly. It goes without saying that more pulses means more
switching losses, so that gains from the use of PWM have to be sufficient to justify the increase in
switching losses. There are also resonant PWM converter topologies that incorporate current-zero
type soft switching in order to reduce the switching losses. Such converters are being increasingly
utilized in some low power applications, but with the known topologies they have not been
justifiable at high power levels due to higher equipment cost [3].
52
Chapter 5
Voltage-Sourced Converters
Fig.5-12. shows the PWM output voltage waveform corresponding to a PWM frequency of
three times the main frequency: The first waveform shows the control signals, similar to those in
Fig.5-12. The second waveform is the phase a to DC neutral voltage vaN. It is seen that it has one
notch in the centre of each half-cycle. The third waveform is vbN, the output voltage of phase b to
DC neutral voltage, which is obviously the same as vaN, except the delay is 120 degrees. Subtracting
vaN from vbN gives the phase-to-phase vab, as shown by the fourth waveform. This shows two
notches resulting from the crossing of control signals.
+Vd/2
Va
1,1A
1,1A
180°
-Vd/2
1,1A
2α
σ
Va
4,4A
1,1A
1A, 4A
3,3A
4,4A
Vb
3A,5A
5,5A
+Vd/2
Vd
Va-Vb
-Vd
Fig.5-11 Operation of thee-level converter, output AC voltage.
The next waveform is that of the vaN, the voltage between the floating neutral n of a wyeconnected floating secondary and the DC neutral. This is obtained by adding and averaging the
three AC voltages, vaN, vbN, and vcN, (vcN not shown). Subtracting vnN from vaN gives the last
waveform shown in Fig.5-12, that of the transformer phase-to-neutral voltage. Because of the halfwave symmetry, all the AC waveforms are free from even harmonics. Waveforms vab and van are
triplen harmonics and the van lags vab by 30 degrees. As explained earlier, combining these two
waveforms through separate wye and delta transformers will result in 12-pulse converter, which
will have adequate flexibility of rapid AC voltage control without having to change the DC voltage
level. The control of the DC voltage can then be optimized for other considerations. The pulses are
wider in the middle of each half sine wave compared to the ends of the half sine wave [3], [11].
5.7 Summary
The VSC is an important element in FACTS applications and it have the following advantages
which can be summarized as follows:
• Continuous operation, compensation, and control for reactive power requirements and
voltage control/ stability applications
• Rapid and continuous response characteristics for smooth dynamic control.
53
Chapter 5
•
•
Voltage-Sourced Converters
Independent control of voltage and power flow for direct power transfer applications.
Automated real and reactive power control for both steady-state and dynamic system
conditions.
a
Vabc,Vtri [p.u]
1,0
b
c
0,5
0,0
-0,5
-1,0
VaN [p.u.]
0,000
1,0
0,005
0,010
0,015
t [s]
0,020
0,005
0,010
0,015
t [s]
0,020
0,5
0,0
-0,5
-1,0
0,000
VbN
1,0
VbN [p.u.]
0,5
0,0
-0,5
-1,0
0,000
0,005
Vab [p.u.]
2
0,010
0,015
t [s]
0,020
0,010
0,015
t [s]
0,020
Vab
1
0
-1
-2
0,000
0,005
Fig.5-12 Operation of a PWM converter with switching frequency of three times the
fundamental frequency
•
•
•
•
•
Superior performance for weak system conditions (low short circuit ratio application).
Inherent modularity and redundancy for increased reliability and availability.
Advanced control methodologies for high-performance Operation.
Elimination or reduced requirements for harmonic Filtering.
Ability to add energy storage as the sourcing element (e.g., batteries, super-conducting
elements, etc.).
• Compact size and reduced volume for installation flexibility and reduced construction
costs.
• Easy expansion and mobility for future system Considerations.
Advanced power semi-conductor technologies for lower losses, reduced operating costs, and high
reliability [9].
54
Chapter 6
Current-Sourced Converter and Self- and Line-Commutated
6 Current-Sourced Converter and Self- and Line-Commutated
6.1 Introduction and basic concept of Current-Sourced Converter
A Current-Sourced Converter is characterised by the fact the DC current flow is always in one
direction and power flow reverses with the reversal of the DC voltage. In this respect, it differs from
the Voltage-Sourced Converter in which the DC voltage always has one polarity and the power
reversal takes place with reversal of the DC current. Fig.6-1 conveys the difference between the
current sourced and the voltage sourced converter. In Fig.6-1(a), the converter box for VoltageSourced Converter is symbolically shown with a turn-off device with a reverse diode, whereas the
converter box for the Current-Sourced Converter is shown without a specific type device in Fig.61(b). This is because the Voltage-Sourced Converter requires turn-off devices with reverse diodes,
the Current-Sourced Converter may be based on diodes, conventional Thyristors or the turn-off
devices.
Id
Active power
dc power
Vd
Reactive power
(a)
Id
dc power
or
or
Active power
or
Reactive power
or Vd
(b)
Fig.6-1 Voltage-sourced and Current-Sourced Converter concepts: (a) voltage sourced
converter; (b) Current-Sourced Converter.
There are three principle types of Current-Sourced Converter as demonstrated in Fig.6-2:
• Diode converter, which is shown in Fig.6-2(a), which simply converts AC voltage to DC
voltage. It utilizes AC system voltage for commutation of DC current from one valve to
another. Obviously the diode-based line-commutating converter just converts AC power to
DC power without control and also in doing consumes some reactive power on the AC side.
• Line-commutated converter based on conventional thyristors (with gate turn-on but without
gate turn-off capability), shown in Fig.6-2(b) and it utilizes an AC system voltage for
commutation of the current from one valve to another. This converter can convert and control
active power in either direction, but in doing so consumes reactive power on the AC side. It
can not supply reactive power to the AC system.
• Self-commutated converter, which is shown in Fig.6-2(c), is based on turn-off devices
(GTO’s, MTO’s, IGCT’s, IGBT’s, etc.). Here the commutation of current from valve to
valve takes place with the device turn-off action and provision of AC capacitors to facilitate
55
Chapter 6
Current-Sourced Converter and Self- and Line-Commutated
transfer of current from valve to valve. In a Voltage-Sourced Converter the commutation of
the current is supported by a stiff DC bus with a DC capacitor. In a self-commutated CurrentSourced Converter, the AC capacitors provide a stiff AC bus for supplying the fast changing
current pulses needed for the commutation. A way from its capability of controlled flow in
either direction, this converter (like the Voltage-Sourced Converter) can also supply or
consume controlled reactive power. However, it is interesting to note that even though the
converter can supply reactive power, sources of reactive power, i.e. capacitors and AC filters
are needed in any case. One advantage of the converters with turn-off devices (selfcommutating converters) is that they offer greater flexibility including the PWM mode of
operation [3].
Active & reactive
power
dc current
dc voltage
dc power
Filters & capacitors
(a)
dc current
Active power
dc power
dc voltage
Reacctive power
Filters & capacitors
(b)
dc current
dc power
Active power
dc voltage
Reacctive
power
Capacitors
Filters
(c)
Fig.6-2 Types of Current-Sourced Converter, (a) diode rectifier; (b) Thyristor linecommutated converter, (c) self-commutated converter.
It is worth mentioning have that when the converters are based on turn-off devices, the VoltageSourced Converters are preferred over the Current-Sourced Converters. In fact, none of the
converter-based controllers described here are based on Current-Sourced Converters. However,
with evolution in device characteristics and functional details of the converters, this situation may
change in the future. Therefore, the Current-Sourced Converters with turn-off devices are not
discussed in detail here. When reactive power management is not a problem, where controlled
reactive power supply is not required and the active power consumed by the converters can be
56
Chapter 6
Current-Sourced Converter and Self- and Line-Commutated
supplied from the system capacitors and/or filters, the line-commutated converters have decisive
economic advantages over self-commutated converters. For conversion AC to DC and DC to AC, in
HVDC transmission, line commutated converters are used almost exclusively where reactive power
is managed through capacitors, filters and the power system. The converters for superconducting
storage can be Current-Sourced Converters since the superconducting reactor it self is a current
source. Also, the DC power supply for storage means, can be Current-Sourced Converters, in order
to drive Voltage-Sourced Converter-based phase-angle regulators. The economic advantage of
conventional Thyristor-based converters arises from the fact that on a per device basis thyristors can
handle two to three times the power than the next most powerful GTO’s, IGCT’s, MTO’s, etc.
Other converters which are variations of the basic types above of Current-Sourced Converters,
such as thyristors converters with artificial commutation, resonant converters and hybrid converters,
are not discussed here. Since the DC voltage in a Current-Sourced Converter can be in either
direction, the converter valve must have both forward and reverse blocking capability. The
conventional thyristors are usually made as symmetric devices. They have both the forward and
reverse blocking capability. This is because they are on the other hand easier and cheaper to make
and can be made with peak blocking voltage as 12kV along with a high current carrying capability.
On the other hand, the turn-off devices have a high on-state forward voltage drop when they are
made symmetric devices. Given that the high production volumes of asymmetric turn-off devices
dictated by the industrial market, it may be advantageous to connect an asymmetric turn-off device
and a diode in series to get a symmetric devices combination. This results in higher forward voltage
drop and losses. Based on this and on other aspect, such as fast-switching characteristics of IGBT’s,
the industrial converter market has shifted very fast towards the PWM Voltage-Sourced Converter
as discussed before [3].
6.2 Single-phase bridge rectifier
A single-phase bridge rectifier is presented in Fig.6-3. These types of rectifiers do not suffer
from the problem of DC magnetization and low device and transformer utilization. They also offer
higher DC output voltage for a given AC supply voltage. This is at the cost of lower efficiency,
because there are two diode drops between the load voltages. The AC supply voltage is given in the
following:
π
2
vS = Vmax sin(ω t ), Vd = ∫ sin(ω t )d (ω t ) = Vmax
(6-1)
0
π
where Vmax is the peak of the input AC voltage to the rectifier. The PRV (Peak Reverse Voltage) of
each diode is Vmax, not 2Vmax, as in the case of the centre-tapped rectifier [12].
6.3 Three-phase bridge rectifier
A three-phase bridge rectifier (diode rectifier) is presented in Fig.6-4. In this bridge rectifier, the
diodes 1, 3 and 5, whichever have a more positive voltage at its anode, conduct respectively.
Similarly, diodes 2, 4 and 6, whichever have a more negative voltage at its cathode, return the load
current. With the numbering of diodes as indicated in Fig.6-5 the conduction patterns are 12-23-3445-56-61-12 for a positive voltage sequence a-b-c. For the negative voltage sequence a-c-b, the
patterns are 16-65-54-43-32-21-16.
57
Chapter 6
Current-Sourced Converter and Self- and Line-Commutated
D2
i1
iP
iL
D3
L
Vd
vS
R
D2
D4
Fig.6-3 Diode bridge rectifier.
When any of the diodes connected with the top (+ve) rail conducts, the potential of the rail is
the corresponding AC line voltage When any of the diodes connected with the bottom (-ve) rail
conducts, the potential of the rail is the corresponding AC line voltage. When any of the diodes
connected with the bottom (-ve) rail conducts, the potential of the rail is the corresponding AC line
voltage.
vO
van
D1
ia
D3
L
vbn i
b
Vd
ic
n
iL
D5
R
vcn
D4
D2
Load
D6
Fig.6-4 A three-phase bridge rectifier.
The voltage across the load is the difference between the +ve and the -ve rail potentials. Assuming
that the load current is continuous (i.e., non zero) at all times, each diode conducts for 120° in each
half cycle of the AC waveform, followed by 240° of non conduction. The output voltage waveform,
the Vd can be given by the following Equation:
Vd =
6
2π
π /6
∫π
− /6
3
Vmax l −l cos(ω t )d (ω t ) = Vmax l −l
π
(6-2)
where Vmaxl-l is the peak value of the line-line voltage [12].
6.4 Phase-controlled AC-DC converters
The controlled AC-DC rectifier circuits with thyristors are commonly used in applications
requiring continuously variable DC supplies from a few kilowatts to several hundreds or thousands
58
Chapter 6
Current-Sourced Converter and Self- and Line-Commutated
of kilowatts. The thyristor switch may be viewed as a controlled diode which is turned on by the
gate current; a few milliamp or at least amps will turn even the largest device on, when its anode to
cathode voltage is positive. Once the thyristor is fired or triggered on. The Thyristor will turn off
when the anode current falls, brought about by the AC source and load, below a threshold close to
zero. The thyristors in an AC rectifier, therefore, must be triggered synchronously with the AC
supply each cycle, by means of a gate control circuit which is interfaced with the AC mains. The
firing angle α, is normally defined to be the angle for which the output DC voltage is maximum.
The thyristors are triggered with short (a few volts, mA level) pulses, one time in each AC cycle, as
shown in Fig.6-5. These pulses are obtained from a firing controller circuit which is synchronised
with the AC mains. In some converter circuits, the firing pulses for each thyristor are maintained for
the intended duration of conduction for the Thyristor [13].
Vmax sin(ω.t )
L
o
a
d
Firing
Control
Circuit
+10 V
-10 V
-10 V
Fig.6-5 The rectifier and the phase controller for a half-wave converter.
6.4.1 Single-phase, fully-controlled bridge rectifier
This diagram of this converter is presented in Fig.6-6. Assuming continuous conduction, the DC
output voltage is given by:
Vd =
1
π +α
π ∫α
Vmax sin(ω t )d (ω t ) =
2 Vmax
π
cos α
(6-3)
where Vmax is peak value of the input line-line voltage. For the same DC output voltage, the input
AC voltage is now half that of the CT rectifier. This converter operates in quadrants one and four.
There are now two device drops (about 3V) between the transformer and the load. Also, the
transformer secondary current is bi-directional and its phase angle with respect to the AC supply
voltage to the converter is roughly given by the firing angle α. With continuous conduction, each
thyristor conducts for 180° in each cycle. This 2-pulse converter operates in two quadrants [13].
6.5 Three-phase fully controlled bridge converters
Three-phase fully controlled bridge converter circuit is presented in Fig.6-7, the thyristor
connected with the positive DC rail, which has the most positive voltage at its anode, conducts
59
Chapter 6
Current-Sourced Converter and Self- and Line-Commutated
when triggered. The thyristors connected with the negative DC rail, the thyristor with the most
negative voltage at its cathode returns the load current, if triggered. It will be useful to see the
numbering of the thyristors and the sequential triggering of the thyristors. Commutation of the load
current from one thyristor to the next occurs at the firing instant, when the incoming thyristor
reverse biases the previously conducting thyristor, as presented in Fig.6-7.
T1
i1
iL
T3
iP
L
Vd
vS
R
T2
T4
Fig.6-6 A single-phase fully-controlled bridge rectifier (thyristor with p=2).
Having established the conduction times of the thyristors, the output DC voltage waveform is
determined by the difference of potentials of the positive and negative rails. For continuous
conduction, the potentials of each rail are known at all times from the firing angles and the input
AC voltages, regardless of the load. Assuming continuous conduction, the output voltage Vd can be
formulated as:
Vd =
3
2π / 3+α
π ∫π
/ 3+α
Vmax l −l sin(ω t )d (ω t ) =
3 Vmax l −l
π
cos α
(6-4)
This converter operates in quadrants 1 and 4, developing both positive and negative polarity DC
output voltage. For firing angles 0° ≤ α ≤ 90° , the converter operates in quadrant 1 (giving positive
output power, i.e., rectifier operation) and for 90° ≤ α ≤ 180° , the operation is in quadrant 4 (giving
negative output power, i.e., inverter operation). Operation in quadrant 4 is of course possible only
when the load includes an active DC source, able to supply power into the AC supply circuit [13].
van
T1
ia
T3
T5
L
vbn i
b
Vd
ic
n
vO
iL
R
vcn
T4
T2
T6
Fig.6-7 Three-phase, fully-controlled bridge converter circuit.
60
Load
Chapter 6
Current-Sourced Converter and Self- and Line-Commutated
6.6 Current-Sourced Converter with turn-off devices
The Current-Sourced Converter with turn-off devices is also referred to as current-stiff converter.
As mentioned previously, the turn-off devices must have the reverse withstand voltage capability
(symmetric device) or have diodes in series if they are asymmetric devices.
In the Current-Sourced Converter with the conventional thyristor which discussed before, the
operation of the converter is limited to the third and fourth quadrant (lagging power factor). This is
because thyristors don’t have turn-off capability and the DC current has to be commutated from
valve to another while the anode-cathode voltage of the incoming valve is still positive. Also, such a
converter needs an AC voltage source for commutation.
In the Voltage-Sourced Converters which is discussed in chapter5, there is a DC capacitor,
which facilitates rapid transfer of current from an outgoing turn-off valve to the opposite valve in a
phase-leg, irrespective of the direction of the AC current. The capacitor is assumed to be large
enough to handle alternate charging and discharging without substantial change in the DC voltage.
With turf-off capability, the valves can be turned off as well. However, turn-off devices in order to
turn off still require an alternate path for rapid transfer of the current. Otherwise, they will have to
dissipate a large amount of energy to turn off the current in an inductive circuit. It can be visualized
that if the capacitors are placed between the phases, on the AC side of the valves as presented in
Fig.6-8(a). The capacitors can facilitate fast transfer of the current from the outgoing turn-off valve
to the incoming valve.
The commutation of the current from the valve 1 to the valve 3 is illustrated in Fig.6-8(b). Given
low inductance of the AC shunt capacitor and the bus connection, the transfer (commutation) is fast
and there is no commutation angle to speak of as far as the valves are concerned. Actually, with
respect to the turn-on di/dt limit of the devices, the inductance of the capacitors and the bus
connections can be duly exploited. Also it is to be noted, that a valve turns off, valve 1 in Fig.68(b), its rate of rise of the voltage is cushioned by the AC capacitor. It is a complex and important
matter in terms of the devices losses and snubber requirements. These capacitors need to handle a
sustained alternating charge/discharge current of the converter valves. Unlike the line commutated
converter using conventional thyristors, this converter with turn-off valves can operate even with
leading power factor and does not need a pre-existing AC voltage for commutation. It can in fact
operate as an inverter into passive or an active AC system.
Fig.6-8(c) shows the anode-bus current connected to the anode side of valves 1, 3, and 5 and
transfer of this incoming DC current from valve 1 to 3, to 5, to 1, etc., in a closed three-valve
sequence in a three-phase converter. Similarly shown are the cathode bus current, the outgoing DC
current, and how it transfers from valve 2, to 4, to 6, to 2 etc., in a closed three-valve sequence. The
two sequences are phase shifted by 60 degrees and they are together from a three-phase, full-wave
bridge converter. Consequent injected AC current in the three phase as it is shown in Fig.6-8(c),
which is same as for the conventional thyristor converter when neglecting commutation angle, see
Fig.6-8(b).
The currents are injected without the support of the AC system voltage, and therefore, the phase
angle and the frequency of this injected AC current can be controlled. To understand the operation
of this converter, it is appropriate to visualize it as an AC current generator connected to an AC
system, which is front-ended with AC capacitor as presented in Fig.6-8(d). The AC side
61
Chapter 6
Current-Sourced Converter and Self- and Line-Commutated
fundamental and harmonics are a function of the AC system and the injected current. A DC side
converter voltage will also have harmonics.
Id
4
2
6
or
3
1
AC system or
passive load
5
(a) 6-pulse converter
2
3
1
(b) Commutation process.
Id
3
1
6
4
2
Cathode bus current
1
5
6
1
Anode bus current
1
ia
4
3
6
ib
6
5
2
2
ic
(c) Current waveforms.
(d) System interface.
Fig.6-8 Self-commutating current-sourced converter: (a) six-pulse converter, (b)
commutation process, (c) current waveforms, and (d) system interface.
62
Chapter 6
Current-Sourced Converter and Self- and Line-Commutated
Various PWM concepts, which like e.g. those discussed in Chapter5, are applicable to the
current-sourced converters; the advantage of the PWM operation is that the commutation capacitor
size will decrease [3].
6.7 Current-Sourced versus Voltage-Sourced Converters
There are some advantages and disadvantages of current-sourced versus voltage-sourced
converters:
• Diode-based converters are the lowest cost converters, if control of active power by the
converter is not required.
• If the leading reactive power is not required, then a conventional thyristor-based converter
provides a low-cost converter with active power control. It can also serve as controlled
lagging reactive power load (like a thyristor-controlled reactor).
• The current-sourced converter does not have high short-circuit current, as does the voltagesourced converter. For current-sourced converters, the rate of rise of fault current during
external or internal faults is limited by the dc reactor. For the current-sourced converters,
the capacitor discharge current would rise very rapidly and can damage the valves.
In a current-stiff converter, the valves are not subject to high dv/dt, due to the presence of the ac
capacitors
• AC capacitor required for current-stiff converters can be quite large and expensive,
although their size can be decreased by adoption of PWM topology. In general the problem
of satisfactory interface of current-sourced converters with the ac system is more complex.
• With the presence of capacitors, which are subjected to commutation charging and
discharging, this converter will produce harmonic voltages at a frequency of resonance
between the capacitors and the ac system inductance. Adverse effects of this can be
avoided by sizing the capacitors such that the resonance frequency does not coincide with
characteristic harmonics.
• The harmonics as well as the presence of dc reactor can result in over-voltages on the valves
and transformers.
Widespread adoption of asymmetrical devices, IGBT’s and GTO’s, as the devices of choice for
lower on-state losses, has made current-sourced converters a favourable choice when turn-off
capability is necessary. The devices market is generally driven by high-volume industrial
applications and as a result symmetrical turn-off devices by high-voltage rating and required
operating characteristics may not be readily available until the volume of the FACTS market
increases [3].
63
Chapter 7
Snubber Circuits
7 Snubber Circuits
7.1 Introduction
When a power electronic converter stresses a power semiconductor device beyond its ratings,
there are two basic ways of relieving the problem. Either the device can be replaced by one whose
ratings exceed the stresses or a snubber circuit can be added to the basic converter to reduce the
stresses to safe levels. The final choice will be a trade-off between cost and availability of the
semiconductor device with the required electrical ratings compared to the cost and the additional
complexity of using a snubber circuit. The power electronics circuit designer must be familiar with
the design and operation of basic snubber circuits in order to make the comparison trade-off. This
chapter discusses the fundamentals of the snubber circuits commonly used in power electronics to
reduce electrical stresses on power semiconductor devices.
7.2 Function and types of snubber circuits
The function of a snubber circuit is to reduce the electrical stresses placed on a device during
switching by a power electronic converter to levels that are within the electrical ranges of the
devices. More explicitly, a snubber circuit reduces the switching stresses to safe level by:
• Limiting the voltages applied to a device during turn-off transients.
• Limiting the device currents during the turn-on transients.
• Limiting the rate of rise (di/dt) of the current through the devices at turn-on.
• Limiting the rate of rise (dv/dt) of the voltage across the devices at turn-off or during
reapplied forward blocking voltage.
• Shaping of the switching trajectory of the device as it turns on and off.
From the circuit topology perspective, there are three broad classes of snubber circuits. These
classes include:
• Unpolarized series RC snubbers used to protect diodes and thyristors by limiting the
maximum voltage and dv/dt at reverse-recovery.
• Polarized RC snubbers. These snubbers are used to shape the turn-off portion of the
switching trajectory of controllable switches, to clamp voltage applied to the devices to safe
levels, or to limit dv/dt during device turn-off.
• Polarize RL snubbers. These snubbers are used to shape the turn-on switching trajectory of
controllable switches and /or to limit di/dt during device turn-on.
Switching stresses are also controlled by utilizing a board class of power electronic converter
circuit termed resonant or quasi-resonant converters. It must be emphasized that snubbers are not a
fundamental part of power electronic converter circuit. The snubber circuit is an additional part to
the basic converter, which is added to reduce the stresses on an electrical component. Usually, in a
power semiconductor device, snubbers may be used singly or in combination depending on the
requirements. As mentioned before, the additional complexity and the cost added to the converter
circuit by the presence of the snubber must balance against the benefits of limiting the electrical
stresses on critical circuit components [8].
64
Chapter 7
Snubber Circuits
7.3 Diode snubber
Snubbers are needed in diode circuits to minimize over-voltage occurring in circuits such as the
step-down converters in Fig.7-1(a) and the other diode applications like in FACTS systems, due to
the stray or leakage inductance in series with the diode and snap-off the diode reverse recovery
current at the turn-on switch T (see Fig.7-1(a)). The analysis of the snubber circuit that will protect
the diode will be based on this step-down converter circuit, where Lσ is the stray inductance. It is
shown later that for the purpose of snubber analysis, this circuit is an equivalent circuit for almost
any converter where diodes are used. An RS-CS snubber is commonly used across the diode for
over-voltage protection as shown in Fig.7-1(a).
To simplify the analysis, the diode reverse recovery current is assumed to snap off
instantaneously as shown in Fig.7-1(b). The load is inductive and it is assumed that the load current
I0 is constant during the switching transient [8], [14], and [15].
iDf(t)
Lσ
+
iDf
Vd
-
iLσ
Df
IO
Io
RS
+
vD
t
V
di
=− d
dt
Lσ
CS
Irr
-
VDf(t)
Vd
t
T
Lσ
diL σ
dt
(a)
(b)
Fig.7-1 (a) A step-down converter circuit with stray inductance and a snubber circuit for the
free-wheel diode, (b) the Diode reverse-recovery current and diode voltage.
7.3.1 Capacitive snubber
Although the capacitive snubber (RS = 0) is not used in practice, it provides an easily analyzed
starting point for analysis that illustrates the basic concept. In obtaining an equivalent circuit, the
switch in Fig.7-2(a) is assumed to be ideal, which results in a worst-case analysis of the circuit.
Treating the instant of diode snap-off at the peck reverse recovery current Irr (Repetitive reverse
current) at t =0, the initial inductor current in the equivalent circuit of Fig.7-2(c), is Irr and the initial
snubber capacitor voltage is zero. To establish a baseline circuit, the snubber resistance RS is
assumed to be zero as in Fig.7-2(b), the capacitor voltage which is the negative of the diode voltage
in this baseline circuit is given by:
vCS = Vd − Vd cos(ω0 t ) + I rr
Lσ
sin(ω0 t )
CS
(7-1)
where
ω0 = 1/( Lσ CS )
65
(7-2)
Chapter 7
Snubber Circuits
Introducing a baseline capacitance Cbase given by:
Cbase = Lσ [
I rr 2
]
Vd
(7-3)
It is possible to express in Equation (7-1) as:
Cbase
sin(ω0 t )]
CS
vCS = Vd [1 − cos(ω0 t ) +
(7-4)
Either by a time derivative or a phasor approach, the maximum value of vCs in Equation (7-4) can be
estimated as:
vCS ,max = Vd [1 + 1 +
Cbase
]
CS
(7-5)
The waveform and inductor current are shown in Fig.7-2(c), for CS = Cbase . In this case the
maximum reverse-diode voltage is the same as VCs,max calculated from Equation (7-5). For small
value of CS, the maximum diode voltage becomes excessive [8], [14], and [15].
7.3.2 Effect of adding a snubber resistor
When the diode snubber resistor RS is included, the equivalent circuit for the snubber becomes
as shown in Fig.7-3. In analyzing this modified circuit the instant of diode snap-off is treated
as t = 0 , and the initial inductor current is Irr and the initial capacitor voltage is zero. The
differential equation governing the behavior of the diode voltage is:
Lσ CS
d 2 vD f
dt
2
+ RS CS
dvD f
dt
+ vD f = Vd
(7-6)
The boundary conditions are: vD f (0+ ) = − I rr RS and:
dvD f (0+ )
dt
The solution of the Equation (7-6) is given by:
vD f (dt ) = −Vd −
=−
I rr RS Vd I rr RS2
−
−
CS
Lσ
Lσ
Lσ I rr
e α t × cos(ωα t − φ − γ )
C S cos(φ )
(7-7)
where
ωα = 1 −
R
V −I R /2
ω
α2
; α = S ; φ = tan -1 ( d rr S ) ; and γ =tan -1 ( α )
2
ω0
ωα Lσ I rr
α
2 ωα
(7-8)
The time t = tm at which the voltage given by the Equation (7-7) is a maximum can be found by
setting the derivative dvD f / dt equal to zero and solving for time. Doing these yields:
tm =
φ +γ −π / 2
≥0
ωα
66
(7-9)
Chapter 7
Snubber Circuits
iLσ
Lσ
I rr
+
RS
Cathode
Vd
_
Lσ
iC
iLσ
+
+
Diode
snap-off
Vd
+
Anode
CS
CS
T
_
vCs
_
(a)
vCs
_
(b)
vCs
VCs ,max
Irr
Vd
0
i Lσ
di
Lσ
= Vd
dt
t
(c)
Fig.7-2:(a) Equivalent circuit of the step-down converter at the instant of diode reverse-recovery
current snap-off (b) the simplification that results when the snubber resistance is zero and (c)
The voltage and current waveforms for RS = 0 and CS = Cbase.
_
Lσ
+
Vd
RS
VDf(t)
_
+
CS
Fig.7-3 Equivalent circuit with snubber resistance RS.
Substituting t = tm into the Equation (7-7) yields the maximum reverse recovery voltage across the
diode as:
⎧⎪
Vmax
C
R
R
⎪⎫
= 1 + ⎨ 1 + base + S − 0.75( S ) 2 ⎬ e −α tm
Vd
CS
Rbase
Rbase ⎪⎭
⎪⎩
(7-10)
In the Equation (7-10), the baseline capacitive Cbase is given by the Equation (7-3) and the
resistance Rbase is given by:
V
Rbase = d
(7-11)
I rr
Typical circuit waveforms for t > 0 are shown in Fig.7-4 as an example, for CS = Cbase , in these
waveforms, the oscillations damped by RS, and the maximum diode voltage depends on the values
of RS and CS which are used. For selected values of CS, the maximum diode voltage varies with RS
67
Chapter 7
Snubber Circuits
as an example for the plotted curve in Fig.7-6 as a function of RS /Rbase. It can be seen that for this
value of CS, there is an optimum value of RS = Ropt = 1.3 Rbase that minimizes Vmax. A snubber design
monogram is shown in Fig.7-6, where the optimum snubber resistance and corresponding Vmax are
plotted as a function of CS. In this monogram all quantities are normalized.
Fehler! Es ist nicht möglich, durch die Bearbeitung von Feldfunktionen Objekte zu
erstellen.
Fig.7-4 The current and the voltage waveforms after diode snaps-off at t = 0.
The energy loss in the resistor RS is given by the following Equation:
1
WR = [ Lσ I rr2 + CS Vd2 ]
2
(7-12)
where, WR is normalized with respect to ( 1/ 2( Lσ I rr2 ) , the peak energy stored in the leakage
inductance as the diode snaps off, is also plotted in Fig.7-6. At the end of the current oscillation, the
energy stored in CS is equals to:
WCS = 1/ 2 CS Vd2
(7-13)
This is dissipated in the diode at the next turn-on of the diode. Assuming a instantaneously turn-on
of the diode, the total energy dissipated in the diode and the snubber resistance is given by the
following:
C
1
1
Wtot = WR + WCS = [ Lσ I rr2 + CS Vd2 ] = Lσ I rr2 (1 + 2 S )
(7-14)
2
2
Cbase
The normalized Wtot is also plotted in Fig.7-6 as a function of the normalized CS. It can be seen from
Fig.7-5 that the maximum voltage decreases only slightly by increasing CS beyond Cbase. However,
the total energy dissipation increases linearly with CS. Therefore, a snubber capacitor with CS in a
range close to Cbase would be used. Once CS has been selected, RS = Ropt can be obtained directly
from Fig.7-6. In this analysis, it is assumed that the reverse-recovery current of the diode snaps off
instantaneously. In practice, the diode reverse-recovery current can be assumed to decay
exponentially. This can be accommodated in the equivalent circuit in Fig.7-2 by adding a timevarying current source. This analysis can be carried out by computer simulation, and the result
shows that the snubber design remains essentially the same [8].
68
Chapter 7
Snubber Circuits
Wtot
( Lσ I rr2 ) / 2
3.0
Vmax
Vd
3
2.41
2.0
Vmax
for RS , RS ,opt
Vd
CS = Cbase
1
1
RS I rr
Vd
1
R
2 S
Rbase
0
Fig.7-5 Maximum over-voltage across the
diode as a function of the snubber
resistance for a fixed value of the snubber
capacitance.
WR
( Lσ I rr2 ) / 2
RS ,opt
Rbase
0
1
2.0 CS / Cbase 3.0
Fig.7-6 Snubber energy loss and the maximum
diode voltage for the optimum value of the
snubber resistance RS as a function of the snubber
capacitance CS
7.4 Snubber circuits for Thyristors
There are several versions of thyristors as mentioned before with turn-off capability relevant to
the FCATS technology. The thyristor should be protected against the reverse-recovery current
which is generated when they are reversing biased may result in unacceptably large over-voltage
because of the series inductance if snubbers are not used. In the previous section, it was shown that
the equivalent circuit for a step-down DC-DC converter could be used to analyze the diode overvoltage snubber in any converter. The equivalent circuit is used here for a three-phase linefrequency thyristor converter, in which the AC-side inductance shown in Fig.7-7(a), are due to line
reactance’s plus any transformer leakage inductance. The DC side is represented by a current source
where id is assumed to flow continuously. It is assumed that thyristors Th1 and Th2 have been
conducting and that thyristor Th3 is gated on at delay angleα, as shown in Fig.7-7(b). The current id
will commutate from thyristor (connected to phase a) to thyristor Th3 (connected to phase b). The
voltage vab is responsible for the commutation of the current. The sub-circuit consisting of Th1 and
Th3 is shown in Fig.7-7(c) with Th3 on and Th1 off and at its reverse recovery time at ω t1 , with iσ =
Irr. The voltage source in the circuit of Fig.7-7(c) can be assumed to be a constant DC voltage with
a value of vab at ω t1 because of the slow variation of 50Hz voltages compared to the fast voltage
and current transient in this circuit. The snubber voltage and current waveforms will be identical to
those described in Fig.7-3. To discuss the design of the snubber, a worst-case line impedance of 5%
is used as explained in the previous section, and Equation (7-15) becomes:
(0.05 VLL )
xC = ω LC =
(7-16)
3 Id
where VLL is the rms line-line voltage and Id is the load current. For the worst-case design, the
voltage source in Fig.7-7(c) will have its maximum value of 2 VLL which corresponds to α = 90.
Here the reverse-recovery time is assumed to be 10µs. Thus, during the current commutation,
assuming that the commutation voltage has a constant value of 2 VLL , the di/dt through the
thyristor Th1 is given by the following:
69
Chapter 7
Snubber Circuits
di
2VLL
=
dt
LC
(7-17)
6 VLL trr I d
⎛ di ⎞
I rr = ⎜ ⎟ trr =
= 0.09 I d
0.1 VLL
⎝ dt ⎠
(5-18)
and therefore,
where trr = 10µs.
As was discussed in the previous section, CS = Cbased is close to an optimum value. Relating to
Fig.7-7(c) and Fig.7-2(a) and the Equation (5-3) gives by the following Equation:
Cbase
⎛I ⎞
= LC ⎜ rr ⎟
⎝ VLL ⎠
2
(7-19)
Substituting Lc from the Equation (7-16) at ω = 377 and Irr from the Equation (7-18) into the
Equation (7-19) yields:
CS = Cbase ( µ F) = 0.6 I d / VLL
(7-20)
RS =Ropt can be obtained from Fig.7-6. Here, assuming the normalized RS = Ropt = 1.3 Rbase , and the
value Rbase = 2 VLL / I rr , it will give, using the Equation (7-18).
RS = Ropt = 1.3 2 VLL / I rr = 20VLL / I d
(7-21)
In order to estimate the loss in each snubber, the voltage waveforms across a thyristor having a
worst-case trigger angle of α = 90. It can be shown that the total energy loss in each snubber equals:
Wsnubber = 3 CS VLL2
(7-22)
or using the Equation (7-20) gives:
Wsnubber = 1.8 ×10−6 I d VLL
(7-23)
If the power of the three-phase converter kVA is S, then at 50Hz, each snubber has a power loss
equals:
Psnubber (in watts) = 10−4 S
(7-24)
A similar procedure can be followed for any value of trr and AC-line inductance. A conservative
design may require CS to be big than Cbase, and therefore RS would be smaller than the value found
above. In that case, the snubber losses would be higher since they are proportional to CS [8], [15],
[31].
70
Chapter 7
Snubber Circuits
P
RS
RS
Th1
van
vbn
vcn
-
~
~
~
CS
RS
Th3
Th5
CS
CS
LC
+
A
+
id
B
LC
+
C
LC
50 Hz
RS
Th4
RS
Th6
CS
RS
Th2
CS
CS
(a)
α
van
vbn
0
ωt
ωt1
vbn-van = vab
(b)
iLc
P
2LC
~
vab (ω.t1 )
CS
Th1 after
recovery
= 2VLL
ith1
RS
Th3 on
A
(c)
Fig.7-7 Turn-off snubbers for Thyristors in a three-phase line-frequency converter circuit: (a)
three-phase line-frequency converter, (b) trigger time, and (c) the equivalent circuit.
7.5 Need for snubber circuits for the transistor
Snubber circuits are used to protect the transistors, which are used in power applications like
FCATS systems and HVDC, by improving their switching trajectory. There are three basic types of
snubbers:
• Turn-off snubbers.
• Turn-on snubbers
• Over-voltage snubbers.
To explain the need of snubber, a step down converter without snubbers is shown in Fig.7-8(a)
where the stray inductances in the various parts of the circuit are shown explicitly. For the purpose
of illustration, a bipolar junction power transistor is used for controlled switch. However, the
discussion that follows applied to controlled switches including MOSFET’s, IGBT’s; GTO’s, and
new devices such as MCT’s. Initially, the transistor is conducting and iC = IO. During the turn-off
71
Chapter 7
Snubber Circuits
switching, at t = t0, the transistor voltage begins to rise, but the current in the various part of the
circuit remain the same until t1, when the freewheel diode begins to conducts, then the transistor
current begins to decrease, and the rate at which it decreases is dictated by the transistor properties
and its base drive. The transistor voltage can be expressed by:
di
vCE = Vd − Lσ C
(7-25)
dt
where Lσ = L1 + L2 + L3 + L4 The presence of stray inductances results in an over-voltage since
diC/dt is negative. At t3, at the end of the current fall time, the voltage comes down to Vd and stays at
that value. During the turn-on transition, the transistor current begins to rise at t4 at a rate dictated
by the transistor properties and the base drive circuit. The equation is still valid, but due to a
positive diC/dt the transistor voltage vCE is slightly less than Vd. Due to the reverse-recovery current
of the freewheel diode iC exceeds IO the freewheel diode reverse-recovery at t5 and the voltage
across the BJT decrease to zero at t6 at a rate dictated by the device properties (see Fig.7-8(c)).
+
L2
L5
Io
Vd
_
t5
ic
L1
Id
t6
Idealized
Switching loci
t0
Turn-off
t1
Turn-on
L3
Cd
L4
T
t4
t3
Vd
(a)
vEC
(b)
Lα
diC
dt
Lα
diC
dt
(c)
Fig.7-8 (a) A step-down converter circuit with stray inductance shown explicitly with (b)
associated switching trajectory and (c) the current and voltage waveforms during turn-on and
turn-off.
These switching waveforms can be represented by switching loci as shown in Fig.7-8(b). The
dotted lines represented idealized switching loci both for turn-on and turn-off, assuming zero
inductances and no reverse-recovery current through the diode. They show that the transistor
72
Chapter 7
Snubber Circuits
experiences high stresses at turn-on and turn-off when both its voltage and current are high
simultaneously, thus causing a high instantaneous power dissipation. Moreover, the stray
inductances result in over-voltage beyond Vd, and the diode reverse-recovery current causes beyond
IO. If necessary, snubber circuits are used to reduce these stresses. An important assumption that
simplifies the snubber circuit analysis is that the transistor current change linearly in time with a
constant di/dt, which is only dictated by the transistor and its base drive circuit. Therefore, di/dt,
which may be different at turn-on and turn-off, is assumed not to be affected by the addition of the
snubber circuit. This assumption provides the basis for a simple design procedure for a laboratory
prototype. The final design may be somewhat different depending on what is revealed by laboratory
measurements on the prototype circuit [8], [16], [17], [31].
7.6 Turn-off snubber:
The main aim of a turn-off snubber is to provide a zero voltage across the transistor in order to
avoid the turn-off problems. This can be approached by connecting a RCD network across the BJT
as shown in Fig.7-10(a). Here, the stray inductances are ignored initially for ease of explanation.
Prior to turn-off, the transistor current is IO and the transistor voltage is essentially zero. At turn-off
in the presence of this snubber, the transistor current iC decreases with a constant di/dt and
( I O − I C ) flows into the capacitor through the snubber diode DS. Therefore, for a current fall time of
tfi, the capacitor current can be written as:
iCS = I o t fi / t , 0 < t < 1
(7-26)
where iCs is zero prior to turn-off time at t = 0 . The capacitor voltage, which is the same as the
voltage across the transistor when DS is conducting, can be written as:
vCS = vCE =
1
CS
∫
t
0
iCS dt =
Io t 2
2 CS t fi
(7-27)
The Equation (7-27) is valid during the current fall time so long as the capacitor voltage is less
than or equal to Vd. The equivalent circuit that represents this condition is shown in Fig.7-10(b).
The voltage and the current waveforms are shown in Fig.7-9(c) for three values of the snubber
circuit capacitance CS [19], [20].
For a small value of capacitance, the capacitor voltage reaches Vd before the current fall time is
over. At that time, the freewheel diode Df turns on and clamps the capacitor and the transistor to Vd
and iCs drops to zero due to dvCS / dt is equal to zero. The next sets of waveforms in Fig.7-9(c) are
drawn for a value of CS = CS1, which causes the capacitor voltage to reach the Vd exactly at the
current fall time tfi. In this case; the value of the capacitor CS1 can be calculated by substituting t = tfi
and vCS = Vd in the Equation (7-27) and is given by:
CS 1 =
Io t f i
(7-28)
2 Vd
For large snubber capacitance with CS > CS1, the waveforms in Fig.7-9(c) show that the
transistor voltage rises slowly and takes longer than tfi to reach Vd. Beyond tfi, the capacitor current
equals IO and the capacitor and the transistor voltages rise linearly to Vd. The turn-off switching loci
with the three values of CS used in Fig.7-9 are shown in Fig.7-10 [8].
73
Chapter 7
Snubber Circuits
+
iDf
Io
Df
+
Df
Vd
DS
Vd
Io - iC
RS
T
CS
_
_
iCS
iC
(a)
CS
(b)
iC
iC
Io
iC
IO
iDf
tfi
Vd
iDf
tfi
Vd
iDf
tfi
Vd
vCs
Cs small
Cs = Cs1
Cs Large
(c)
Fig.7-9 (a) turn-off snubber circuit, (b) its equivalent circuit during the transient and (c) current
and voltage waveforms during the turn-off transient. (The shaded areas in Fig.7-9 (c) represent
the charge put on the snubber capacitance during turn-off that will be dissipated in the power
switching device at the next turn-off.)
To optimize the snubber design it is necessary to consider the transistor turn-on in the
presence of turn-off the snubber. To understand the transistor behavior at turn-on, initially it is
assumed that the resistor is essentially zero, which means a pure capacitance without RS and DS is
used as the turn-off snubber, as shown in Fig.7-11(a) The presence of CS causes the turn-on current
to increase beyond IO and the freewheel diode reverse-recovery current. It is still assumed that
diC/dt is constant during turn-on. The shaded area in Fig.7-11(a) represents the charge of the
capacitor that is discharged into the transistor. This charge is equal to the area of one of the shaded
areas in Fig.7-9(c) depending on the value of CS used. In the absence of the snubber capacitor CS,
the transistor voltage would have fallen almost instantaneously (since the voltage fall time is usually
quite small) as shown by the dashed line in Fig.7-11(a). Hence, the energy dissipated in the
74
Chapter 7
Snubber Circuits
transistor during the voltage turn-on would have been small. The presence of CS lengthens the
voltage fall time so the additional energy is dissipated in the transistor. The additional energy
dissipated in the transistor during the capacitor discharge time can be expressed by the following
Equation:
∆WQ = ∫
t2
tri + trr
iC vCE dt = ∫
t2
tri + trr
iCS vCE dt + ∫
t2
tri + trr
I o vCE dt
(7-29)
The first term, in the right-hand side, equals the energy stored in the capacitor, which is dissipated
in the transistor at turn-on. However, there is additional energy dissipated in the transistor as
expressed by the second term in the Equation (7-29).
iC
RBSOA
IO
Cs = 0
Cs small
Cs = Cs1
Cs Large
Vd
vCE
Fig.7-10 Switching trajectory during turn-off with various values of snubber capacitance CS.
The dissipated energy is normally larger. This energy dissipation is due to the lengthening of the
voltage fall time brought about by the presence of CS. The transistor turn-on waveforms in presence
of the snubber resistance RS is shown in Fig.7-11(b). Here, unlike the pure capacitively snubbed
transistor, the voltage can be assumed to fall almost instantaneously. Therefore, no additional
energy dissipation due to the snubber occurs in the transistor at turn-on. The capacitor energy,
which is dissipated in the snubber resistor, is given by:
1
WR = CS Vd2
2
(7-30)
The snubber resistance in Fig.7-11(b) should be chosen so that the peak current through it is less
than the reverse-recovery current Irr of freewheel diode, which can be formulated as follows:
Vd
< I rr
(7-31)
RS
The circuit designer usually attempts to limit Irr to 20% I O or less so that the Equation (7-31)
becomes approximately:
Vd
= 0.2 I rr
(7-32)
RS
75
Chapter 7
Snubber Circuits
Discharge
of CS
+
Io
Df
vCE
IO
Vd
Vd
_
CS
T
0
iC
0
tri
t2
tri+trr
(a)
vCE
trr
+
Io
Df
Vd
T
_
Vd
0
RS
iDf
DS
iC
CS
0
Irr
IO
Irr
vCs (t = 0) Vd
=
RS
RS
(b)
Fig.7-11 Effect of the snubber capacitance CS on the turn-off transient without (a) snubber
resistance RS and (b) with the resistance.
Based on the above assumption, comparing of Fig.7-11(a) with Fig.7-11(b) indicates that
including the resistance RS has the following beneficial effect during the transistor turn-on:
• All the capacitor energy is dissipated in the resistor which is easier to cool than the transistor.
• No additional energy dissipation occurs in the transistor due to the turn-off snubber.
• The peak current that the transistor must conduct is not increased due to the turn-off snubber.
In order to support choosing an appropriate value for CS, the energy dissipated in the transistor
during turn-off and the energy dissipated in the snubber resistance RS during turn-on are plotted as
functions of the CS in Fig.7-12. Based on the previous assumption, these plots are independent of RS
76
Chapter 7
Snubber Circuits
and there is no additional energy dissipation in the transistor during turn-on due to the presence of
the turn-off snubber. CS should be chosen based on the following issues:
• Keeping the turn-off switching locus within the reverse-bias safe operation area.
• Reducing the transistor losses based on its cooling consideration.
• Keeping the sum (shown as a dashed line in Fig.7-12) of the transistor turn-off energy
dissipation and snubber resistance energy dissipation low.
Fehler! Es ist nicht möglich, durch die Bearbeitung von Feldfunktionen Objekte zu
erstellen.
Fig.7-12 Turn-off energy dissipation in the power switching device and the
snubber resistance RS as a function of the snubber capacitance CS.
Having made initial selection of RS based on the Equation (7-32) and CS based on design tradeoff, previously discussed, the designer must ensure that the capacitor has sufficient time to
discharge down to low voltage, 0.1Vd, during the minimum on-state time of the transistor in order
that the turn-off snubber be effective at the next turn-off interval [8].
During the on-state of the transistor, the capacitor discharge with a time constant τ C = RS CS
and the capacitor voltage given by:
vCS = Vd e −t /τ C
(7-33)
therefore, discharging vCs down to 0.1Vd requires a time interval of 2.3τc thus:
ton state > 2.3 RS CS
(7-34)
As an example, assuming CS = CS1 (given in the Equation (7-28)) and RS is chosen using the
Equation (7-32). Then the minimum on-state time of the transistor must be six times the transistor
current fall time tfi (see Fig.7-15) [8].
7.7 Over-voltage snubber
The stray inductions were neglected, when the turn-off snubber was described. Hence, there was
no over-voltage. The over-voltages at turn-off due to stray inductances, such as the one shown in
Fig.7-11(a), can be minimized by means of the over-voltage snubber circuit shown in Fig.7-13. It is
assumed here that it is possible to lump all the stray inductances together as indicated in Equation
(7-25). The operation of over-voltage snubber can be described as the follows: initially the
77
Chapter 7
Snubber Circuits
transistor is conducting and the voltage vC ,OV across the over-voltage snubber capacitor equals Vd.
At turn-off, assuming the BJT current fall time to be small, the current through Lσ is essentially IO
when the transistor current decreases to zero, and the output current then free-wheel diode Df. At
this stage, the equivalent circuit is shown in Fig.7-13(b), where the Df, Io combination appears as a
short circuit, and the transistor is an open circuit. Now the energy stored in the stray inductances
gets transferred to the over-voltage capacitor through the diode DOV and the over-voltage ∆VCE
across the transistor (in the state, the capacitor COV and the transistor have the same voltage) can be
obtained by replacing the recharged capacitor with its equivalent circuit as shown in Fig.7-11(c).
Using the energy consideration and noting that ∆VC ,OV = ∆VCE gives:
(COV ∆VCE2 ,max )
2
Lδ
+
( L σ I o2 )
=
2
Lδ
Io
Df
ROV
Vd
(7-35)
DOV
+
Cd
ROV
Vd
DOV
_
COV
T
COV
_
(a)
(b)
Lσ COV
kVd
Lδ
+
_
ROV COV
Discharging
iLσ
∆VCOV
Vd
Charging
1Vd
COV
+
Vd
_
vCE
Without COV
0.0
tfi
0.0
With COV
(c)
(d)
Fig.7-13 (a) Over-voltage snubber, (c, b) its equivalent circuit during transient turn-off, (d) the
collector-emitter voltage with and without the snubber.
The equation above shows that a large value of COV will minimize the over-voltage ∆VCE,max.
Once the current through the Lσ has decreased to zero, it can reverse its direction due to the diode
DOV, and the over-voltage on the capacitor decreases to Vd through the resistor ROV. The capacitor
78
Chapter 7
Snubber Circuits
discharge time constant ROV COV should be small enough so that the capacitor voltage has decayed
approximately to Vd prior to the next turn-off of the transistor. To support the estimation of the
proper value of COV, the circuit waveforms with and without the over-voltage snubber are shown in
Fig.7-13(d). The observed over-voltage of k Vd without the over-voltage snubber is used to estimate
Lσ as given by the following Equation:
k Vd = Lσ I o / t fi
(7-36)
If an over-voltage, for example, ∆ VCE , max = 0.1 Vd is acceptable, then using Equation (7-35) and
substituting for Lσ from the Equation (7-36) yields:
COV =
(100 I o t f i )
Vd
, (∆ VCE , max = 0.1 Vd )
(7-37)
In terms CS1which given by (7-28), COV from Equation (7-37) can be rewritten as:
COV = 200 k CS 1
(7-38)
This shows that the substantially larger capacitance is needed for over-voltage protection
compared to the values used in the turn-off snubber, which are on the order of CS1. It can be shown
that even with a large value of COV the dissipated energy in ROV is of the same order as the energy
dissipated in the resistor of the turn-off snubber. Both the turn-on and the over-voltage protection
snubbers should be used simultaneously [8], [19], [20].
7.8 Turn-on snubber
There is a large FBSOA (Forward Biased Safe Operating Area) of the most controlled switches
including BJT’s, MOSFET’s, GTO’s, and IGBT’s. So the turn on snubbers are only used to reduce
turn-on switching losses at high switching frequencies and for limiting the maximum diode reverse
recovery current. Turn-on snubber works by reducing the voltage across the switch (transistor) as
the current builds up. A turn-on snubber can be in series with the transistor as shown in Fig.7-14(a)
or in series with the freewheel diode as shown in Fig.7-14(b). In both circuits the turn-on and turnoff switching waveforms across the transistor and the freewheel diode are identical. The reduction
in the voltage across the transistor during turn-on is due to the voltage drop across LS .This
reduction is given by:
∆VCE = LS I O / tri ⇒ LS = ∆VCE tri / I O
(7-39)
where, tri is the current rise time as shown in Fig.7-14(c) for small values of LS. For such small
values, di/dt is dictated only by the transistor and its base drive circuit and is assumed to be the
same as without the turn-on snubber. Therefore, the diode peak reverse-recovery current is also the
same as without the turn-on snubber. If it is important to reduce the diode peak reverse-recovery
current, it can be achieved with a large value of LS as shown by the waveforms in Fig.7-14(d). Here
the current rate of rise is di/dt = Vd/LS and the voltage across the transistor is almost zero during the
current rise time. During the on-state of the transistor, LS conducts Io. When the transistor turns off,
the energy stored in the snubber inductor ( LS I o2 ) / 2 will be dissipated in the snubber resistor RLS.
The snubber time constant is τ L = LS / RLS . When selecting RLS, the following two factors must be
considered. First, during transistor turn-off, the turn-on snubber will generate an over-voltage across
the transistor which is given by:
79
Chapter 7
Snubber Circuits
∆VCE ,max = RLS I o ⇒ RLS =
(7-40)
IO
Io
Df
+
∆VCE ,max
+
Io
DLS
Io
LS
RLS
Vd
Vd
DLS
LS
Df
RLS
_
_
T
(a)
(b)
LS small
LS
Vd
T
LS Large
iC
di
dt
iC
vCE
vCE
trí
trr
(c)
(d)
Fig.7-14 Turn-on snubber circuit (a) in series with the power switching device or (b) in
series with the free-wheel diode, (c) The power switching device voltage and current
waveforms for small value of Lσ and (d) for Large values of Lσ.
Second, during the off-state the inductor current must decay to a low value, for example 0.1IO,
so that the snubber can be effective during the next turn-on. Therefore, the minimum interval for the
off state of the BJT should be:
toff state > 2.3( LS / RLS )
(7-41)
Thus a large inductance will result in lower turn-on voltage and lower turn-on losses. But it will
cause over-voltage during turn-off, lengthen the minimum required off-state interval, and result in
higher losses in the snubber. Therefore, LS and RS must be selected based on the above design tradeoffs following a procedure similar to that described for the turn-off snubber. Since the turn-on
snubber inductance must carry the load current, which makes this snubber expensive, it is rarely
used alone.
However, if the turn-off snubbers are to be used in transistor bridge configurations, then turn-on
snubbers must be used. It is possible to use all snubbers simultaneously or in any other combination.
80
Chapter 7
Snubber Circuits
A circuit configuration that include all three snubber but having a reduced components count (the
Undeland snubber) is shown in Fig.7-15 [8], [14], [15], [18], and [19].
DF
IO
COV
CS
Cd
DS
RS
LS
Fig.7-15 A modified circuit with an over-voltage snubber, a turn-on snubber, and turnoff snubber; the Undeland snubber for step-down converter.
7.9 GTO snubber circuit consideration
It was pointed out that the GTO’s almost always need snubber circuit in its applications like
FACTS systems. While snubbers for GTO’s have the same configurations as for other controlled
switches (see Fig.7-16), the large voltages and currents found in GTO circuits place additional
requirements on the snubber circuits. Some of additional considerations are discussed below:
•
A GTO is capable of turning off significantly larger current compared to its rms or average
current capability. The maximum controllable current for a given GTO in the circuit of
Fig.7-16, depends on the turn-off snubber capacitance CS. This dependence is related to the
maximum rate of change in the increase in the anode-cathode voltage at turn-off. Exceeding
this maximum dv AK / dt |max would cause regenerative of the GTO back into the on state due
to large displacement currents. Now dv AK / dt is inversely proportional to CS according
to dvAK / dt = I o / CS . So for a given dv AK / dt , the larger CS is, the larger Io can be. It is
assumed that the maximum controllable anode current given on the GTO specification sheet
is not exceeded. A large CS, however, results in a higher overall switching losses and large
current through the GTO at turn-on. Therefore, the capacitance CS should be just sufficient
to turn-off the maximum current dictated by particular application.
•
The capacitor CS should have a low internal inductance and large peak current rating. In
practice, this may require parallelizing of many capacitors to achieve these required
properties for CS.
The turn-off snubber diode DS needs to carry the entire load current for a short time. Its
average current is very low, since its dynamic forward voltage at turn-on must be low, but
often a diode with a large average current rating is chosen.
The turn-off resistance RS must be selected based on trade-off between maximum additional
discharge current into the GTO and requirement on the minimum on-state time of the GTO
to discharge CS so it can be properly operated during the next turn-off, as was described in
•
•
81
Chapter 7
Snubber Circuits
the turn-off snubber section. There is a considerable power loss in RS and therefore it may
require mounting on heat sink.
+
IO
Df
Vd
RON
Cd
Turn-on
snubber
LON
DON
Turn-off
snubber
RS
GTO
DS
_
CS
Fig.7-16 Step-down converter circuit using a GTO as the switching
device with turn-on and turn-off snubbers
It has already been described why the stray inductance in the turn-off snubber current loop
should be as small as possible. To achieve this objective, the snubber components should be
mounted as close to the GTO as possible. The design considerations for turn-on snubber for the
GTO are similar to those described in the turn-on snubber section [8], [15], [19], and [20].
7.10 IGBT Snubber design
7.10.1 Over-voltage causes and their suppression
Due to the high switching transistors of IGBTs at turn-off or during D1 (FWD Freewheel Diode)
reverse recovery, the current change rate (di/dt) is very high. Therefore, the circuit wiring
inductance to the module can cause a high turn-of surge voltage V = L (di / dt ) .
For example, using IGBT’s waveform at turn-off will introduce the causes and the method of
their suppression, as well as illustrate a concrete example of a circuit (using an IGBT and FWD
together). Fig.7-17 shows the principle of a half-bridge, used as a test circuit, and the resulting
voltage and current waveforms when switching IGBT1. The stray inductance LS, shown as a
concentrated circuit element, represents all distributed inductances within the commutation loop.
When turning off the upper IGBT1, its current, whose magnitude is maintained by the inductive
load, commutates into the diode D2 of the lower module for free-wheeling. IGBT1 takes over
voltage up to the value of Ed and then, during the following fall time, the current is reduced through
IGBT1 and at the same time is built up in diode D2 (see Fig.7-18). The current rate-of-rise di/dt,
82
Chapter 7
Snubber Circuits
dependent on current and voltage as well as temperature, is typically in the range of 3 − 6kA / ms ;
values of up to 10kA / ms can be reached under short circuit conditions. Due to the falling current a
voltage drop of (− LS (dioff / dt )) occurs across the stray inductance LS.
LS
IC1
IGBT1
D1
VGE1
VCE1
Load
Ed
ID2(=-IC2)
LO
D2
IGBT2
VD2(=-VCE2)
RO
Fig.7-17 Test chopper circuit.
(Ed: DC supply voltage, LS: main circuit wiring inductance, RO, LO: The load.)
If IGBT1 is turned on again, the load current commutates back from the branch of the diode D2
and is taken over again by IGBT1. Due to the rising current in this path a voltage drop of
LS (dion / dt ) occurs over the stray inductance. This reduces the DC link voltage as long as diode D2
is still conducting. No voltage is taken over, until the peak of the reverse recovery current is
exceeded. If this point is reached, it depends strongly on the recovery behavior of the diode, with
which rate-of-rise the current goes through zero and with which rate-of-rise diode and anti-parallel
IGBT must take over blocking voltage. High stray inductances and/or a snappy diode behavior may
lead to considerable overvoltage spikes VCESP at this point. The VCESP: the turn-off surge voltage
peak can be calculated as follows:
VCESP = Ed + (− LS (diC / dt ))
(7-42)
where: diC / dt is the maximum collector current change rate at turn-off. If the voltage VCESP
exceeds the IGBTs C-E (VCES) rating, then the module will be destroyed [21], [22], [23], [24], and
[25].
7.10.2 Over-voltage suppression methods
Several methods for suppressing turn-off surge voltage (as discussed before in this chapter). The
sources of the over voltages are listed below:
• Control the surge voltage by adding a protection circuit (snubber circuit) to the IGBT. Use a
film capacitor in the snubber circuit. Place it as close as possible to the IGBT in order to
bypass high frequency surge current.
• Adjust the IGBT driver circuits (-VGE) or RG in order to reduce the di/dt value.
• Place the electrolytic capacitor as close as possible to the IGBT in order to reduce the
effective inductance of the wiring. Use a low inductance capacitor.
83
Chapter 7
Snubber Circuits
• To reduce the inductance of the main as well as snubber circuits wiring, and use thicker and
short wires. It is also very effective to laminate the copper bars in wiring [21], [22], and [23].
IGBT1 turn-on
IGBT1 turn-off
0
VGE
VIGBT1
VCEPS1
VCEPS2
I IGBT1
0
I D2
0
VIGBT2
Fig.7-18 IGBT Switching waveforms during the turn-off and turn-on processes
7.10.3 Type of IGBT snubber circuits and their features
Snubber circuit is a supplementary circuit used in the converter circuit to reduce stress put on
the power semiconductor device. The ultimate goal of the snubber circuit is to improve the
transient waveform. The snubber circuit suppresses over-current or over-voltage or improve dv/dt
and di/dt to ease the transient waveform to reduce stress on the power semiconductor switching
device. Snubber circuit can be divided into those connected to each IGBT and those connected in
between the DC power supply and bus and ground. The first types of circuits include RC snubber
circuits, charge and discharge RCD snubber circuits, and discharge-suppressing snubber circuits;
while the second type includes C snubber circuit and RCD snubber circuits [21], [22], [23], and
[24].
7.10.3.1 RC Snubber circuit
The RC snubber circuit, which is presented in Fig.7-19(a), is effective in turn-off surge voltage
and suitable for chopper circuit. It is also effective for oscillation by parasitic reactance and dv/dt
noise. However, when it is applied in large capacity IGBT, resistance for the snubber must be set to
low to dissipation heat, so it has the disadvantage of worsening loading conditions at turn-on. Loss
at the snubber itself is quite large, so it is not suitable for high frequency. In very large capacity
IGBT circuit, it is better to use small snubber “RC snubber circuit” along with main snubber
“discharge-suppressing RCD snubber circuit”. When used together, it helps parasitic oscillation
control of the main snubber circuit loop. Main applications of this snubber circuit include arc
welder and switching power supply [22].
84
Chapter 7
Snubber Circuits
7.10.3.2 Charge and discharge RCD Snubber circuit
This snubber suppresses over-voltage at turn-off to reduce switching losses at turn-off, and its
effectiveness in surge voltage suppression is about average (see Fig.7-19(b)). The snubber capacitor
is completely discharge at turn-on, and it is fully recharged at turn-off. Unlike the dischargesuppressing RCD snubber circuit below which acts as a clamp, this circuit reduces IGBT dv/dt
during turn-off. As such, soft switching possible, and IGBT loss is reduced.
Since the structure of this circuit is snubber diode added to an RC snubber, snubber resistance
can be increased, which alleviate the loop problem at turn-on. It is effective chopper applications,
which use large current and low DC link voltage. Its advantage also includes no oscillation at its DC
link voltage. The power loss due to the resistance is as follows:
P = 1/ 2[( L I o2 f ) + (CS Ed2 f )]
(7-43)
where L: the wiring inductance of the main circuit, IO: the collector current at IGBT turn-off, CS: the
capacitor of the snubber circuit, Ed: DC supply voltage and f: the switching frequency.
However, losses from this circuit (mostly from snubber resistance) are significantly larger than
the same in discharging suppressing RCD snubber circuit. As such, it is not suitable for high
frequency switching applications. There are a lot of turn-off losses with bridge configuration. The
disadvantages of this snubber circuit are relatively many parts and difficulties in selecting parts
[21], [22].
7.10.3.3 Discharge-suppressing RCD snubber circuit
The functions of this circuit as similar to those of voltage clamp snubber circuit (see Fig.719(c)). Snubber capacitor is charged to the DC link voltage while the IGBT is in conduction, and
VCE rises rapidly when IGBT is turn-off. Due to the stray inductance of the DC loop, VCE rapidly
rises above the DC link voltage, so the snubber diode is forward biased conduction, and the snubber
begins operation.
The energy stored in the stray inductance moves to snubber capacitor, which absorbs the energy
without a rise in voltage. It has the advantages of small oscillation in DC link, and it is most
practical in mid-to-large current applications. It is effect on turn-on voltage transient is neither large
nor small. It is ideal for high frequency switching as its losses from the snubber circuit is small. The
losses can be calculated with the following Equation:
P = 1/ 2[( L I o2 f )
(7-44)
where L: the wiring inductance of main circuit, IO: the collector current at IGBT turn-off and f: the
switching frequency.
It is disadvantages are that it has many necessary parts and is less than effective on turn-off
surge voltage. It is often used in inverters [21], [22].
7.10.3.4 C snubber circuit
This is the simplest snubber circuit, so it has the advantages of suppressing over-current at
minimum cost (see Fig.7-19(d)). It is effective in mid-to-low current, low power applications, and
as the power level increases, it becomes more likely for the circuit to oscillation as the snubber
capacitor and the main circuit inductance from LC resonance circuit. It is often used in inverters
[21], [22].
85
Chapter 7
Snubber Circuits
P
P
RCD
P
RC
RCD
N
N
N
(a)
(b)
(c)
P
C
RCD
N
N
(d)
(e)
Fig.7-19 Schematic type of individual snubber circuits: (a) RC snubber circuit, (b) Charge
discharge RCD snubber circuit, (c) Discharge suppressing RCD snubber circuit, (d) C snubber
circuit and (e) RCD snubber circuit.
7.10.3.5 RCD snubber circuit
The RCD snubber circuit operates in the same manner as the C snubber circuit, but it is different
in that it operates during turn-off switching (see Fig.7-20(e)). It is a circuit that solved oscillation of
the C snubber circuit by using the fast recovery diode. Energy that stored in DC loop inductance
moves to the capacitor while the IGBT turns off. The snubber diode prevents oscillation from taking
place. The charge from the capacitor is discharged through the snubber resistor. (RC time constant
should be about 1/3 of the switching cycle. ( τ = T / 3 = 1/ 3 f ). This circuit reduces turn-off voltage
transient directly. Switching waveform is significantly smoother and snubber loss is small. Effect
on the turn-on voltage transient is fine, and it has the advantage of stable wave as the snubber diode
blocks oscillation. It is practical in medium current range, but operation in large capacity IGBT,
parasitic inductance increases to present problems in controlling over-voltage. In such large current
applications, discharge-suppressing RCD snubber circuits are generally used. The functions of the
discharge-suppressing RCD snubber circuit are similar to the functions of the RCD snubber circuits,
86
Chapter 7
Snubber Circuits
but the discharge-suppressing RCD snubber circuit has the advantage of small loop inductance as it
is attached to the collector and the emitter of each device. This circuit cannot use in low inductance
snubber capacitors designed to attached directly to the IGBT, and the blocking diode added to
protection circuit can increase the total snubber inductance. Furthermore, if the recovery
characteristics of the diode are not good, VCE over-shoot and dv/dt at either sides of the IGBT/diode,
or the output voltage can oscillate. The Turn-off mechanism is nearly the same as that of the
discharge-suppressing RCD snubber circuit [21], [22].
7.10.4 Discharge-suppressing RCD snubber circuit design
The discharge suppressing RCD snubber can be considered the most suitable snubber circuit for
IGBTs. Basic design methods for this type of circuit will be explained in the following points.
7.10.4.1 A Study of applicability
Fig.7-20 shows the turn-off locus waveform of an IGBT in a discharge-suppressing RCD
snubber circuit. Fig.7-21 shows the IGBT current and voltage waveforms at turn-off. The dischargesuppressing RCD snubber circuit is activated when the IGBT C-E voltage starts to exceeds the DC
supply voltage [20]. The dotted line diagram in Fig.7-20 shows the ideal operating locus of an
IGBT. In an actual application, the wiring inductance of the snubber circuit or a transient forward
voltage drop in the snubber diode can cause a spike voltage at IGBT turn-off.
This spike voltage causes the sharp-concerned locus indicated by the solid line in Fig.7-21. The
discharge-suppressing RCD snubber circuits’ applicability is decided by whether or not the IGBTs
operating locus is within the RBSOA (Reverse Biased Safe Operating Area) at turn-off. The spike
voltage at IGBT turn-off is calculated as follows:
VCESP = Ed + VFM + (− LS (diC / dt ))
(7-45)
where: Ed: DC supply voltage, LS: the snubber circuit wiring inductance, VFM: the transient forward
voltage drop and diC / dt maximum collector current change rate at IGBT turn-off. The reference
values for the transient forward voltage drop in snubber depending on diode are: 600 V class: 20 to
30V, and 1200V class: 40 to 60V [18], [21], [22].
IC
(Pulse)
VCE
RBSOA
iC
I0
VCEP
VCESP
VCES
VCESP
VCES
VCE
Fig.7-20 Turn-off locus waveform of IGBT.
Fig.7-21 Voltage and current waveforms at turnoff. IGBT.
87
Chapter 7
Snubber Circuits
7.10.4.2 Calculating the capacitance of the snubber circuit capacitor
The necessary capacitance of a snubber capacitor CS is calculated as follows:
CS = L I o2 /((VCEP − Ed ) 2 )
(7-46)
where: L: the main circuit wiring inductance, VCEP: the snubber capacitor peak voltage, Io: the
collector current at IGBT turn-off and Ed: DC supply voltage. The VCEP must be limited to less than
or equal to the IGBT C-E withstand voltage [21].
7.10.4.3 Calculating snubber resistor
The function required of snubber resistance RS is to discharge the electric charge accumulated in
the snubber capacitor before the next IGBT turn-off. To discharge 98% of the accumulated energy
by the next IGBT turn-off, the snubber resistance must be satisfying the following inequality:
RS ≤ 1/(2.3 CS f )
(7-47)
where: f is the switching frequency. If the snubber resistance is set too low, the snubber circuit
current will oscillate and the peak collector current at the IGBT turn-off will increase. Therefore,
the snubber resistance is set in a range below the value calculated in the Equation (5-46).
Irrespective of the resistance, the power dissipation loss P (RS) is calculated as follows [21]:
(7-48)
P ( RS ) = ( L I o2 f ) / 2
7.10.4.4 Snubber diode selection
A transient forward voltage drop in the snubber diode is one factor that can cause a spike
voltage at IGBT turn-off. If the reverse recovery time of the snubber diode is too long, then the
power dissipation loss will also be much greater during high frequency switching. If the snubber
diodes reverse recovery is too hard, then the IGBT C-E voltage will drastically oscillate. The
selecting of the snubber diode that has a low transient forward voltage, short reverse recovery time
and a soft recovery [18], [20], [21].
7.10.4.5 Snubber circuit wiring precautions
The snubber circuits wiring inductance is one of the main causes of spike voltage, therefore it is
important to design the circuit with the lowest possible inductance [21].
88
Chapter 8
Simulation results of three-level VSC snubber circuits
8 Simulation results of three-level VSC snubber circuits
8.1 Introduction
In this chapter, the three-level Voltage-Sourced Converter with the common snubber and the
new suggested snubber circuit will be discussed. Depending on the snubber circuit strategy, which
was studied in the chapter7, recommended and new snubber circuit design will be presented which
are suitable for the three-level Voltage-Sourced Converter. The Study will be supported by
computer simulations and practical tests.
8.2 Common snubber circuit for three level inverters
In the recent years, modern power semiconductor switching devices like GTO’s (Gate Turn-off
Thyristors), IGBT’s (Insulated Gate Bipolar Transistors), and IGCT’s (Integrated Gate Commutated
Thyristors) (discussed in Chapter3), which have the ability to be turned-off via a gate-control, have
attracted more and more attention to be used in very large voltage-sourced inverters, FACTS
(Flexible AC Transmission Systems) devices or modern HVDC systems. These systems with a rated
power of 300MVA and above make it possible to increase the power transmission capacity of the
existing lines or to improve the stability of the today and future power systems. The inverters require
specialized high power gate controlled valves due to their high MVA rated power. At present, the
GTO and the IGCT is still the most practical power switching device available for use in these
circuits. GTO’s with the maximum voltage and maximum current of are now commercially
available. Several of these devices must be connected in series to constitute a single valve for the
above applications. In general, very specific protection systems (snubber circuits as discussed in
Chapter7) should be used to reduce and prevent the stresses, which result from the switching (on-off
and off-on) processes.
The main goal is to keep the power semiconductor device within the SOA (Safe Operation Area)
defined by the manufacturer [26], [27], [31]. Basically, the power semiconductor switching devices,
which can be used in the high-power systems described above, require a series snubber circuit
(inductance) that limits the current-time-divertive di/dt at turn-on, and a parallel snubber circuit
(capacitance) that limits the voltage-time-divertive dv/dt during the turn-off. The two snubber
circuits cannot be considered as disjoint circuits but they must be seen as a coupled auxiliary circuit
design (see Fig.8-1) [28], [29], [30]. In fact, snubber circuits can be divided in three types:
Unpolarized series RC snubbers, polarized RCD snubbers and polarized RLD (which were discussed
in Chapter7). In details, a single-phase schematic of a three level converter is shown in Fig.8-1. The
snubber circuit for the shown partial network of a three level converter uses separate elements and
integrated into separate sub-circuit [31], [32], [33], [34], and [35].
8.3 Double snubber circuit for three level inverters
Here a new snubber circuit configuration will be subjected; the proposed double snubber
circuit uses an additional RC snubber to optimize the damping of the turn-off snubber configuration.
Therefore, the over voltage can be reduced effectively by an optimization of the passive
components and its parameters in the new design. The double snubber makes it possible to reduce
the high peak power of the losses during switching off the semiconductor devices associated with
89
Chapter 8
Simulation results of three-level VSC snubber circuits
the simultaneous maximum of the voltage and the current. In this way, the proposed circuit allows a
minimization of both the over voltage and the switching losses [31], [34], and [36]. The common
snubber circuit shown in Fig.8-1 composes of two standard turn-on RLD snubbers (RON1, LON1 DON1,
RON2, LON2, and DON2), but the double snubber configuration consisting of the same snubber circuit
configuration and an extra RC network connected in parallel to the RCD as shown in Fig.8-2.
Turn-on snubber
Turn-off snubber
LON1
DS1
DON1
RON1
Df1
RS1
GTO1
CS1
Ed
C1
DC1
Df2
GTO2
DS2
RS2
CS2
Rd
The Load
Vd
Df3
DC2
Ed
GTO3
DS3
RS3
CS3
C2
RON2
DON2
Df4
GTO4
DS4
RS4
CS4
LON2
Fig.8-1 Single phase of a three level converter (common snubber circuit).
The RC snubber has several advantages: It allows reducing losses in the circuit and the
switching device. Additionally, the peak voltage is limited. During the charging of CS which is
limiting the dv/dt during the off process, the effective value of RS is essentially close to zero. This is
not the optimum value for RS. In most cases, transient phenomena will cause an essentially higher
over voltage or at least more than necessary [8], [37], [38], [39], [40], [41], [42], [43], and [44].
Table 8.1 shows the total number of protection elements RSµ, CSµ, Ronµ, Lonµ and snubber diodes Dµ
in the common- and proposed double snubber circuit configurations. However, the common
snubber circuit uses fewer elements than the proposed double snubber circuit, where the proposed
double snubber circuit design needs more passive components. Therefore, the implementation of the
90
Chapter 8
Simulation results of three-level VSC snubber circuits
double snubber circuit is favorable, especially, in cases when the optimization of losses and/or over
voltages is of highest interest [28], [30], [42], [43], [44], and [45].
Components
Capacitors
Diodes
Inductors
Resistors
Common Snubber Circuit configuration
4
6
2
6
Double Snubber Circuit configuration
8
6
2
10
Table 8.1 the total number of snubber elements for the different snubber designs. [28], [32].
Double snubber citcuit
LON1
DS1
DON1
RON1
Df1
RS1
GTO1
CS1
Ed
RP1
CP1
C1
DC1
Df2
GTO2
DS2
Rd
RS2
RP2
CS2
CP2
RS3
RP3
CS3
CP3
RS4
RP4
CS4
CP4
The Load
Vd
Df3
DC2
Ed
GTO3
DS3
C2
RON2
DON2
Df4
GTO4
DS4
LON2
Fig.8-2 Proposed double snubber circuit configuration in a three level inverter system.
8.4 An optimized snubber design for three level inverters
The other proposed design is an optimized snubber design which consists of a centre-tapped
inductor (LON1, LON2), a damping resistor (RON), four diodes (DON1, DON2, DS1, DS2), four capacitors
(CS1, CS2, CS3, CS4), and two resistors (RS1, RS2) for each arm of the three-phase inverter system as
shown in Fig.8-3 [8], [28], [36], [47], and [48]. Table 8.2 compares between the number of
components for the common snubber circuit (RLD/RCD) and the new optimized snubber design.
The new optimized design includes only three resistors while the common snubber needs six
resistors [8], [36].
91
Chapter 8
Simulation results of three-level VSC snubber circuits
Components
Capacitors
Diodes
Inductors
Resistors
RLD/RCD Snubber
4
6
2
6
Optimized snubber design
4
4
2
3
Table 8.2 the total number of snubber elements for different snubber designs.
RS1
GTO1
Df 1
Ed
DS1
CS1
C1
Turn-Off snubber
DCD1
GTO2
Df 2
CS2
DON1
Rd
The Load
Vd
LNO1
LNO2
Turn-on
snubber
RON
DON2
Ed
DCD2
Df 3
GTO3
Df 4
GTO4
CS3
C2
DS1
CS4
RS1
Fig.8-3 An optimized snubber design for Single phase three level GTO inverters.
Additionally, the new optimized snubber design allows leaving out two diodes and has the
following advantages:
•
•
•
•
•
Inverters based on the optimized snubber design are suitable for high voltage applications
(like FACTS equipments and HVDC systems) since the voltage sharing between serially
connected power devices is guaranteed.
The number of elements, which are necessary for the optimized snubber design, is less than
those which are needed for the conventional circuit.
The manufacturing cost, the complexity, and also the converter size can be reduced.
The performance of the optimized snubber design in the direction of the over-voltage
protection of the power semiconductors is better than that of conventional snubber circuits.
There are only two resistors used to discharge the capacitors. So the total losses are lower
than those of the conventional design [50], [51], [52], [53], and [54].
92
Chapter 8
Simulation results of three-level VSC snubber circuits
Because of the fewer elements in the snubber circuit, the theoretic reliability will be higher.
Therefore, the implementation of the optimized snubber design is favorable especially in cases of
the optimization of the losses and/or over-voltage is of highest inertest. To analyze the physical
background of the protection performance, some of the basics of different switching operations
have to be discussed. Therefore, the possible switching states of a single phase of a three level
inverter system are listed in Table 8.3. Due to the symmetry, it is sufficient to consider only a
complete cycle of communication process: S0 → S1 → S0 → S−1 assuming that the load current is
flowing.
Switching states
GTO1
GTO2
GTO3
GTO4
S1
S0
S-1
ON
ON
OFF
OFF
OFF
ON
ON
OFF
OFF
OFF
ON
ON
Table 8.3 the three Switching States.
There is an important assumption which simplifies the analysis of the snubber design: The
Thyristor current changes linearly in time with a constant di/dt (the load current is constant, In most
applications the time constant of the load (some milliseconds) is much larger than the switching
time (in the sub-microseconds), so the load current IL remains almost constant during the switching
period), which is only dictated by the GTO-Thyristor current (or other devices) and its base driver’s
circuit. Therefore, di/dt, which may be different for the turn-on and the turn-off processes, is
assumed not to be affected by adding the snubber circuits. Also, it is assumed that the time constant
of the load (milliseconds) is much larger than the switching time (in the microsecond range). So the
load current IL remains almost constant during the switching period. Based on these premises the
relevant transition or commutation phases will be closer analyzed later in this Chapter [50].
Commutation from S0 to S1:
The current transition in this phase is defined by the turning-off of GTO3. After a defined dead
time, GTO1 is turned on. During that period, the load current is flowing through the clamping diode
DC1 and GTO2. The snubber capacitors CS1 and CS4 are charged up to Ed and CS2, CS3 are totally
discharged (see Fig.8-4(a)).
Phase 1: The semiconductor switching device GTO1 starts turning on, so the current flowing
through it begins to increase and the current through the clamping diode DC1 starts to decrease and
the capacitor CS1 begins to discharge in the loop which is formed through GTO1 and RS as shown in
Fig.8-4(b).
Phase 2: In this step, the current through GTO1 continues to increase during the current rise time.
As soon as the clamping diode DC1 ceases to conduct, the load terminal voltage goes up charging
CS3. At the end of this step the capacitor CS1 is completely discharged and the energy which is stored
in LON2 will be dissipated through the loop DON2, RON, DON1, LON1 (see Fig.8-4(c)).
Commutation from S1 to S0:
During this phase GTO1 is turned off and GTO3 is turned on after the necessary dead time. The
initial condition for this period is the final state of the previous commutation. During the turn-off
period of GTO1 the load current is absorbed by CS1 (see Fig.8-5(a)) [50].
93
Chapter 8
Simulation results of three-level VSC snubber circuits
RS1
GTO1
(OFF)
Df 1
Ed
C1
DCD1
GTO2
(ON)
Df 2
Rd
DS1
CS1
Ed
DCD1
CS2
DON1
Ed
DCD2
TheLoad
LNO2
Df 3
GTO3
(ON)
C2
GTO4
(OFF)
Df 4
GTO1 DS1
CS1
(ON)
Df 2
GTO2
CS2
(ON) D
ON1
Rd
LNO1
Vd
Df 1
C1
Ö
RON
LNO1
Vd
LNO2
TheLoad
DON2
Ed
CS3
DCD2
Df 3
GTO3
(OFF)
Df 4
GTO4
(OFF)
C2
CS4
DS2
RS2
RON
DON2
CS3
CS4
DS2
RS2
( a)
(b)
RS1
Ed
Df 1
GTO1
(ON)
Df 2
GTO2
CS2
(ON) D
ON1
C1
DCD1
Rd
Ö
DS1
CS1
LNO1
Vd
Ed
DCD2
TheLoad
LNO2
Df 3
GTO3
(OFF)
Df 4
GTO4
(OFF)
C2
RON
DON2
CS3
CS4
DS2
RS2
(c)
Fig.8-4 Commutation path of the transition form S0 to S1: (a) initial state, (b) phase 1 and (c)
phase-2
Phase 1: In Phase 1, iGTO1 decreases linearly from ILOAD to zero during the current fall time period
and the load current starts to charge CS1 through the turn-off diode DS1.This step ends when iGTO1
reduces to zero see (Fig.8-5(b)).
Phase 2: In Phase 2 the CS1 is completely charged. Finally the complete load current flows through
the clamping diode DC1. After that, GTO3 begins to turn-on after a suitable dead time. In this case
the commutation process is completed (see Fig.8-5(c)) [28], [37].
94
Chapter 8
Simulation results of three-level VSC snubber circuits
RS1
RS1
Df 1
Ed
C1
DCD1
GTO1
(ON)
GTO2
(ON)
Df 2
Rd
DS1
CS1
Ed
TheLoad
DCD2
LNO2
Df 3
GTO3
(OFF)
Df 4
GTO4
(OFF)
C2
DCD1
CS2
DON1
LNO1
Vd
Ed
RON
Df 1
GTO1
(OFF)
Df 2
GTO2
(ON)
C1
Rd
DS1
CS1
CS2
DON1
LNO1
Ö
Vd
DON2
Ed
CS3
DCD2
TheLoad
LNO2
Df 3
GTO3
(ON)
C2
CS4
DS2
GTO4
(OFF)
Df 4
RON
DON2
CS3
CS4
DS2
RS2
RS2
( a)
(b)
RS1
Ed
Df 1
GTO1
(OFF)
Df 2
GTO2
(ON)
C1
DCD1
Rd
DS1
CS1
CS2
DON1
LNO1
Ö
Vd
Ed
TheLoad
DCD2
LNO2
Df 3
GTO3
(OFF)
Df 4
GTO4
(OFF)
C2
RON
DON2
CS3
CS4
DS2
RS2
( c)
Fig.8-5. Commutation path of the transition form S1 to S0: (a) initial state, (b) phase 1 and (c)
phase-2.
Commutation from S0 to S−1:
During this phase GTO2 is turned off and GTO4 is turned on after the corresponding dead time.
The initial conditions are the following: CS1 and CS4 are completely charged up to Ed; CS2 and CS3
are completely discharged and the load current flows through the free wheeling diodes Df3 and Df4
as presented in Fig.8-6(a).
95
Chapter 8
Simulation results of three-level VSC snubber circuits
Phase 1: When GTO2 is turned off the load current will be absorbed by CS2 charging it. During
GTO2 is turned off, the voltage drop across GTO1 remains Ed, while the voltage drop across GTO2 is
increasing. iGTO2 decreases linearly from ILOAD to zero during the current fall time. This step ends
when iGTO2 reached zero (see Fig.8-6(b))[28], [31], [36], and [50].
Phase 2: The voltage over GTO2 continues to rise in this which ends when VC reaches Ed. The
S2
energy which is stored in LON1 will be dissipated through the loop DON2, RON, DON1, LON1 (see Fig.86(c)) [28], [31], [36], and [50].
Note: During the discharging process of the capacitor CS2, the discharging current will flow through
the following elements: GTO2, LON1, LON2, DON2, RON, CS2 (see Fig.8-6c).
Df 1
Ed
C1
DCD1
GTO1
(OFF)
GTO2
(OFF)
Df 2
Rd
RS1
RS1
DS1
CS1
Df 1
GTO1 D
(OFF) CS1
S1
Df 2
GTO2
(OFF)
Ed
C1
DCD1
CS2
Rd
DON1
LNO1
LNO1
Vd
Ed
DCD2
TheLoad
LNO2
Df 3
GTO3
(OFF)
RON
Vd
DON2
Ed
CS3
TheLoad
LNO2
Df 4
(a)
RON
DON2
DCD2
Df 3
GTO3
(OFF)
Df 4
GTO4
(OFF)
C2
C2
CS2
DON1
GTO4 CS4
(OFF) D
S2
CS3
CS4
DS2
RS2
RS2
(b)
RS1
Ed
Df 1
GTO1 DS1
(OFF) CS1
Df 2
GTO2
(OFF) CS2
DON1
C1
DCD1
Rd
Ö
CS2
discharging
-path
LNO1
Vd
Ed
TheLoad
DCD2
Df 3
C2
Df 4
LNO2
GTO3
(ON)
GTO4
(ON)
RON
DON2
CS3
CS4
DS2
RS2
(c)
Fig.8-6 Commutation path of the transition form S0 to S-1: (a) initial state, (b) phase 1 and (c)
phase-2.
96
Chapter 8
Simulation results of three-level VSC snubber circuits
8.4.1 Performance of the optimized snubber design
The main advantage of the optimized snubber design is that the three snubber circuits (turn-on,
turn-off, and over-voltage) are used in combination and that the number of the required elements is
less than those which are necessary for the common snubber circuit. The operation of the power
semiconductor switching devices still remains in the SOA. Additionally the converter size, its cost,
and the losses can be reduced. Fig.8-7 shows a three level inverter with the corresponding
optimized snubber design.
• The turn-off snubber circuit protects the power switching semiconductor device from the
over-voltage, which appears during the turn-off process, because of the high current
change rate di/dt. Therefore, the circuit wiring inductance can cause a high turn-off surge
voltage. The charging process takes place as in the common snubber circuit, through the
turn-off diode. The discharging process occurs through the power semiconductor
switching device, the turn-on resistor, and the mid-point inductance for the second and
third GTO. The charging and discharging of GTO1 and GTO4 is realized in the ‘normal’
way by DS1/DS2 and RS1/RS2, respectively.
• The turn-on snubber’s circuits limits the di/dt during turning on and reducing the voltage
across the power semiconductor switching device due to the voltage drop across the
inductance when the current builds up. During the on-state of the power semiconductor
switching device, the load current flows through the turn-on inductance. When the power
semiconductor switching device is turned off, the stored energy in the snubber inductance
will be dissipated in the resistor RON. When GTO1 and GTO2 are switched on and off, the
inductor LON2 will decay the current to low value. So the snubber circuit will be effective
in the next turn-on, and vice versa, LON1 will do the same during GTO3 and GTO4
switching.
• The over-voltage snubber circuit minimizes the over-voltage which is caused by the strayand turn-on inductance across the power semiconductor switching device to safe levels.
The energy stored in LON1 gets transferred to the turn-off capacitor CS2 through the diode
DON1. So, there is no over-voltage across the power semiconductor switching device
because it will be limited to Ed through the resistor RON. This snubber circuits consist of
the following elements: CSµ, RON, DONµ, and DSµ [50], [38], [39], and [40].
8.5 Dual-use snubber circuit for three-level inverter
The suggested snubber circuit “the dual-use snubber circuit,” is the modified design of the
optimized snubber design. The modification in is in the position of the turn-off snubber resistor RSµ.
The turn-off resistor RSµ will be replaced to have a new position as presented in Fig.8-8. The new
location of the resistor RSµ will give the snubber circuit design the same advantages of the optimized
snubber circuit and an extra advantage. While the resistor RSµ in this case is much more useful
during the turn-off processes and has two functions [8], [28], [31], [36], and [50]:
• The first function is during the discharge process of CS1. The capacitor energy will be
dissipated in it and the discharge current will be limited.
• The second function is during the charging process of CS2, The resistor has a damping effect
of the over-voltage across the power switching device while charging process of CS2.
Therefore, the over-voltage will be reduced, especially in the beginning of turn-off process
[55], [56], and [57].
97
Chapter 8
Simulation results of three-level VSC snubber circuits
Ed
Df 1
Df 2
GTO2
Vd
CS2
DON1
Rd
The Load
CS1
RS1
C1
DCD1
DS1
GTO1
LNO1
RON
LNO2
DON2
Ed
DCD2
Df 3
GTO3
Df 4
GTO4 DS1
RS1
CS3
C2
CS4
Fig.8-7 One phase of a three level inverter with the new dual-use snubber circuit design.
8.6 Dual-inductive snubber circuit for three-level Inverter
This is the developed module of the dual-use snubber circuit to update the dual-use snubber circuit
design in order to get more new features and optimized performance of the snubber circuit. It
comprises the three snubber circuit (turn-on, turn-off and over-voltage) and has a parallel inductor
with the turn-on resistor (see Fig.8-8). The performance of the dual-inductive snubber circuit is
better than the performance of common snubber circuits, especially the limitation of the current,
suppressing the over-voltage and minimization of the losses. The dual-inductive snubber circuit has
additionally more advantages than the dual-use snubber design which are mentioned in the
following:
• The dual-inductive snubber design suppresses the over-voltage protection of the power
semiconductors much more than that of conventional snubber circuits, because of the new
arrangement of the snubber circuit elements.
• The parallel inductor will decay the current during the turn-on process to a lower value. So
the turn-on snubber circuit is more effective.
• The resulting voltage over RON will drop nearly to zero when the turn-on process is over
because of the parallel inductor LONP. Therefore the losses in RON during the on-states drop
fundamentally when the transient processes are decayed.
• The parallel inductor can be a primary winding of transformer which can be effective as a
power recovery system.
98
Chapter 8
Simulation results of three-level VSC snubber circuits
• The theoretical reliability is higher because of the fewer elements in the snubber circuit
[36], [50], and [57].
Ed
GTO1
Df 2
GTO2
CS1
RS1
C1
DCD1
DS1
Df 1
CS2
DON1
Rd
LNO1
RON
Vd
LONP
The Load L
NO2
DON2
Ed
DCD2
Df 3
GTO3
Df 4
GTO4 DS1
RS1
CS3
C2
CS4
Fig.8-8 One phase of a three-level inverter with the new dual-inductive snubber circuit.
8.7 Simulation and discussion of the results
The simulation will be based on a detailed Matlab(R)/SimulinkTM model. The analyzed model
comprises a single phase of a MV three level inverter. The load and source data are presented in the
following Table 8.4. The internal parameters of the GTO and its common snubber circuit (which are
RS, CS) were taken from the data sheet of the used GTO semiconductor devices. Table 8.4 shows the
parameters of the used GTO and the load information.
GTO
parameters
Forwardvoltage
Internal
resistor
Internal
reactance
Vf (V)
RON (Ω)
LON (µΗ)
1
0.001
1
Load and source data
Current fall
time
tfi (µs)
5
Snubber
resistor
Snubber
capacitor
RS (Ω)
CS (µF)
5
1
Load voltage Ed (V)
Load current IL (A)
Power factor Pf
6000
4600
0.8
Table 8.4 the parameters of the used GTO and the load information.
99
Chapter 8
Simulation results of three-level VSC snubber circuits
8.7.1 Description of the PWM in Matlab®/SimulinkTM
Power semiconductor devices are used as mentioned before to control the voltage and the
current of converter systems, because it has lower energy losses compared with other continuous
controller like resistor. The pulsed control system will be used to drive the power semiconductor
devices for achieving high efficiency The pulse width modulation (PWM) is the most widely used
type of control system for the power semiconductor devices when it needs to control the output of
the converter systems. For achieving smooth control of the voltage or current the switching
frequency (commutation frequency) should be relative high (kHz).
The circuitry of the PWM will be built in Matlab®/SimulinkTM tools. The circuitry consists of
the block components, which can give the needed PWM (e.g. 2-3kHz), those components contain
the sine wave (50Hz) which is the voltage reference, the pulse generator which give pulse scheme
which will be generate the saw wave (tri-angle pulse scheme) at the required switching frequency
(see Fig.8-9), and the comparison box which contains the comparison elements like great- or lessthan, and logic elements And/Or, the main aim of this box is to compare between the sine wave and
the saw wave, as presented in Fig.8-9 and Fig.8-10. The output of the comparison will be true (1) if
the sine wave is bigger than the saw wave, this means that the IGBT will switch on during this time,
or false (0) when the saw wave is bigger than the sine wave, as shown in Fig.8-11. Fig.8-12 shows
the pulse scheme of the Matlab®/SimulinkTM simulation which will drive the IGBT at the
frequency of e.g.3 kHz. This pulse scheme will be used as reference in other programming tools
(will be discussed in Chapter9). With the programming tools, the pulse scheme, which will drive the
gates of IGBT converter, is the output of the microcontroller chip which will be the driver system.
Fig.8-9 The main flow chart of the PWM in Matlab®/SimulinkTM.
100
Chapter 8
Simulation results of three-level VSC snubber circuits
Sine, Saw [p.u]
Fig.8-10 The internal part of the PWM in Matlab®/SimulinkTM.
1,0
0,5
Sine wave
Saw wave 1
Saw wave 2
0,0
-0,5
-1,0
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-11 The comparison functions of the PWM in Matlab®/SimulinkTM.
101
Chapter 8
Simulation results of three-level VSC snubber circuits
1,0
IGTO1 0,8
0,6
0,4
0,2
0,0
0,000
1,0
0,8
IGTO2 0,6
0,4
0,2
0,0
0,000
1,0
IGTO3 0,8
0,6
0,4
0,2
0,0
0,000
0,005
0,010
0,015
0,020
t[s] 0,025
0,030
0,005
0,010
0,015
0,020 t[s] 0,025
0,030
0,005
0,010
0,015
0,020
t[s] 0,025
0,030
0,005
0,010
0,015
0,020 t[s] 0,025
0,030
1,0
0,8
IGTO4 0,6
0,4
0,2
0,0
0,000
Fig.8-12 the output of the PWM in Matlab®/SimulinkTM.
8.7.2 Comparison of the proposed double snubber configuration and the common
snubber circuit
In this simulation, the proposed double snubber configuration will be compared with the
common snubber circuit. The parameters of the common snubber circuit are constant (standard
values). The values of the proposed double snubber configuration were chosen exactly in the same
way, but the value of CS and RS were reduced to other values as in Table 8.5. While this reduction
improves the performance of the double snubber circuit, the results for the common snubber
configuration remain nearly the same. Additionally the values of the parallel parameter will be set
to different values to indicate to its effect. Table 8.5 gives the values of snubber circuits in all
simulation cases.
Snubber circuit data
Double snubber configuration
Common snubber
CS (µF)
1
0.5
1
RS (Ω)
5
2.5
5
CP (µF)
0.05
0.01
-
RP (Ω)
5
1
-
Table 8.5 Snubber circuits (common and double configuration) information.
The simulation’s results for the first value ( CP = 0.05µ F ) are presented in Figs 8-13 to 8-18.
Figs 8-13 and 8-14 show the current and the voltage on GTO1 and GTO2. The overvoltage is limited
in a better way by the double snubber configuration and the current peaks are much smaller as
shown in Fig.8-14.
102
Simulation results of three-level VSC snubber circuits
uGTO1 [V]
Chapter 8
7000
uGTO1-C
uGTO1-D
6000
5000
0,000
0,005
0,010
0,015
t [s]
0,020
uGTO2 [V]
uGTO2-C
uGTO2-D
6000
0,0050
0,0075
0,0100
0,0125
t [s]
0,0150
Fig.8-13 The voltages on GTO1 and GTO2 in the common and the double snubber
configuration (CP=0.05µF).
6000
iGTO1 [A]
5000
iGTO1-C
4000
iGTO1-D
3000
2000
1000
0
0,000
0,005
0,010
0,015
t [s]
0,020
0,015
t [s]
0,020
iGTO2 [A]
6000
5000
iGTO2-C
4000
iGTO2-D
3000
2000
1000
0
0,000
0,005
0,010
Fig.8-14 The currents in GTO1 and GTO2 in the common and the double snubber
configuration (CP=0.05µF).
The Figs 8-15, 8-16 and 8-17 show the current and the voltage in the freewheel diode and in the
turn-off diode. It can easily be seen that the current and the voltages resulting from the new
proposed protection circuits are more favorable than those simulated for the conventional circuit. In
Fig.8-18 the total losses in the conventional circuit in the proposed double snubber design are
compared based on energy function over two cycles of the fundamental frequency. As indicated, the
energy lost in the entire new snubber circuit is much lower than in the standard snubber design.
Also, the losses in the GTO itself are less for the new design but do not differ too much.
103
Chapter 8
Simulation results of three-level VSC snubber circuits
iDs1 [A]
5000
4000
iDs1-C
3000
iDs1-D
2000
iDs2 [A]
1000
0
0,000
3000
0,005
0,010
t [s]
0,015
0,020
2500
2000
iDs2-C
1500
iDs2-D
1000
500
0
0,000
0,005
0,010
t [s]
0,015
0,020
Fig.8-15 The currents in Df1 and Df2 in the common and the double snubber Configuration
(CP=0.05µF).
0
uDs1 [V]
-1500
uDs1-C
uDs1-D
-3000
-4500
0,000
0
0,001
0,002
0,003
0,004
t [s]
0,005
uDs2 [V]
-1500
uDs2-C
-3000
uDs2-D
-4500
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-16 The voltages on DS1 and DS2in the common and the double snubber configuration
(CP=0.05µF).
104
Chapter 8
Simulation results of three-level VSC snubber circuits
iDf1 [A]
5000
4000
3000
iDf1-C
2000
iDf1-D
1000
iDf2 [A]
0
0,000
0,005
3000
0,010
0,015
t [s]
0,020
0,010
0,015
t [s]
0,020
iDf2-C
iDf2-D
2000
1000
0
0,000
0,005
Fig.8-17 The currents in DS1 and DS2 in the common and the double snubber configuration
(CP=0.05µF).
140000
Losses [Ws/p.u.]
120000
100000
80000
60000
con- Losses
Double-Losses
40000
20000
0
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-18 The total losses (energy function) in RS1 in the conventional- and in RS1, RP1 in the
proposed double snubber design over one cycle of the fundamental frequency (CP=0.05µF).
In the following figures, the effect of varying the CP to a smaller value is demonstrated
(CP=0.01µF). Despite the additional reduction of the over-voltage and the current in the switching
device GTO1 and GTO2 (Figs.8-19, 8-20), the effect on the over-voltage across GTO1 is small in
comparison with the same parameter on GTO2 .But the current’s spikes are much smaller in the new
design compared with the currents of GTO1, 2 in the common snubber circuit switching devices. The
effect of changing the CP is much better visible in the time functions for the turn-off diode (Figs 8105
Chapter 8
Simulation results of three-level VSC snubber circuits
uGTO1 [V]
21, 8-22, and 8-23). This tendency also holds for the freewheel diode. Also, the total losses were
fewer for the double snubber configuration than those in common snubber circuit (see Fig.8-24).
6500
6000
uGTO1-C
5500
uGTO2 [V]
uGTO1-D
5000
0,000
6500
0,005
0,010
0,015
t [s]
0,020
uGTO2-C
uGTO2-D
6250
6000
0,0050
0,0075
0,0100
0,0125
t [s]
0,0150
iGTO1 [A]
Fig.8-19 The voltages in GTO1 and GTO2 in the common and the double snubber
configuration (CP=0.01µF).
6000
4500
iGTO1-C
iGTO1-D
3000
iGTO2 [A]
1500
0
0,000
0,005
0,010
0,015
t [s]
0,020
0,015
t [s]
0,020
6000
iGTO2-C
4500
iGTO2-D
3000
1500
0
0,000
0,005
0,010
Fig.8-20 The currents in GTO1 and GTO2 in the common and the double snubber configuration
(CP=0.01µF).
106
Chapter 8
Simulation results of three-level VSC snubber circuits
iDs1 [A]
5000
4000
iDs1-C
3000
iDs1-D
2000
iDs2 [A]
1000
0
0,000
3000
0,005
0,010
t [s]
0,015
0,020
2500
2000
iDs2-C
1500
iDs2-D
1000
500
0
0,000
0,005
0,010
t [s]
0,015
0,020
Fig.8-21 The currents in DS1 and DS2 in the common and the double snubber configuration
(CP=0.01µF).
0
-1000
uDs1-C
-2000
uDs1-D
uDs1 [V]
-3000
-4000
-5000
0,000
0
0,001
0,002
0,003
0,004
t [s]
0,005
-1000
uDs2 [V]
-2000
-3000
uDs2-C
-4000
uDs2-D
-5000
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-22 The voltages on DS1 and DS2 in the common and the double snubber configuration
(CP=0.01µF).
107
Simulation results of three-level VSC snubber circuits
iDf1 [A]
Chapter 8
4000
3000
iDf1-C
iDf1-D
2000
1000
iDf2 [A]
0
0,000
0,005
3000
0,010
0,015
t [s]
0,020
0,010
0,015
t [s]
0,020
iDf2-C
iDf2-D
2000
1000
0
0,000
0,005
Fig.8-23 The currents in Df1and Df2 in the common and the double snubber Configuration
(CP=0.01µF).
Losses [Ws/p.u.]
125000
100000
75000
50000
Losses-C
Losses-D
25000
0
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-24 The total losses (energy function) in RS1 in the conventional and in RS1, RP1 in the
proposed double snubber design over two cycles of the fundamental frequency (CP=0.01µF).
8.7.2.1 Conclusion remarks
The simulation results show that the proposed double snubber configuration facilitates reducing
the stress in the switching device (e.g. GTO’s) as well as the other semiconductor devices in the
form of the turn-off and the freewheel diodes. The dependency on the capacitance of the additional
snubber parts is visible but not extremely strong. Therefore, rather small, cheap, and economical
elements should be able to provide this additional advantageous effect. The new double snubber
circuit is suitable for high voltage and high power more level converter systems and FACTS
108
Chapter 8
Simulation results of three-level VSC snubber circuits
systems. The advantages of the new design is the ability to minimize both, the over voltage and the
losses in the entire circuit by using an additional parallel RC damping snubber. Therefore, the
additional RC circuit can compensate the disadvantage of the conventional RCD snubber design,
that the effective value for RS during the charging of snubber capacity CS is essentially zero. But the
main cost of the snubber circuit will be high compared with the common snubber circuit design
while the number of passive elements is large.
8.7.3 Comparing the common and the optimized snubber design
In this simulation, a comparison between the common snubber circuit and the optimized
snubber design was done. The element parameters were chosen in the same way, the only different
is that the value of capacitor CS was reduced to other values (see Table 8.6) in the simulation, while
this reduction improves the performance of the optimized snubber design in the direction of reduced
losses and the over-voltage across the power switching device. The load and source data will be the
same as in Table 8.4
Snubber circuit data
Optimized snubber design
Common snubber
CS (µF)
1
0.5
0.25
1
RS (Ω)
5
5
2
5
Table 8.6 the snubber circuits (common and design) information.
uGTO1 [V]
In the following Figures, the stresses in different semiconductor devices (GTOs and diodes) will
be analyzed for CS =1µF. Figs 8-25 and 8-26 show that the over-voltages across and the currents in
GTO1 and GTO2. The over-voltages across GTO1, 2 are limited in a better way by the optimized
snubber than the common snubber design during the turn-off process, and the current spikes are also
much smaller in GTO2 and about the same in GTO1 than those in the common snubber circuit as
shown in Fig.8-25.
6500
uGTO1-C
6250
uGTO1-Opti
6000
0,005
0,010
0,015
uGTO2 [V]
5750
0,000
6500
t [s]
0,020
uGTO2-C
uGTO2-Opti
6250
6000
0,0050
0,0075
0,0100
0,0125
t [s]
0,0150
Fig.8-25 The voltages in GTO1 and GTO2 in the common and the optimized snubber design.
109
Chapter 8
Simulation results of three-level VSC snubber circuits
iGTO1 [A]
Figs.8-27, 8-28 and 8-29 show the voltages and the currents in diodes DS1 and Don1 and the total
losses as an energy function. It can easily be seen, that the currents resulting from the optimized
snubber design are more adequate than those simulated for the common snubber circuit, especially
in the turn-off diode DS1 and turn-on diode Don1. The current in freewheel diode Df1 is about the
same but in Df2 the current in the optimized snubber circuit design is clearly smaller the common
snubber circuit (see Fig.8-24(a)).
7500
iGTO1-C
5000
iGTO1-Opti
iGTO2 [A]
2500
0
0,000
0,005
0,010
0,015
t [s]
0,020
6000
4500
iGTO2-C
3000
iGTO2-Opti
1500
0
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-26 The currents in GTO1 and GTO2in the common and the optimized snubber design.
iDf1 [A]
6000
4000
iDf1-C
2000
iDf1-Opti
iDf2 [A]
0
0,000
0,005
3000
iDf2-C
2000
iDf2-Opti
0,010
0,015
0,010
0,015
t [s]
0,020
1000
Losses [Ws/p.u.]
0
0,000
0,005
t [s]
0,020
(a)
150000
100000
Losses-C
Losses-Opti
50000
0
0,000
0,005
0,010
0,015
t [s]
0,020
(b)
Fig.8-27 (a) The currents in Df1 and Df2 and comparing the total losses in conventional and (b)
the optimised snubber configuration (energy function).
110
Simulation results of three-level VSC snubber circuits
iDs1 [A]
Chapter 8
4500
iDs1-C
3000
iDs1-Opti
iDon1 [A]
1500
0
0,000
4500
0,005
0,010
0,015
t [s]
0,020
0,015
t [s]
0,020
iDon1-C
3000
iDon1-Opti
1500
0
0,000
0,005
0,010
Fig.8-28 The current in DS1 and Don1 in the common and the optimized snubber configuration.
The over-voltages across Don1 and DS1 are the same as shown in the two snubber circuit as
presented in Fig 8-29. The total losses in the circuit (GTO, diode and snubber element losses) for
the two different snubber designs (conventional snubber circuit and optimized snubber design) are
illustrated in Fig.8-27(b). The losses are shown as an energy function over one cycle of the
fundamental frequency. As indicated, the energy lost in converters based on the optimized snubber
design is much lower than that dissipated in the common snubber design.
0
uDs1 [V]
-1500
uDs1-C
-3000
uDs1-Opti
-4500
0,000
0
0,005
0,010
0,015
t [s]
0,020
0,015
t [s]
0,020
uDon1-C
-1500
uDon1-Opti
uDon1 [V]
-3000
-4500
-6000
0,000
0,005
0,010
Fig.8-29 The voltages in DS1 and Don1 in the common and the optimized snubber
configuration.
111
Chapter 8
Simulation results of three-level VSC snubber circuits
uGTO1 [V]
In the next two simulations, CS was set to 0.5µF and then 0.25µF. Figs 8-30 to 8-34 present the
simulation results for the value of CS. Figs 8-30 and 8-31 show the voltage and the current resulting
in GTO1,2. The over-voltage shown in Fig.8-30 across GTO1 and GTO2 in the optimized snubber
design is more limited than it is in the common snubber design and the current spikes through GTO2
are much smaller as presented in Fig.8-31.
uGTO1-C
6500
uGTO1-Opti
6000
uGTO2 [V]
5500
5000
0,000
6500
0,005
0,010
0,015
t [s]
0,020
uGTO2-C
uGTO2-Opti
6250
6000
0,0050
0,0075
0,0100
0,0125
t [s]
0,0150
iGTO1 [A]
Fig.8-30 The voltages on GTO1 and GTO2 in the common and the optimized snubber design
(CS=0.5µF, RS=5Ω).
8000
6000
iGTO1-C
iGTO1-Opti
4000
2000
iGTO2 [A]
0
0,000
0,005
0,010
0,015
t [s]
0,020
6000
iGTO2-C
4000
iGTO2-Opti
2000
0
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-31 The currents in GTO1 and GTO2 in the common and the optimized snubber design
(CS=0.5µF, RS=5Ω).
The currents through freewheel-, turn-off and turn-on diodes are better improved in the
optimized snubber design than in the common snubber circuit. Fig.8-32 shows currents through DS1
and Don1, and Fig.8-34(a) illustrates the current of Df1,2. The over-voltage across the DS1 and Don1 in
112
Chapter 8
Simulation results of three-level VSC snubber circuits
iDs1 [A]
the optimized snubber design becomes more acceptable than the same voltage in the common
snubber circuit comparatively with the result of the last simulation as clarified in Fig.8-33.The total
losses in the optimized snubber design and the common snubber circuit are compared as an energy
function over one cycle of the fundamental frequency in Fig.8-34(b).
4500
3000
iDs1-C
iDs1-Opti
1500
iDon1 [A]
0
0,000
0,005
0,010
0,015
t [s]
0,020
0,015
t [s]
0,020
4500
3000
iDon1-C
iDon1-Opti
1500
0
0,000
0,005
0,010
Fig.8-32 Currents in DS1 and Don1 in the common and the optimized snubber configuration
(CS=0.5µF, RS=5Ω).
0
uDs1 [V]
-1500
uDs1-C
-3000
uDs1-Opti
-4500
0,000
0
0,005
0,010
0,015
t [s]
0,020
0,015
t [s]
0,020
-1500
uDon1-C
uDon1 [V]
-3000
uDon1-Opti
-4500
-6000
0,000
0,005
0,010
Fig.8-33 Voltages on DS1 and Don1 in the common and the optimized snubber configuration
(CS=0.5µF, RS=5Ω).
113
Chapter 8
Simulation results of three-level VSC snubber circuits
iDf1 [A]
6000
4000
iDf1-C
2000
0
0,000
iDf1-Opti
0,005
0,015
t [s]
0,020
(a)
iDf2-C
3000
iDf2 [A]
0,010
iDf2-Opti
2000
Losses [Ws/p.u.]
1000
0
0,000
150000
0,005
0,010
0,015
t [s]
0,020
t [s]
0,020
100000
50000
0
0,000
Losses-C
Losses-Opti
0,005
0,010
0,015
(b)
Fig.8-34 (a) The currents in Df2 and Df2 in the common and the optimized snubber
configuration, and (b) the comparison of the total losses in the conventional and the optimized
snubber configuration (energy function) (CS=0.5µF, RS=5Ω).
As mentioned before, the total energy lost in the optimized snubber design in this simulation is
much lower than in the common snubber circuit. One of the reasons therefore is that the losses are
related to the capacitor CS. Also, the losses in the GTO’s itself are less for the optimized snubber
design but the difference here is not so much. For the other value of (CS =0.25µF), the performance
of the optimizer snubber will be more typical, especially in the over-voltage protection and
minimization of the total losses. Fig.8-35 and Fig.8-36 show the effect of CS reduction, the overvoltage becomes smaller and also the spikes in the current for GTO1,2.
The current of freewheel diodes and total losses energy function are presented in Fig.8-37. The
current in Df1 in the optimizer snubber design has about the same value, but in Df2 the current is
smaller in comparison of the common snubber circuit, while the total losses are clearly smaller than
these of the common snubber circuit.
Based on the aforementioned results, the optimized snubber design is based on a new and
simple structured snubber, which depends on passive elements. As a result, the number of required
components is reduced, and the reliability increases distinctly. The presented optimized snubber
design provides several additional advantages: It increases the optional performance of the three
level converters due to the lower clamping over-voltages across the switching devices, it improves
the efficiency because of the lower snubber and total losses, its suitable structure can be extended to
114
Chapter 8
Simulation results of three-level VSC snubber circuits
uGTO1 [V]
energy recovery snubbers and there is no unbalance problem. Additionally, the manufacturing costs,
the complexity and therefore the converter size can be reduced.
6500
uGTO1-C
6250
uGTO1-Opti
uGTO2 [V]
6000
5750
0,000
6500
0,005
0,010
0,015
t [s]
0,020
uGTO2-C
uGTO2-Opti
6250
6000
0,0050
0,0075
0,0100
0,0125
t [s]
0,0150
iGTO1 [A]
Fig.8-35 The voltages in GTO1 and GTO2 in the common and the optimized snubber design
(CS=0.25µF, RS=2Ω).
7500
6000
iGTO1-C
4500
iGTO1-Opti
3000
iGTO2 [A]
1500
0
0,000
0,005
0,010
0,015
t [s]
0,020
6000
iGTO2-C
4500
iGTO2-Opti
3000
1500
0
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-36 The currents in GTO1 and GTO2 in the common and the optimized snubber design
(CS=0.25µF, RS=2Ω).
115
Chapter 8
Simulation results of three-level VSC snubber circuits
iDf1 [A]
4500
3000
iDf1-C
1500
iDf1-Opti
iDf2 [A]
0
0,000
0,005
0,010
0,015
t [s]
0,020
0,010
0,015
t [s]
0,020
0,010
0,015
t [s]
0,020
iDf2-C
3000
iDf2-Opti
1500
0
0,000
0,005
(a)
Losses [ws/p.u.]
150000
100000
Losses-C
Losses-Opti
50000
0
0,000
0,005
(b)
Fig.8-37 (a) The currents in Df2 and Df2 in the common and the optimized snubber
configuration, and (b) the comparison of the total losses in the conventional and the optimized
snubber configuration (energy function) (CS=0.25µF, RS=2Ω).
8.7.4 Comparison of the common and the dual-use snubber circuits for different values
for CS-D
In the first simulation, a comparison between the common snubber circuit and the dual-use
snubber design will be done. Two different scenarios were analyzed as follow: In the first, the
parameters were defined by maximum values for modern power semiconductor device for an MV
inverter system. The turn-off snubber parameters were set to the standard parameter values and
other values (see Table 8.4 and Table 8.7) while the load current will be IL= 5000A to get the
optimal values for the snubber circuit elements.
Snubber circuit data
Dual-use snubber design
Common snubber
CS (µF)
0.5
0.25
1
RS (Ω)
2.5
2.5
5
Table 8.7 Snubber circuits (common and double dual-use) information.
In the following figures the stresses for different power semiconductor devices (GTOs and
diodes) will be discussed and analyzed. Fig.8-38 and Fig.8-39 show that the voltages and the
current in GTO1, 2 in the two circuits, the over-voltage (shown in Fig.8-38) is limited in a better way
by the dual-use snubber circuit and the current pecks are smaller as presented in Fig.8-39.
116
Chapter 8
Simulation results of three-level VSC snubber circuits
uGTO1 [V]
Figs.8-40 and 8-41 present the current and the voltage in DS1 and Don1. It is explicitly seen that
the currents in DS1 and Don1 in the dual-use snubber circuit are smaller than those of the common
snubber circuit. The voltage in the diode in the dual-use snubber circuit is suppressed much more
than the common snubber circuit. For the free-wheel diodes, it can be see that the currents of
freewheel diodes resulting from the dual-use snubber circuit are more favorable than those of the
simulated for the common snubber circuit as shown in Fig.8-42(a).
6400
uGTO1-C
6200
uGTO1-Dual
6000
5800
uGTO2 [V]
0,000
0,005
0,010
0,015
t [s]
0,020
t [s]
0,0150
uGTO2-C
6400
uGTO2-Dual
6200
6000
0,0050
0,0075
0,0100
0,0125
Fig.8-38 The voltages on GTO1 and GTO2 in the common and the dual-use snubber
configuration (CS-D=0.5µF, RS=2.5Ω).
iGTO1 [A]
7500
6000
iGTO1-C
4500
iGTO1-Dual
3000
1500
iGTO2 [A]
0
0,000
6000
0,005
0,010
4500
0,015
t [s]
0,020
0,015
t [s]
0,020
iGTO2-C
iGTO2-Dual
3000
1500
0
0,000
0,005
0,010
Fig.8-39 The currents in GTO1 and GTO2 in the common and the dual-use snubber
configuration (CS-D=0.5µF, RS=2.5Ω).
117
Simulation results of three-level VSC snubber circuits
iDs1 [A]
Chapter 8
4500
iDs1-C
3000
iDs1-Dual
1500
iDon1 [A]
0
0,000
0,005
0,010
0,015
t [s]
0,020
0,015
t [s]
0,020
4500
iDon1-C
iDon1-Dual
3000
1500
0
0,000
0,005
0,010
Fig.8-40 The currents in DS1 and Don1 in the common and the dual-use snubber
configuration (CS-D=0.5µF, RS=2.5Ω).
In Fig.8-42(b), the total losses in the power semiconductor devices and in RSµ in the two
inverters are compared as energy function over one cycle of the fundamental frequency.
0
uDs1 [V]
-1500
-3000
uDs1-C
uDs1-Dual
-4500
0,000
0
0,005
0,010
t [s]
0,020
0,015
t [s]
0,020
uDon1-C
-1500
uDon1-Dual
-3000
uDon1 [V ]
0,015
-4500
-6000
0,000
0,005
0,010
Fig.8-41 The voltages in DS1 and Don1 in the common and the dual-use snubber configuration
(CS-D=0.5µF, RS=2.5Ω).
118
Chapter 8
Simulation results of three-level VSC snubber circuits
iDf1 [A]
4500
3000
iDf1-C
1500
iDf1-Dual
iDf2 [A]
0
0,000
0,005
4500
iDf2-C
3000
iDf2-Dual
0,010
0,015
t [s]
0,020
0,010
0,015
t [s]
0,020
1500
Losses [Ws/p.u.]
0
0,000
0,005
(a)
150000
100000
Losses-C
Losses-Dual
50000
0
0,000
0,005
0,010
0,015
t [s]
0,020
(b)
Fig.8-42 (a) The currents in Df2 and Df2 in the common and dual-use snubber configuration,
and (b) comparison of the total losses in the conventional and the optimized snubber
configuration (energy function) (CS-D=0.5µF, RS=2.5Ω).
As indicated, the energy lost in the entire power semiconductor device and snubber circuit
resistors in the dual-use snubber circuit is much lower than that in the common snubber circuit.
Also, the losses in the GTO’s itself are less for the dual-use snubber circuit but do not differ too
much.
This next simulation was carried out for a small value of CS-D = 0.25µF and RS-D = 2.5 Ω. The
simulation results are analyzed as the following: Figs 8-43 and 8-44 illustrate the voltages and the
current in GTO1 and GTO2. The over-voltage across the GTOµ is limited much better, and the peak
of current is smaller than the first simulation. For the turn-off diode, the over-voltages across DS1,
Don1 are also less than in the first simulation; the currents have the same advantages (see Fig.8.-45
and Fig.8-46). The currents in the freewheel diodes Df1 and Df2 are more adequate as shown in
Fig.8-47(a), in which the current in Df2 is smaller than the same current in the other simulation. The
total losses are illustrated in Fig.8-47(b) as energy function: The total energy lost in the inverters
based on the dual-use snubber circuit is lower than that dissipated in the common snubber circuit.
119
Simulation results of three-level VSC snubber circuits
uGTO1 [V]
Chapter 8
6500
uGTO1-C
6250
uGTO1-Dual
uGTO2 [V]
6000
5750
0,000
6500
0,005
0,010
0,015
t [s]
0,020
uGTO2-C
uGTO2-Dual
6250
6000
0,0050
0,0075
0,0100
0,0125
t [s]
0,0150
Fig.8-43 The voltages in GTO1 and GTO2 in the common and the dual-use snubber
configuration (CS-D=0.25µF, RS=2.5Ω).
iGTO1 [A]
6000
4500
3000
iGTO1-C
iGTO1-Dual
1500
0
iGTO2 [A]
0,000
6000
0,005
0,010
0,015
t [s]
0,020
iGTO2-C
4500
iGTO2-Dual
3000
1500
0
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-44 The currents in GTO1 and GTO2 in the common and the dual-use snubber
configuration (CS-D=0.25µF, RS-D =2.5Ω).
120
Simulation results of three-level VSC snubber circuits
iDs1 [A]
Chapter 8
4500
3000
iDs1-C
iDs1-Dual
1500
iDon1 [A]
0
0,000
0,005
0,010
0,015
t [s]
0,020
0,015
t [s]
0,020
4500
3000
iDon1-C
iDon1-Dual
1500
0
0,000
0,005
0,010
Fig.8-45 The voltages in DS1 and Don1 in the common and the dual-use snubber
configuration (CS-D=0.25µF, RS-D =2.5Ω).
0
uDs1 [V]
-1500
uDs1-C
-3000
uDs1-Dual
-4500
0,000
0
0,005
0,015
t [s]
0,020
uDon1-C
-1500
uDon1-Dual
-3000
uDon1 [V]
0,010
-4500
-6000
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-46 The voltages in DS1 and Don1 in the common and the dual-use snubber configuration
(CS-D=0.25µF, RS-D=2.5Ω).
Based on the aforementioned results, an optimized snubber design is proposed to be used in
high voltage and high power more-level converter systems and partially in FACTS devices. The
121
Chapter 8
Simulation results of three-level VSC snubber circuits
iDf2 [A]
iDf1 [A]
main advantage of the new design is the simple structure snubber circuit, which realizes the facility
to minimize the over-voltage, the total losses in the entire circuit, no unbalance problems and the
manufacturing costs of the snubber circuit. Using a Matlab®/ SimulinkTM simulation based on an
MV three-level inverter system, the over-voltages and the losses are compared between the
common and the dual-use snubber circuits. The results clarify the features of the dual-use snubber
circuit especially for the operation close to the limits of the SOA (Safe Operation Area).
4500
3000
iDf1-C
1500
iDf1-Dual
0
0,000
4500
0,005
0,010
0,015
0,010
0,015
t [s]
0,020
iDf2-C
3000
iDf2-Dual
1500
Losses [Ws/p.u.]
0
0,000
150000
0,005
t [s]
0,020
(a)
Losses-C
Losses-Dual
100000
50000
0
0,000
0,005
0,010
0,015
t [s]
0,020
(b)
Fig.8-47 (a) The currents in Df2 and Df2 in the common and dual-use snubber configuration
and (b) comparison between the total losses in the conventional and the optimized snubber
configuration (energy function) (CS-D=0.25µF, RS-D=2.5Ω).
8.7.5 Comparison of the dual-inductive- and common snubber circuit
The simulation will be performed for the standard parameters and another selected optimized
values for the dual-inductive snubber circuit and the common snubber circuit design. The values of
the parameters were set to the standard values for the common snubber circuit and the dualinductive snubber for the first simulation. In the second simulation the parameters of the dualinductive snubber were modified to the next values CS=0.25µF and RS=2.5Ω, but the load
characteristics were the same as in the last comparison. Figs 8-48 and 8-49 show the voltages and
the current in GTO1 and GTO2. The over-voltages across GTOs in the dual-inductive snubber circuit
are suppressed to save values in comparison with the common snubber circuit. Because the reason
of the over-voltage “RON and stray inductance” have very small effect as discussed before. While,
the parallel inductive limits the di/dt, makes the effect RON smaller, and absorbs the total energy in
the turn-on snubber circuit, the current spikes are also restricted from those of the common snubber
122
Chapter 8
Simulation results of three-level VSC snubber circuits
uGTO1 [V]
circuit. The voltage across DS1 is less than the common snubber circuit, but the over-voltage across
the turn-on diode Don1 is a bit bigger than the same diode in the common snubber circuit, because
the Don1-D-Ind has now two functions: turn-on and turn-off diode (see Fig.8-50).
6400
uGTO1-C
6200
uGTO1-D-Ind
6000
5800
0,000
0,005
0,010
0,015
t [s]
0,020
0,0125
t [s]
0,0150
uGTO2-C
uGTO2-D-Ind
uGTO2 [V]
6400
6200
6000
0,0050
0,0075
0,0100
Fig.8-48 The voltages on GTO1 and GTO2 in the common and the Dual-indicative Snubber
design (Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω).
iGTO1 [A]
6000
5000
iGTO1-C
4000
3000
iGTO1-D-Ind
2000
1000
0
0,000
0,005
0,010
t [s]
0,020
iGTO2-C
6000
iGTO2 [A]
0,015
5000
iGTO2-D-Ind
4000
3000
2000
1000
0
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-49 The currents in GTO1 and GTO2 in the common and the Dual-indicative Snubber
design (Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω).
For other simulations parameters, Fig.8-51 presents the currents in DS1 and Don1 in the two
snubber circuits; it is clearly that the current in DS1 in the new snubber circuit is smaller than the
other snubber circuit. The current value depends on the capacitor value and the performance of the
123
Chapter 8
Simulation results of three-level VSC snubber circuits
turn-off snubber circuit. For the turn-on diode current iDon1, the effect of the parallel inductive is
abundantly clear, this current is produced from the stored energy in the turn-on and stray inductors,
so we can get more recovery energy from the turn-on snubber circuit by using recovery transformer
depending on that the inverter efficiency will be high and the losses will be smaller than that those
of the common snubber circuit.
0
uDs1 [V]
-1000
-2000
-3000
uDs1-C
-4000
uDs1-D-Ind
-5000
0,000
0
0,005
0,010
uDon1 [V]
-1000
0,015
-2000
uDon-C
-3000
uDon1-D-Ind
-4000
t [s]
0,020
-5000
-6000
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-50 The voltages on DS1 and Don1 in the common and the Dual-indicative Snubber design
(Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω).
iDs1 [A]
5000
4000
iDs1-C
3000
iDs1-D-Ind
2000
1000
0
0,000
0,005
0,010
0,015
t [s]
0,020
iDon1 [A]
5000
4000
iDon1-C
3000
iDon1-D-Ind
2000
1000
0
0,000
0,005
0,010
0,015
t [s]
0,020
Fig.8-51 The currents in DS1and Don1 in the common and the Dual-indicative Snubber
design (Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω).
124
Chapter 8
Simulation results of three-level VSC snubber circuits
The freewheel diodes current and the losses energy function are shown in Fig.8-52; the Dfµ
currents in new design is smaller than the other snubber circuit (Fig.8-52(a)), and the total losses
energy function for the dual-inductive snubber circuit is under the other function, this result comes
from the big influence of the new snubber circuit design .
iDf1 [A]
5000
4000
3000
2000
1000
0
0,000
iDf1-D-Ind
0,005
Losses [Ws/p.u.]
0,010
0,015
t [s]
0,020
0,010
0,015
t [s]
0,020
iDf2-C
iDf2-D-Ind
iDf2 [A]
3000
2500
2000
1500
1000
500
0
0,000
150000
iDf1-C
0,005
100000
50000
0
0,000
0,005
0,010
0,015
Losses-C
Losses-D-Ind
t [s]
0,020
(b)
Fig.8-52 (a) The currents in Df2 and Df2 in the common and the Dual-indicative snubber
design and (b) comparison of the total losses in the conventional and the optimized snubber
configuration (energy function) (Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω).
This simulation gives an evolution of the common and dual-inductive snubber circuits. The
dual-inductive snubber circuit represents a compromise between snubber size, the number of
devices which compose the snubber circuit, and the total losses in the snubber circuit and in the
power switching devices. This choice will be suitable and fitting for the high voltage and FACTS
systems, because it has permitted losses, the complexity of the circuit, and the weight and the size
of the snubber circuit.
125
Chapter 9
Experimental investigation on the dual snubber circuit design
9 Experimental investigation on the dual snubber circuit design
9.1 Introduction
For the testing the dual snubber circuit design, a three-level inverter system consisting of four
IGBT transistors, including the protection circuits has been established. To drive this circuit during
the test and prototyping phases, two possibilities were available. The first was to use a computer
program named D-Link. The D-Link takes the simulation output waveforms from
Matlab®/SimulinkTM and generates these waveforms on the serial port of the PC. This cheap
solution was not chosen because D-Link can generate a minimum pulse width of 100 microseconds
while the minimum pulse width required for the driving signals is 500 nanoseconds. The alternative
was using complex programmable logic devices (CPLDs) available from Lattice Semiconductor
Corporation. Even though this solution moves expensive, the CPLDs can be used in the prototyping
phases of future projects. This will distribute the CPLDs cost over many projects and hence makes
this solution feasible. The ispMach4A5 CPLD series was used in the signal generator circuit. In
principle series can generate pulse widths of 5ns and hence it satisfies the required minimum pulse
width constraint. The driving circuit logic was designed and verified using the ispDesignExpert
from Lattice. It contains a schematic editor and a simulator in addition to other utilities. The
designed logic was then programmed into the CPLDs using Lattice-Pro software.
9.2 Design Procedure
Gate4
Gate3
Gate2
Gate1
The design of the drive circuit of the inverter system will be in three steps as in the following:
• The first step was to get the needed IGBT-driving pulses that will be generated. The signals
where taken from Matlab®/SimulinkTM simulations of the real system at 900 Hz switching
frequency. The minimization of the switching frequency from the standard 3 kHz to only 900
Hz was made in order to minimize the logic complexity of the driving circuit so it can fit into
only one CPLD. The simulation results of Matlab®/SimulinkTM are drawn in Fig. 9-1.
1,0
0,8
0,6
0,4
0,2
0,0
0,00
1,0
0,8
0,6
0,4
0,2
0,0
0,00
1,0
0,8
0,6
0,4
0,2
0,0
0,00
1,0
0,8
0,6
0,4
0,2
0,0
0,00
G1
0,01
0,02
0,03
t [s]
0,04
0,03
t [s]
0,04
G2
0,01
0,02
G3
0,01
0,02
0,03
t [s]
0,04
0,03
t [s]
0,04
G4
0,01
0,02
Fig.9-1 Pulse scheme of Matlab®/SimulinkTM at 900Hz
• The second step was to build the needed generating logic in the CPLDs software. The
126
Chapter 9
Experimental investigation on the dual snubber circuit design
schematic of the logic circuit was drawn using ispDesignExpert from Lattice. The
ispDesignExpert contains a schematic editor and a functional and timing simulator. The
schematic of the logic circuit is shown in Fig. 9-2 and the functional simulation results are
presented in Fig. 9-3. After the simulating of the circuit, a JEDEC-file (*.jed) will be
generated. The JEDEC file contains the needed information which will configure the CPLD
according to the designed logic circuit.
Fig. 9-2: The schematic diagram of the driving circuit logic.
• The last step was to transfer the program to the CPLD, then test it with and without the
optocoupler system, finally put them together on the same board as shown in Fig. 9-4.
127
Chapter 9
Experimental investigation on the dual snubber circuit design
Fig. 9-3: Functional simulation results from the ispDesignExpert.
(b)
(a)
(c)
Fig. 9-4: The driving circuit including: (a) The 5 and 15 V power supply from a 12 V source
(b) CPLD Chip and (c) 4 optocoupler.
9.3 Driving the IGBT bridge circuit
The signals that must be generated to drive the IGBT bridge circuit are shown in Fig. 9-3 and
redrawn with a bigger scale in Fig. 9-5 for discussing its characteristic.
Fig. 9-5 One period of the driving signals.
128
Chapter 9
Experimental investigation on the dual snubber circuit design
From Fig.9-5 of the pulse scheme, the following characteristics of the signals help in reducing
the logic complexity by factor of approximately 4:
• Each signal contains an active part consisting of many on and off (logic '1' and logic '0')
pulses and an idle part, in which the IGBT stays either on or off. The active part is
furthermore divided in two identical halves. For example, the active part of G1starts with a
narrow logic '1' pulse. The logic '1' pulse width increases until a maximum range in the
middle is reached, then it decreases in the same manner. As it will be described in section 9-4,
a logic block will be used to generate one half of the active part. Generating the second half is
established by using the same block.
• The G1and the G3 pulses are identical except for that the logic '1' in G1 corresponds to the
logic '0' in G3 and vice versa. This means that G3 is an inverted version of G1 and an inverter
was used to get G1 from G3. The same applies for G2 and G4. An inverter was also used to get
G4 from G2.
• The G2 is shifted from G1 by a half period. Achieving this shifting through a delay-line of
memory elements is ineffective because the clock period is 500 ns while the half period
equals 10 ms and a series of 20000 memory elements will be needed. Investing the firstmentioned property of the signals, the same logic block which would be used to generate the
active part of G1, while G2 is idle, would be used to generate the active part of G2 while G1
goes idle. Additional control blocks were added for this purpose.
A detailed description of the logic circuit used is found in section 9-4.
9.4 The Logic of the Driving Circuit
The active part of the pulses (signals) to be generated constitutes of a number of logic '1' and
logic '0' pairs of pulses. The used concept here is to map the pulse duration in the waveform to an
integer number. The integer number is derived from dividing the pulse duration by the clock period
which will drive the logic circuit. A series of integers result for each signal. A digital counter is then
used to count each of the integers in the series. At the end of counting, the counter toggles the
output signal from '1' to '0' or from '0' to '1' and another counter starts counting for the next integer
in the series.
In other words, the series of integers representing the active part of the signal are
implemented on the circuit level as a series of digital counters. The counters outputs are connected
through an OR gate to a toggle flip-flop (T-flip-flop). Each counter in the series is also connected to
its neighbours through a logic block as shown in Fig. 9-6. This logic block does the two tasks:
•
When the counter finishes counting, the control block enables the next counter in the series.
•
One clock later, it gives the counter itself a reset pulse and disables it from counting. This
prepares the counter for the next counting.
129
Chapter 9
Experimental investigation on the dual snubber circuit design
Fig. 9-6: The logic block schematic diagram.
The series of counters counts continuously. The counting begins with first counter in the
series (first_1 output in Fig. 9-7), continues until the tenth forming the first half of active period
(forward counting). Backward counting starts directly that after and continues until the first counter
(first_2 output in Fig. 9-7). Each forward and backward run of the counters in the series generates
on active part. The control block shown in Fig. 9-7 switches the output signal between the T-flipflop dedicated to generate G1 and the T-flip-flop dedicated to generate G2. When the active part of,
say, G1 is being generated, the G2 flip-flop keeps its last state, and hence, the idle part is generated
automatically.
One problem arises at the two ends of the counters' series. The first counter must count twice,
one in the forward counting and one in the backward counting. Instead of, two identical counters
(with first_1 and first_2 outputs in Fig. 9-7) were used. The count of the last counter in series (tenth
output) was doubled to generate the wide pulse in the middle of the active part as shown in Fig. 9-5.
Fig. 9-7: Control block of the driving circuit.
130
Chapter 9
Experimental investigation on the dual snubber circuit design
9.5 Output Circuit
The driving circuit consists of the ispMach4A5 128/256 CPLD from Lattice Semiconductor
Co. and four optocouplers, the HCPL 316J, one for each of the driving signals. The CPLD was
programmed to generate the signals G1, G2, G3, and G4. The optocouplers serve two purposes:
• Affording electrical ground isolation between the CPLD side and the IGBTs side. This is
important to protect the CPLD from the spikes that may be generated during high-speed
switching of the IGBTs.
• Making the interface between the 5V logic levels at the CPLD side and the less than 12V
driving levels at the IGBTs side.
9.6 Experimental results
Three-level IGBTs (MG150J2YS1, CM100DU-12F) inverter system was constructed for
testing the performance of the dual-use and dual-inductive snubber circuit design especially in the
protection from over-voltage across the power switching devices (see Fig-8-7). The parameters of
the inverter system and the load information are shown in Table 9-1, the power factor of the loads is
Pf =0.8. The total test system consists of: adjustable transformer supplies the three-phase diode
bridge-rectifier, the rectifier is connected to two capacitors (DC-link) which is coupled to the threelevel IGBT inverter system, the inverter will be loaded with ohmic- and inductive-burden. The
voltage will be measured across the IGBTs, snubber-diode and the load, and so the load current.
The measuring system consists of amplifier; oscilloscope, ampere meter, current-transformer and
voltmeter (see Fig 9-8).
CS [µF]
RS [Ω]
LON [µH]
RON [Ω]
LP [µH]
IO [A]
Rload [Ω]
Lload [mH]
0.47
33
1
0.33
2
0.5
50,66
121,31
Table 9.1 Dual-use- and dual-inductive snubber circuits information.
Load
3 level IGBT
inverter system
3~
Diode bridge
rectiefer
Driver
Vin1 Vout1
0
Vout1
Vout2
0
0
Amplifier
Vin2 Vout2
0
0
Fig. 9-8 Control blocks of the complete Three-level IGBT inverter system.
131
Chapter 9
Experimental investigation on the dual snubber circuit design
9.6.1 Test of the dual-use snubber circuit
In order to test the effectiveness of the dual-use snubber circuit regarding the reduction the overvoltage transients during the turn-off process of the IGBTs in the mentioned three-level inverter, the
following procedure was done.
Fig.9-9 shows the curves of the emitter-collector voltage across IGBT1, 2”UIGBT1, 2” during two
periods. It is clearly that the voltage wave looks almost identical to the simulated result in Fig.8-43.
The disturbance of the cable and measuring equipment produces some spikes in the voltage
waveform; another reason was the amplifier which magnifies the waveform and the spikes for ten
times. The measured current in IGBT1, 2 is presented in Fig.9-10 in comparison the simulated and
the measured currents have the same curves so the current peaks are smaller and reduced (see Fig.844 and Fig 9-10).
The voltage waveforms across the turn-off snubber diode DS1 and turn-on snubber diodeDON1
are illustrated in Fig.9-11, it can easily be seen that the over-voltage waveforms are the same as
those resulted from the simulation results in Matlab®/SimulinkTM. The over-voltage is within the
range of the rated voltage values.
The output voltage and current of inverter are shown in Fig.9-12 for an inductive load. The load
voltage is a square wave and alternates between +Ed and - Ed While the load current waveform is a
sinusoidal wave.
UIGBT1
UIGBT2
Dual-use snubber circuit design
Fig. 9-9 Voltage waveforms across IGBT1, 2.
132
Chapter 9
Experimental investigation on the dual snubber circuit design
IIGBT1,2
Dual-use snubber circuit design
Fig. 9-10 Current waveforms through IGBT1, 2.
UDS1
UON1
Dual-use snubber circuit design
Fig. 9-11 Voltage waveforms across DS1, DON1.
9.6.2 Test of the dual-inductive snubber circuit
In this experimental investigation, the parallel impedance will be added to the snubber circuit
133
Chapter 9
Experimental investigation on the dual snubber circuit design
of the three-level IGBTs inverter system. Then the waveforms of the voltage and current of the
IGBTs, the diodes and the load will be observed (see Fig.8-8).
UL
IL
Dual-use snubber circuit design
Fig. 9-12 Load voltage- and current-waveforms.
The voltage and current waveforms of IGBT1, 2 are illustrated in Fig.9-13 and Fig.9-14. The
voltages across the IGBTs are the same of the simulated result in Matlab®/ SimulinkTM which are
presented in Fig.8-48, and also the voltages are much more suppressed than the dual-use snubber
circuit design in the first experiment (see Fig.9-13 and Fig.9-9). The current waveforms in IGBT1, 2
are identical with those of Matlab®/ SimulinkTM which are shown in Fig.8-49 and there are no
peaks. The current is smoother than the current waveform of the first experiment. The voltages
across the snubber circuit diodes are better in comparison with the dual-use snubber circuit in first
test (see Fig.9-15). The parallel inductive is effective in the turn-on snubber circuit in limiting the
current and the measured current through it is much bigger than the current through the turn-on
snubber resistor, so the over-voltages will be suppressed more and more and the current is restricted
another time. Fig.9-16 presents the load voltage and current of the inverter system. Also, the
voltage was square waveform changing between ± Ed and the current is of a sinusoidal wave. The
load voltage waveform in this test is better than in the first test and much more restrained. In the
other side, the current waveform is identical, pure and no more peaks.
The dual-inductive snubber design is suitable for high rated power applications for e.g., in the
converter systems which are used in HVDC and FACTS devices. The most important feature of the
dual-inductive snubber circuit is the new simple structured snubber. So, the number of required
components will be minimized and the performance of the snubber circuits is improved. The dualinductive snubber design improves the efficiency due to low snubber losses and a better clamping
134
Chapter 9
Experimental investigation on the dual snubber circuit design
the over-voltages across the switching devices. Additionally it reduces the manufacturing costs and
the complexity so the converter size can be reduced. The experimental investigation were carried
out and very good findings. It shows the advantages of the dual-inductive snubber circuit especially
for the operation close to the limits of the SOA (safety operating area).
UIGBT1
UIGBT2
Dual-indicative snubber circuit design
Fig. 9-13 Voltage waveforms across IGBT1, 2.
IIGBT1,2
Dual-inductive snubber circuit design
Fig. 9-14 Current waveforms through IGBT1, 2.
135
Chapter 9
Experimental investigation on the dual snubber circuit design
UDS1
UON1
Dual-indicative snubber circuit design
Fig. 9-15 Voltage waveforms across DS1, DON1.
UL
IL
Dual-indicative snubber circuit design
Fig. 9-16 Load voltage- and current-waveforms.
136
Chapter 10
Conclusion
10 Conclusion
In this work new and simple snubber circuits for three-level inverter system were proposed
and suggested, which have several new features. These suggested circuits were tested by using a
Matlab®/SimulinkTM Simulation which was based on a single phase model of a MV three level
inverter system. The overvoltages and the losses are compared between the conventional and the
other suggested snubber circuit configurations: the double snubber circuit, the optimized snubber
circuit design, the dual-use snubber circuit and the dual-inductive snubber circuit. The results
clarify the advantages of the proposed designs especially for the operation close to the limits of the
SOA (Safety Operating Area) as in the following:
•
•
•
•
The double snubber circuit, which is suitable for high voltage and high power more level
converter systems and FACTS devices, has the advantages of both the RC and RCD
snubber circuit. This means, the facility to minimize both, the over voltage and the losses
in the entire circuit. Therefore the additional RC circuit can compensate the disadvantage
of the conventional RCD snubber design, that the effective value for RS during the
charging of the snubber capacity CS is essentially zero. But still the number of the total
snubber circuit elements is large, so the total losses will be bigger compared ti the
following snubber circuit designs.
The optimized snubber design is proposed to be used in high voltage and high power more
level converter systems and partially FACTS devices. The optimized snubber design is
based on a new and simple structured snubber, which depends on passive elements. As a
result, the number of required components is reduced, and the reliability increases
distinctly. The presented optimized snubber design provides several additional advantages:
It increases the optional performance of the three level converters due to the lower
clamping over-voltages across the switching devices, it improves the efficiency because of
the lower snubber- and total losses, its suitable structure can be extended to energy
recovery snubbers and there is no unbalancing problem. Additionally, the manufacturing
costs, the complexity and therefore the converter size can be reduced.
The dual-use snubber design has the same advantages of the optimized snubber design;
and more than that: the new position of turn-off snubber resistor RS which has two new
functions the first is during the discharge process of CS1, and the second is during the
charging process of CS2. So the resistor has a damping effect of the over-voltage across the
power switching device while charging process of CS2, this means that the over-voltage
will be reduced, especially in the beginning of turn-off process.
The dual-inductive snubber design is also expedient for using in high- voltage and power
level converter systems and a FACTS devices. The dual-inductive snubber design has the
same structure und advantages of the dual-use snubber circuit. But the added inductance
suppresses the over-voltage across the switching device more effectively and improves the
efficiency due to minimize the losses.
In the experimental part of the dissertation, the dual-use and dual-inductive snubber circuits of
three-level inverter systems were tested. The investigation results show that:
• The dual-use snubber circuit has perfectly reduced the over-voltage across the IGBTs,
restricted the current changing, and suppressed the over-voltages across the snubber circuit
diodes. The results were compatible with those of the simulation results in
Matlab®/SimulinkTM.
• The dual-inductive snubber circuit tests were carried out and also the results correspond to
the simulation results of the Matlab®/SimulinkTM studies. The added inductive in the turn
on snubber circuit absorbs a big part of the current while a small part will flow through the
137
Chapter 10
Conclusion
turn on snubber circuit resistor which causes the over-voltage. The value of the measured
current through the resistor was about 10% of the total current of the turn on snubber
circuit. The shape of the load voltages and currents were much better than in the dual-use
snubber circuit. So, the dual-inductive snubber design will improve the total efficiency due
to the mentioned features.
138
Chapter 11
Zusammenfassung
11 Zusammenfassung
In
der
vorliegenden
Arbeit
werden
Schutzbeschaltungen
für
DreipunktZwischenkreisstromrichter entwickelt und vorgeschlagen, die neuartige Möglichkeiten beinhalten.
Die vorgeschlagenen Schaltungen wurden mit Matlab®/SimulinkTM Modellen entwickelt und
überprüft. Verwendet wurde ein einphasiges Modell eines Dreipunktumrichters für
Mittelspannungsanwendungen. Die Überspannungen und die Verluste werden zwischen der
konventionellen und den hier vorgeschlagen Schutzbeschaltungen verglichen: der
Doppelschutzbeschaltung oder auch „Double Snubber Circuit“, der optimierten Schutzbeschaltung,
oder „Optimized Snubber Circuit Design“, der Mehrzweckschutzbeschaltung oder „Dual-use
Snubber Circuit“ und der Doppel-induktiv Schutzbeschaltung oder „Dual-inductive Snubber
Circuit“. Die Resultate erklären die Vorteile der vorgeschlagenen Entwürfe besonders für den
Betrieb nahe der Grenze der SOA (Sicherer Arbeitsbereich) wie folgt:
•
Der „Double Snubber Circuit“, der für Hochspannungsanwendungen und mehrstufige
Umrichtersysteme für FACTS-Anlagen zweckmäßig ist, vereint die Vorteile der RC und
RCD Schutzbeschaltungen. Die zusätzliche RC Schutzbeschaltung kann den Nachteil der
konventionellen RCD Schutzbeschaltung ausgleichen, so dass der wirkungsvolle Wert von
RS während der Aufladung des Schutzbeschaltungskondensators CS naher Null ist. Jedoch
ist die Zahl der Beschaltungselemente groß, und somit sind die Gesamtverluste größer als
bei den andren vorgeschlagenen Schutzbeschaltungsentwürfen.
•
Die vorgeschlagene optimierte Schutzbeschaltung oder „Optimized Snubber Circuit“ kann
in mehr stufige Hochspannungs- und Hochleistungsstromrichtern und teilweise in FACTSAnlagen verwendet werden. Der optimierte Schutzbeschaltungsentwurf basiert auf einer
neuen und einfacher strukturierten Schutzbeschaltung, insbesondre bezügliche der
passiven Elemente. Als Ergebnis, wird die Zahl der erforderlichen Bauteile verringert und
die
Zuverlässigkeit
wird
deutlich
erhöht.
Der
dargestellte
optimierte
Schutzbeschaltungsentwurf gewährt mehrere zusätzliche Vorteile: Er erhöht die
Leistungsfähigkeit des Dreipunkt-Zwischenkreisstromrichters wegen der niedrigeren
Überspannungen über dem Leistungshalbleiter, verbessert die Leistungsfähigkeit, wegen
der geringeren Schutzbeschaltung und die Gesamtverluste. Die verwendete Struktur kann
zu einer Schutzbeschaltung mit Energierückgewinnung erweitert werden und es gibt keine
Probleme bzgl. einer unsymmetrischen Spannungsaufteilung. Zusätzlich können die
Herstellungskosten, die Komplexität und damit die Konvertergröße verkleinert werden.
•
Der zweifache Schutzbeschaltungsentwurf oder „Dual-use snubber circuit“ hat neben
denVorteilen des optimierten Schutzbeschaltungsentwurfs weitere: die neue Position des
Abschaltschutzwiderstandes RS, besitzt zwei neue Funktionen. Die Einerseits während des
Entladungsprozesses von CS1 und andererseits während des Aufladungsprozesses von CS2.
Der Widerstand hat einen Dämpfungseffekt bezüglich der Überspannungen über den
Leistungshalbleitern während der Aufladung von CS2. Dies gilt insbesondere für den
Beginn des Abschaltprozesses.
•
Die Doppel-induktive Schutzbeschaltung oder „Dual-inductive Snubber Circuit“ ist
ebenfalls für die Verwendung in mehrstufigen
Hochspannungs- und
Hochleistungsstromrichtersystemen und FACTS-Anlagen geeignet. Die doppel-induktive
Schutzbeschaltung hat in wesentlichen die Struktur und die Vorteile des zweifachen
Schutzbeschaltungsentwurfs. Jedoch unterdrückt die zusätzliche Induktivität die
139
Chapter 11
Zusammenfassung
Überspannungen über den Leistungshalbleitern noch stärker und verbessert die
Leistungsfähigkeit durch eine Minimierung der Gesamtverluste.
Im experimentellen Teil der Dissertation, wurden die zweifache- und die doppel-induktive
Schutzbeschaltung der Dreipunktstromrichtersysteme überprüft. Die Resultate zeigen:
•
Die zweifache Schutzbeschaltung konnte die Überspannungen über den IGBTs in sehr
guter Weise verringern, die Flankensteilheit des Stromes wird begrenzt und die
Überspannungen über den Schutzbeschaltungsdioden unterdrückt. Die Resultate stimmen
mit denen der Simulation in Matlab®/SimulinkTM überein.
•
Bei der doppelinduktiven Schutzbeschaltung entsprechen die Messergebnisse den
Simulationsergebnissen mit Matlab®/SimulinkTM. Die zusätzliche Induktivität in der
Einschaltschutzbeschaltung nimmt einen großen Teil des Stromes auf, während ein
kleiner Teil den Einschaltwiderstand durchfließt. Die Spannungs- und Stromform
bezüglich der Last konnte gegenüber der zweifachen Schutzbeschaltung weiter verbessert
werden. Somit optimiert die doppel-induktive Schutzbeschaltung die GesamtLeistungsfähigkeit des Stromrichtersystems weiter.
140
Appendix A1
Abbreviation and symbols
Appendix A1 Abbreviations and symbols
INom
: Forward conduction mode.
VD-Nom
: Forward blocking mode.
VR-Nom
: Reverse blocking mode.
t1
: Needed time for junction biased reverse.
t2
: Charge storage time in the bulk semiconductor material.
trr
: Reverse recovery time.
Irr
: Reverse recovery current.
IRR
: Peak reverse recovery current.
Qrr
: Reverse recovery charge.
VF
: Diode forward voltage drop.
ICE
: Collector-Emitter current.
IC
: Collector current.
VCE
: Collector-Emitter voltage.
VGE
: Gate-Emitter voltage
td(on)
: Turn-on delay time.
tr
: Rise time.
td(off)
: Turn-off delay time.
tf
: Fall time.
VGET
: Threshold voltage of the Gate-Emitter voltage.
SCR
: Silicon Controlled Rectifier.
IG
: Gate current.
td
: Delay time.
IT
: Full anode conducting current.
IRM
: Maximum reverse-recovery current.
VRRM
: Maximum reverse-recovery voltage.
VAK
: Anode-Cathode voltage.
tq
: Turn-off time of the Thyristor.
Tj
: The temperature of the junction.
IT
: Thyristor current.
VR
: Reverse recovery voltage.
VDRM
: Peak Off-state voltage.
VG
: Gate voltage.
rv
: Radii of GTO wafer.
Vd
: DC voltage.
FACTS
: Flexible AC Transmission Systems.
141
Appendix A1
Abbreviation and symbols
Vo
: Output voltage.
io
: Output current.
VS
: DC voltage.
Id
: Supply DC current.
δ
: Phase angle (pulse width).
Von
: The rms value of the nth order AC output voltage.
Vo1
: The rms fundamental output voltage.
eC
: Maximum value of the sinusoidal modulating voltage.
fo
: Output frequency.
vtri
: Saw-wave voltage.
vtri,max
: Maximum value of the saw-wave voltage.
Vo1max
: Fundamental output voltage.
vABC
: Line-line voltages.
VAN:
: Line-neutral voltage for phase A.
vl-N
: Line-neutral voltage.
mf
: Frequency modulation ratio.
Vmax
: Peak value of the input AC voltage.
Lσ
: Stray inductance.
IO
: Load current.
RS
: Snubber resistor.
CS
: Snubber capacitor.
vCS
: Snubber capacitor voltage.
ω
: Angular frequency.
Irr
: Reverse recovery current.
Cbase
: Baseline capacitor.
vCS ,max
: Maximum value of the snubber capacitor voltage.
vD f
: Free-wheel diode voltage.
Rbase
: Baseline snubber resistor.
Ropt
: Optimum value of the snubber resistor.
WR
: Energy loss in RS
WCS
: Energy stored in CS.
Wtot
: Total energy dissipated in the diode and its snubber resistor.
xC
: Line impedance.
VLL
: Line-to-line voltage.
Id
: Load current.
Wsnubber
: Total energy loss in the snubber circuit.
142
Appendix A1
Abbreviation and symbols
Psnubber
: Total power loss in the snubber circuit.
iCS
: Charging current of the snubber capacitor.
tfi
: Current fall time.
ton state
: Turn-on time
FBSOA
: Forward biased safe operation area.
RBSOA
: Reverse biased safe operating area
∆VCE
: Over-voltage across the transistor.
VCE,max
: The maximum over-voltage across the transistor.
COV
: Over-voltage capacitor.
ROV
: Over-voltage resistor.
k
: Constant factor.
tri
: Current rise time.
τC
: Time constant of the turn-off snubber circuit.
toff state
: Turn-off time.
RLS
: The resistor of the turn-on inductance.
τL
: Time constant of the turn-on snubber circuit.
VCESP
: Turn off surge voltage peak.
VFM
: Transient forward voltage drops in the snubber diode.
VCEP
: Snubber capacitor peak voltage.
f
: Switching frequency.
P
: Power dissipation caused by the snubber resistor.
σ
: On time (Pulse width duration per half cycle for ON period).
α
: Off time. (Pulse width duration per half cycle for Off period)
V1max
: Maximum value of the fundamental voltage V1.
SOA
: Safe Operating Area.
CP
: Parallel snubber capacitor.
RP
: Parallel snubber resistor.
143
Appendix A2
List of Figures
Appendix A2 List of Figures
Fig.3-1 Device operational states for (a) symmetrical device and (b) asymmetrical device. ..............7
Fig.3-2 Diode: (a) diode symbol, (b) diode structure, and (c) more detailed diode structure. ..........11
Fig.3-3 (a) Diode forward- and (b) reverse-recovery-biased.............................................................11
Fig.3-4 Diode characteristics. ............................................................................................................13
(a).Input waveform applied to the diode in Fig.3-3(a), (b) The excess-carrier density at the junction,
(c) The diode current, (d) The diode voltage, (e) Diode reverse-recovery characteristics. ...............13
Fig.3-5 IGBT transistor: (a) IGBT transistor structure and the location of the equivalent circuit, (b)
the equivalent circuit, (c) IGBT symbol. ...........................................................................................14
Fig.3-6 IGBT current-voltage (IC-VGE) characteristics for a given value of VCE. ..............................15
Fig.3-7 Switching characteristic of an IGBT. ....................................................................................16
Fig.3-8 Thyristor (a) Thyristor symbol, (b) Thyristor structure, (d) two-transistor structure, and (d)
Thyristor equivalent circuit................................................................................................................18
Fig.3-9 Switching characteristics of the thyristor. .............................................................................19
Fig.3-10 Gate turn-off thyristor (a) GTO -symbol, (b) -structure and (c) -equivalent circuit. ..........21
Fig.3-11 (a) A picture of GTO surface (b) A picture of GTO wafer including definition of radii rv.21
Fig.3-12 GTO turn-on and turn-off process: (a) turn-on and (b) turn-off..........................................23
Fig.3-13 Conduction and conventional Turn-off of a GTO...............................................................24
Fig.3-14 MOS Turn-off (MTO) thyristor (a) MTO symbol, (b) MTO structure; (c) MTO equivalent
circuit, and (d) more detailed equivalent circuit. ...............................................................................25
Fig.3-15 Emitter Turn-off (ETO) thyristor: (a) ETO symbol, (b) ETO equivalent circuit, and (c)
ETO structure.....................................................................................................................................25
Fig.3-16 IGCT thyristor (a) IGCT symbol (b) IGCT structure with a Gate-Commutated Thyristor
and reverse diode. ..............................................................................................................................26
Fig.3-17 MOS-Controlled Thyristor (MCT) (a) MCT symbol, (b) MCT structure (c) MCT
equivalent circuit................................................................................................................................27
Fig.3-18 Control characteristics of power switching devices. ...........................................................28
Fig.4-1 Basic types of FACTS Controllers (a) general symbol for FACTS controller; (b) series
controller; (c) shunt controller; (d) unified series-series controller; (e) coordinated series and shunt
controller; (f) unified series-shunt controller; (g) unified controller for multiple lines; (h) series
controller with storage; (i) shunt controller with storage; (j) unified series-shunt controller with
storage. ...............................................................................................................................................32
Fig.4- 2 Shunt-connected Controllers (a) Static Synchronous Compensator (STATCOM) based on
voltage-sourced and current-sourced converter; (b) STATCOM with storage, i.e., Battery Energy
Storage System (BESS), Superconducting Magnet Energy System and large capacitor; (c) Static
VAR Compensator (SVC); (d) Static VAR Generator (SVG), Static VAR System, ThyristorControlled Reactor (TCR), Thyristor-Switched Capacitor (TSC), and Thyristor-Switched Reactor
(TSC); (d) Thyristor-Controlled Braking Resistor.............................................................................35
Fig.4-3 (a) Static Synchronous Compensator (SSSC) (b) SSSC with storage; (c) ThyristorControlled Series Capacitor (TCSC) and Thyristor Switched Series Capacitor (TSSC), and (d)
Thyristor-Controlled Series Reactance (TCSR) and Thyristor-Switched Series Reactance. ............39
Fig.4-4 (a) Thyristor-Controlled Phase-Shifting Transformer (TCPST) or Thyristor-Controlled
Phase Angle Regulator (TCPR); (b) Unified Power Flow Controller (UPFC)..................................41
Fig.4- 5 Various other controllers (a) Thyristor-Controlled Voltage Limiter (TCVL), (b) ThyristorControlled Voltage Regulator (TCVR) based on tap changer, (c) Thyristor-Controlled Voltage
Regulator (TCVR) based on voltage injection...................................................................................41
Fig.5-1 Basic principle of Voltage-Sourced Converter: (a) Valve for a Voltage-Sourced Converter;
(b) Voltage-Sourced Converter concept; (c) Single-valve operation.................................................44
Fig.5-2 Single-phase half-bridge converter. ......................................................................................45
144
Appendix A2
List of Figures
Fig.5-3 Single-phase full-bridge converter with RL-load..................................................................46
Fig.5-4 The regulation of the output voltage by means of a phase-controlled. .................................47
Fig.5-5 The regulation of the output voltage by means of a DC-DC converter. ...............................47
Fig.5-6 The waveform of the output voltage of Full-bridge converter..............................................48
Fig.5-7 Three-phase full-wave bridge converter................................................................................49
Fig.5-8 The switching scheme of three-phase Voltage-Sourced Converter. .....................................50
Fig.5-9 The switching scheme for three-phase VSC 120° conducting stead of 180°........................51
Fig.5-10 One phase-leg of a three-level converter.............................................................................52
Fig.5-11 Operation of thee-level converter, output AC voltage. .......................................................53
Fig.5-12 Operation of a PWM converter with switching frequency of three times the fundamental
frequency............................................................................................................................................54
Fig.6-1 Voltage-sourced and Current-Sourced Converter concepts: (a) voltage sourced converter;
(b) Current-Sourced Converter. .........................................................................................................55
Fig.6-2 Types of Current-Sourced Converter, (a) diode rectifier; (b) Thyristor line-commutated
converter, (c) self-commutated converter. .........................................................................................56
Fig.6-3 Diode bridge rectifier. ...........................................................................................................58
Fig.6-4 A three-phase bridge rectifier................................................................................................58
Fig.6-5 The rectifier and the phase controller for a half-wave converter. .........................................59
Fig.6-6 A single-phase fully-controlled bridge rectifier (thyristor with p=2). ..................................60
Fig.6-7 Three-phase, fully-controlled bridge converter circuit..........................................................60
Fig.6-8 Self-commutating current-sourced converter: (a) six-pulse converter, (b) commutation
process, (c) current waveforms, and (d) system interface..................................................................62
Fig.7-1 (a) A step-down converter circuit with stray inductance and a snubber circuit for the freewheel diode, (b) the Diode reverse-recovery current and diode voltage. ..........................................65
Fig.7-2:(a) Equivalent circuit of the step-down converter at the instant of diode reverse-recovery
current snap-off (b) the simplification that results when the snubber resistance is zero and (c) The
voltage and current waveforms for RS = 0 and CS = Cbase..................................................................67
Fig.7-3 Equivalent circuit with snubber resistance RS. ......................................................................67
Fig.7-4 The current and the voltage waveforms after diode snaps-off at t = 0..................................68
Fig.7-5 Maximum over-voltage across the diode as a function of the snubber resistance for a fixed
value of the snubber capacitance. ......................................................................................................69
Fig.7-6 Snubber energy loss and the maximum diode voltage for the optimum value of the snubber
resistance RS as a function of the snubber capacitance CS ................................................................69
Fig.7-7 Turn-off snubbers for Thyristors in a three-phase line-frequency converter circuit: (a) threephase line-frequency converter, (b) trigger time, and (c) the equivalent circuit................................71
Fig.7-8 (a) A step-down converter circuit with stray inductance shown explicitly with (b) associated
switching trajectory and (c) the current and voltage waveforms during turn-on and turn-off...........72
Fig.7-9 (a) turn-off snubber circuit, (b) its equivalent circuit during the transient and (c) current and
voltage waveforms during the turn-off transient. (The shaded areas in Fig.7-9 (c) represent the
charge put on the snubber capacitance during turn-off that will be dissipated in the power switching
device at the next turn-off.)................................................................................................................74
Fig.7-10 Switching trajectory during turn-off with various values of snubber capacitance CS.........75
Fig.7-11 Effect of the snubber capacitance CS on the turn-off transient without (a) snubber
resistance RS and (b) with the resistance............................................................................................76
Fig.7-12 Turn-off energy dissipation in the power switching device and the snubber resistance RS as
a function of the snubber capacitance CS. ..........................................................................................77
Fig.7-13 (a) Over-voltage snubber, (c, b) its equivalent circuit during transient turn-off, (d) the
collector-emitter voltage with and without the snubber.....................................................................78
Fig.7-14 Turn-on snubber circuit (a) in series with the power switching device or (b) in series with
the free-wheel diode, (c) The power switching device voltage and current waveforms for small
value of Lσ and (d) for Large values of Lσ. ........................................................................................80
145
Appendix A2
List of Figures
Fig.7-15 A modified circuit with an over-voltage snubber, a turn-on snubber, and turn-off snubber;
the Undeland snubber for step-down converter. ................................................................................81
Fig.7-16 Step-down converter circuit using a GTO as the switching device with turn-on and turn-off
snubbers .............................................................................................................................................82
Fig.7-17 Test chopper circuit. ............................................................................................................83
Fig.7-18 IGBT Switching waveforms during the turn-off and turn-on processes .............................84
Fig.7-19 Schematic type of individual snubber circuits: (a) RC snubber circuit, (b) Charge discharge
RCD snubber circuit, (c) Discharge suppressing RCD snubber circuit, (d) C snubber circuit and (e)
RCD snubber circuit...........................................................................................................................86
Fig.7-20 Turn-off locus waveform of IGBT......................................................................................87
Fig.7-21 Voltage and current waveforms at turn-off. IGBT..............................................................87
Fig.8-1 Single phase of a three level converter (common snubber circuit). ......................................90
Fig.8-2 Proposed double snubber circuit configuration in a three level inverter system...................91
Fig.8-3 An optimized snubber design for Single phase three level GTO inverters. ..........................92
Fig.8-4 Commutation path of the transition form S0 to S1: (a) initial state, (b) phase 1 and (c) phase2..........................................................................................................................................................94
Fig.8-5. Commutation path of the transition form S1 to S0: (a) initial state, (b) phase 1 and (c) phase2..........................................................................................................................................................95
Fig.8-6 Commutation path of the transition form S0 to S-1: (a) initial state, (b) phase 1 and (c) phase2..........................................................................................................................................................96
Fig.8-7 One phase of a three level inverter with the new dual-use snubber circuit design. ..............98
Fig.8-8 One phase of a three-level inverter with the new dual-inductive snubber circuit.................99
Fig.8-9 The main flow chart of the PWM in Matlab®/SimulinkTM. ...............................................100
Fig.8-10 The internal part of the PWM in Matlab®/SimulinkTM. ...................................................101
Fig.8-11 The comparison functions of the PWM in Matlab®/SimulinkTM. ....................................101
Fig.8-12 the output of the PWM in Matlab®/SimulinkTM...............................................................102
Fig.8-13 The voltages on GTO1 and GTO2 in the common and the double snubber configuration
(CP=0.05µF).....................................................................................................................................103
Fig.8-14 The currents in GTO1 and GTO2 in the common and the double snubber configuration
(CP=0.05µF).....................................................................................................................................103
Fig.8-15 The currents in Df1 and Df2 in the common and the double snubber Configuration
(CP=0.05µF).....................................................................................................................................104
Fig.8-16 The voltages on DS1 and DS2in the common and the double snubber configuration
(CP=0.05µF).....................................................................................................................................104
Fig.8-17 The currents in DS1 and DS2 in the common and the double snubber configuration
(CP=0.05µF).....................................................................................................................................105
Fig.8-18 The total losses (energy function) in RS1 in the conventional- and in RS1, RP1 in the
proposed double snubber design over one cycle of the fundamental frequency (CP=0.05µF)........105
Fig.8-19 The voltages in GTO1 and GTO2 in the common and the double snubber configuration
(CP=0.01µF).....................................................................................................................................106
Fig.8-20 The currents in GTO1 and GTO2 in the common and the double snubber configuration
(CP=0.01µF).....................................................................................................................................106
Fig.8-21 The currents in DS1 and DS2 in the common and the double snubber configuration
(CP=0.01µF).....................................................................................................................................107
Fig.8-22 The voltages on DS1 and DS2 in the common and the double snubber configuration
(CP=0.01µF).....................................................................................................................................107
Fig.8-23 The currents in Df1and Df2 in the common and the double snubber Configuration
(CP=0.01µF).....................................................................................................................................108
Fig.8-24 The total losses (energy function) in RS1 in the conventional and in RS1, RP1 in the proposed
double snubber design over two cycles of the fundamental frequency (CP=0.01µF)......................108
146
Appendix A2
List of Figures
Fig.8-25 The voltages in GTO1 and GTO2 in the common and the optimized snubber design. ......109
Fig.8-26 The currents in GTO1 and GTO2in the common and the optimized snubber design.........110
Fig.8-27 (a) The currents in Df1 and Df2 and comparing the total losses in conventional and (b) the
optimised snubber configuration (energy function).........................................................................110
Fig.8-28 The current in DS1 and Don1 in the common and the optimized snubber configuration. ...111
Fig.8-29 The voltages in DS1 and Don1 in the common and the optimized snubber configuration. .111
Fig.8-30 The voltages on GTO1 and GTO2 in the common and the optimized snubber design
(CS=0.5µF, RS=5Ω)..........................................................................................................................112
Fig.8-31 The currents in GTO1 and GTO2 in the common and the optimized snubber design
(CS=0.5µF, RS=5Ω)..........................................................................................................................112
Fig.8-32 Currents in DS1 and Don1 in the common and the optimized snubber configuration
(CS=0.5µF, RS=5Ω)..........................................................................................................................113
Fig.8-33 Voltages on DS1 and Don1 in the common and the optimized snubber configuration
(CS=0.5µF, RS=5Ω)..........................................................................................................................113
Fig.8-34 (a) The currents in Df2 and Df2 in the common and the optimized snubber configuration,
and (b) the comparison of the total losses in the conventional and the optimized snubber
configuration (energy function) (CS=0.5µF, RS=5Ω). .....................................................................114
Fig.8-35 The voltages in GTO1 and GTO2 in the common and the optimized snubber design
(CS=0.25µF, RS=2Ω)........................................................................................................................115
Fig.8-36 The currents in GTO1 and GTO2 in the common and the optimized snubber design
(CS=0.25µF, RS=2Ω)........................................................................................................................115
Fig.8-37 (a) The currents in Df2 and Df2 in the common and the optimized snubber configuration,
and (b) the comparison of the total losses in the conventional and the optimized snubber
configuration (energy function) (CS=0.25µF, RS=2Ω). ...................................................................116
Fig.8-38 The voltages on GTO1 and GTO2 in the common and the dual-use snubber configuration
(CS-D=0.5µF, RS=2.5Ω). ...................................................................................................................117
Fig.8-39 The currents in GTO1 and GTO2 in the common and the dual-use snubber configuration
(CS-D=0.5µF, RS=2.5Ω). ...................................................................................................................117
Fig.8-40 The currents in DS1 and Don1 in the common and the dual-use snubber configuration (CSD=0.5µF, RS=2.5Ω). .........................................................................................................................118
Fig.8-41 The voltages in DS1 and Don1 in the common and the dual-use snubber configuration (CSD=0.5µF, RS=2.5Ω). .........................................................................................................................118
Fig.8-42 (a) The currents in Df2 and Df2 in the common and dual-use snubber configuration, and (b)
comparison of the total losses in the conventional and the optimized snubber configuration (energy
function) (CS-D=0.5µF, RS=2.5Ω). ...................................................................................................119
Fig.8-43 The voltages in GTO1 and GTO2 in the common and the dual-use snubber configuration
(CS-D=0.25µF, RS=2.5Ω). .................................................................................................................120
Fig.8-44 The currents in GTO1 and GTO2 in the common and the dual-use snubber configuration
(CS-D=0.25µF, RS-D =2.5Ω)...............................................................................................................120
Fig.8-45 The voltages in DS1 and Don1 in the common and the dual-use snubber configuration (CSD=0.25µF, RS-D =2.5Ω).....................................................................................................................121
Fig.8-46 The voltages in DS1 and Don1 in the common and the dual-use snubber configuration (CSD=0.25µF, RS-D=2.5Ω). ....................................................................................................................121
Fig.8-47 (a) The currents in Df2 and Df2 in the common and dual-use snubber configuration and (b)
comparison between the total losses in the conventional and the optimized snubber configuration
(energy function) (CS-D=0.25µF, RS-D=2.5Ω). .................................................................................122
Fig.8-48 The voltages on GTO1 and GTO2 in the common and the Dual-indicative Snubber design
(Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω). ...............................................................................123
Fig.8-49 The currents in GTO1 and GTO2 in the common and the Dual-indicative Snubber design
(Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω). ...............................................................................123
147
Appendix A2
List of Figures
Fig.8-50 The voltages on DS1 and Don1 in the common and the Dual-indicative Snubber design
(Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω). ...............................................................................124
Fig.8-51 The currents in DS1and Don1 in the common and the Dual-indicative Snubber design
(Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω). ...............................................................................124
Fig.8-52 (a) The currents in Df2 and Df2 in the common and the Dual-indicative snubber design and
(b) comparison of the total losses in the conventional and the optimized snubber configuration
(energy function) (Ed=6000V, IL=5000A, CS=0.25µF, RS=2.5Ω). ..................................................125
Fig.9-1 Pulse scheme of Matlab®/SimulinkTM at 900Hz..................................................................126
Fig. 9-2: The schematic diagram of the driving circuit logic...........................................................127
Fig. 9-3: Functional simulation results from the ispDesignExpert. .................................................128
Fig. 9-4: The driving circuit including: (a) The 5 and 15 V power supply from a 12 V source (b)
CPLD Chip and (c) 4 optocoupler ...................................................................................................128
Fig. 9-5 One period of the driving signals .......................................................................................128
Fig. 9-6: The logic block schematic diagram...................................................................................130
Fig. 9-7: Control block of the driving circuit...................................................................................130
Fig. 9-8 Control blocks of the complete Three-level IGBT inverter system. ..................................131
Fig. 9-9 Voltage waveforms across IGBT1, 2....................................................................................132
Fig. 9-10 Current waveforms through IGBT1, 2................................................................................133
Fig. 9-11 Voltage waveforms across DS1, DON1. ..............................................................................133
Fig. 9-12 Load voltage- and current-waveforms. ............................................................................134
Fig. 9-13 Voltage waveforms across IGBT1, 2..................................................................................135
Fig. 9-14 Current waveforms through IGBT1, 2................................................................................135
Fig. 9-15 Voltage waveforms across DS1, DON1. ..............................................................................136
Fig. 9-16 Load voltage- and current-waveforms. ............................................................................136
148
Appendix A3
List of Figures
Appendix A3 List of Tables
Table 1.1 the available diodes information’s. ....................................................................................12
Table 3.1 Switching signals for uni-polar single-phase converter.....................................................49
Table 8.1 the total number of snubber elements for the different snubber designs. . ........................91
Table 8.2 the total number of snubber elements for different snubber designs. ................................92
Table 8.3 the three Switching States..................................................................................................93
Table 8.4 the parameters of the used GTO and the load information................................................99
Table 8.5 Snubber circuits (common and double configuration) information.................................102
Table 8.6 the snubber circuits (common and design) information. .................................................109
Table 8.7 Snubber circuits (common and double dual-use) information.........................................116
Table 9.1 Dual-use- and dual-inductive snubber circuits information.............................................131
149
Appendix A4
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