146
ACTA ELECTROTEHNICA
Design of Series Resonant Parallel
Loaded Transformer less DC Power
Supply
T.T. WAGHMARE, V.B. WAGHMARE and Vitthal BANDAL
Abstract - This paper highlights transformer less, high voltage, series resonant mode DC power supply.
Switching frequency is limited to a lower value due to thermal reasons. As compared to electrical switching
limit, thermal limit is much lower. Switching frequencies in MHz range are possible to reduce size, weight
and converters which involve ZVS and ZCS. A transformer less, high voltage power supply utilizes energy
storage elements and power semiconductors switches. Advent of modern high voltage power
semiconductors and their power capacity operate in radio frequency range makes it possible to eliminate
the transformer in high voltage DC power supply. Circuit used in this paper is modified form of basic series
inverter, which does not require snubber circuit protection. Soft switching is achieved with the help of
saturable reactor. Features of the above circuit are supported by hardware experimental results. Proposed
power supply can be useful in missile applications, battery charging in railways, laser applications and for
DC motors used in coal mines.
Index Terms – Series Resonant, SMPS, Control Circuit, Zero Voltage Switching (ZVS), Zero Current
Switching (ZCS), Resonant Converter, Soft Switching.
1. INTRODUCTION
Power supply is an essential requirement
of all electronic equipments and systems. A
power supply is needed to operate under
varying conditions of load and input supply.
Under all conditions it has to maintain the
output within a close tolerance. In addition, it
has to protect the equipment against line born
disturbances.
Our aim is to design the transfer less high
voltage DC power supply using parallel
loaded, series resonant dc-dc converter which
is snubberless, soft switched assisted and high
voltage dc power supply.
The efforts in power electronics are
always to increase the efficiency, to increase
the power factor, to reduce the harmonics, to
reduce the stress on power devices etc. The
resonant converter fulfills many of these
requirements.
In all switch mode power supplies, switch
closes with voltage across it and opens with
© 2009 – Mediamira Science Publisher. All rights reserved.
current through it. The operating point remain
in active region for considerable period and
energy is converted into heat inside the
device. The average heat developed is directly
proportional to switching frequency. When
switching speed is increased heat developed
does not have enough time to flow towards
heat sink and creates higher thermal gradients.
Thus, it becomes a problem of thermal
transients. Dissipative snubbers shifts
switching power loss from the switch to the
snubber circuits and therefore do not provide a
reduction in the overall switching power loss.
Higher switching frequency reduces
weight, size and hence cost of the power
supply. Also, it provides fast transient
response and less and eases of filtering
requirements. Higher switching frequencies
are possible with resonant converters which
use either zero voltage switching (ZVS) or
zero current switching (ZCS) or both. Lower
transients on sudden application of load or
Volume 50, Number 2, 2009
3. MAIN CIRCUIT DAIGRAM
Fig. 2 shows circuit diagram of parallel
loaded, series resonant dc-dc converter
derived from basic series inverter circuit and
which is the heart of transformer less resonant
mode power supply.
Fuse
G1
A.C.
Supply
G3
S1
_
+
load rejection and availability of possible with
higher switching frequency.
This paper presents design of parallel
loaded, series resonant, snubber less, soft
switching full bridge DC to DC step up
converter. i.e. Transformer less dc power
supply. Lower the value of ξ (damping ratio)
higher is the voltage step up. Zero current
switching (ZCS) is used in the present scheme
which reduces switching losses to lower value
and provision of snubber circuit is not
necessary.
147
C
Ls
C1
C.T.
S3
B
A
G2
G4
S2
S4
To Driver
2. SERIES RESONANT DC-DC
CONVERTER
Circuit
L
Load
+
+
Vd/2
R
L
C
Vd/2
S1
S2
Fig. 1. Basic Series Inverter.
Analyzing above circuit, [1]
− ξπ
V1 1 + e
=
Vd
1− e
1−ξ 2
(1)
−ξπ
1−ξ
2
Where, V1= Initial Voltage across capacitor
Vd= Supply Voltage
ξ = Damping Factor
Also,
R C
(2)
2 L
It means damping factor (ξ) should be as
small as possible to get highest possible
amplification.
ξ=
Vo
C2
_
Series inverter circuit is shown in
figure 1. In above circuit initial voltage across
capacitor goes on increasing as number of
cycles pass. This will be unbounded if circuit
resistance were zero. With resistance in the
circuit, variables stabilize when average
power consumed by resistance equals average
power supplied by source.
_
Fig. 2. Main Circuit Diagram (Parallel loaded, series
resonant DC-DC converter).
Above power circuit uses Power
MOSFET’s. ON&OFF time of Power
MOSFET’s are small as compared to that of
BJT. Power MOSFET is a faster device than
BJT. Also we are using Power MOSFET of
higher rating than of required because it’s
forward conduction drop will be less for the
same value of current as compared to
MOSFET of the required rating. Power
MOSFET’s are chosen because they are work
at higher voltage with less losses.
The dv/dt stress across Power MOSFET
will be low & provision of snubber circuit is
not necessary.
Here the small DC link capacitor C1
(unlike large filters capacitor used in a
conventional rectifier.) is used to filter the
switching frequency components entering the
line and to draw the line current for the most
of the parts 50 Hz cycle. 1,2,3 & 4 are four
MOSFET’s switches. The resonant tank
circuit is formed by the inductors Ls, and
capacitor Cs. For the constant dc output
voltage Vo at the load, a large capacity filter
C2 is used to filter 150 Hz frequency voltage
ripple i.e. transfer from the input section to the
output section. The inductor filter L is used to
filter the HF switching frequency current
ripple in id.
ACTA ELECTROTEHNICA
148
The value of resonant inductor and the
capacitor is chosen according to the decided
resonant frequency. The gauge of the wire
used for winding the inductor is such that it is
capable of carrying a current more than 5 A.
the ferrite core is chosen such that it is capable
of handling 500W power. The 1-26-38 EE
core can handle up to 1500 W power. Also the
window area is sufficient to accommodate the
required number of windings of required
gauge.
At standard value of capacitor is chosen
as the resonant capacitor. The resonating
frequency is about 20 kHz to 40 kHz.
3.1. Control Circuit
Fig. 3 & 4 shows control card or circuit of
voltage controller, over voltage protection &
over current protection.
Voltage
Voltage
Driver
G1
Circuit 1
S1
the separate output voltage isolate from each
other. Turns ratio between primary and
secondary are adjusted so as to obtain +12 V
and -12 V which is given to different IC’s
used in the control card and to the driver card
and all supply are isolated from each other.
This circuit is highly accurate 12 V, 3 V
secondary regulated fly back power supply.
That will operate from 70 V to 130 V input.
High voltage supply is applied to the primary
winding of transformer (t1). The other side of
the transformer is driven by the integrated
high voltage MOSFET transistor within the
TOP 100 (V1). The circuit operates at a
switching frequency near about 25 kHz set by
the internal oscillator.
A TL 431 shunt regulator directly senses
and accurately regulates the output voltage.
The effective voltage can be fine turned by
adjusting the resistor divider formed by R5,
R6 & R7.
VCO
Feedback
Controller
3.3. Driver Circuit
Optoisolator
Over
Voltage
Protection
Trip
Circuit
Optoisolator
Over
Current
Protection
Current
Feedback
Driver
G2
Circuit 2
S2
Fig. 3. Block diagram of Control Circuit.
3 Pin CPU
Fuse
D1
D4
O
+
C1
O
C2
_
_
A
O
+
C6
D2
O
+ 12 V
+
R2
_
Output of the VCO is applied to the
power devices via NOT gates, optocoupler
and the pulse amplifier circuit. TLP 250 is
used for driver circuit. The VCO output is
given to the inverter such that there is a delay
in the firing pulses for the two arms. Thus the
unit avoids the cross conduction of MOSFET.
Power supply given to the two optocoupler
should be isolated from each other.
O Return
NMOS
Main Winding
D3
TOP 223
B
O + 12
C5
C4
+
D5
+
R1
C7
_
C3
_
O Return
D7
L
+ 12 V
Auxilliary Winding
C
O + 12
MCT 2E
R4
R5
C12
D6
R6
+
C8
R3
R7
+
+
C9
_
_
C10
_
O Return
Feedback Winding
C11
TL 431
R8
Fig. 4. Circuit Diagram of Control Circuit.
3.2. SMPS
This power supply generates a voltage of
12 V. it has SMPS IC multiplied transformer
as it’s main components. SMPS IC TOP
SWITCH 101 is responsible for regulating the
output voltage to a desired value. Transformer
has multiplied secondary winding, to generate
3.4. Current Feedback by using C.T.
Current transformer is used to transfer the
current from primary to secondary side to
measure high circulating current in the
resonant circuit we use current transformer.
Current transformer’s primary is connected in
series with the resonant circuit and it;
secondary sides are never left open. The turn
ratio of the transformer is 1:400, it reduces the
secondary current by a factor of 400. This is
then converted to the corresponding voltage
by connecting a resistor of suitable value as a
load. The voltage across this load is given to
the over current protection circuit of the
control card circuit.
Volume 50, Number 2, 2009
3.5. Voltage Feedback
The part of the output voltage is sampled
and given the voltage controller and over
voltage protection card. There negative
feedback is used so as to operate the circuit
above the resonant frequency.
149
= 0.0003125 Watt-Sec.
Calculate the electrical conditions ‘Ke’
Ke= 0.145 x (Po) x (Bm)2 x 10-6
= 0.145 x (500) x (0.3)2 x 10-6
= 6.525 x 10-6
Calculate the core geometry ‘Kg’
3.6. Rectifier and Filter
High frequency sine wave is applied to
the bridge rectifier. The diodes of bridge
rectifier must be fast recovering diodes. The
rectified voltage is then filtered by LC type
filter. The inductor is wound on a ferrite since
the ripple frequency is very high. A core is
selected such that it can handle 500 W power.
This output is given to lamp load.
3.7. Design for a Gapped E-Core
Inductor
1. Inductance L= 0.4 mH = 0.4*10 –3 [H]
2. DC current I0 = 0
[A]
3. AC current ∆I = 2.5
[A]
4. Output power Po = 500
[W]
5. Flux density, Bm = 0.3
[T]
6. Ripple frequency, F = 25
[KHz]
7. Regulation, x = 1%
8. Core material: Silicon
9. Core configuration: E Core
H
Since, L = 0.4 x 10 –3
Resonant frequency,
1
f =
2π LC
C = 0.1 μF
Calculate the energy handling capability,
Energy = LI
Watt-Sec.
2
2
I=
I0 +
ΔI
2
Amp
2.5
2
= 1.25 Amp
= 0+
Energy =
LI 2
Watt-Sec.
2
[0.4 × 10 −3 ].[1.25] 2
2
= 3.125 x 10-4
=
( Energy ) 2
Kg = Ke × α
[0.0003125] 2
−6
= [6.525 × 10 ] × 1
= 0.0149664 cm5
Select a comparable core geometry
coefficient Kg from the specified core section
and record the appropriate data
AWG = 14
Kg = 0.044456 cm5
Ap = 0.76 cm4
MLT = 6.2 cm
Ac = 0.91 cm2
Wa = 0.84 cm2
At = 33.8 cm2
G = 1.321 cm
MPL = 5.8 cm
Wtfe = 40 g
Calculate the current density J. Use the
area product Ap found,
2(Energy) × 10 4
Bm × A p × Ku
J=
A/cm2
Window utilization factor Ku = 0.4
2(0.0003125) x10 4
J = (0.3)(0.84)(0.4)
= 68.53 A/cm2
Calculate the bare wire size Aw(B),
ΔI
I0 +
2
J
cm2
Aw(B) =
2.5
0+
2
Aw(B) = 68.53
ACTA ELECTROTEHNICA
150
Aw(B) = 0.01824 cm2
180
Calculate the effective window area
Wa(eff)
Use window area Wa found in above step
Wa(eff) = Wa. S3 cm2
A typical value for S3 is 0.75
∴ Wa(eff) = 0.84 x 0.75
= 0.63 cm2
Calculate the number of turns. Use wire
area Aw found in above step.
Wa ( eff ) × S 2
[turns]
N=
Aw
A typical value for S2 is 0.6
(0.63)(0.6)
= 16 turns
N=
0.02295
Observation Table for Load Regulation:
Input A.C.
Output DC
Output DC
Sr.
No. Voltage (Volts) Voltage (Volts) current (Amps)
160 V
160 V
160 V
160 V
45 V
56 V
70 V
87 V
0.3 A
0.7 A
1.1 A
1.6 A
Observation Table for Efficiency:
Sr.
No.
1
2
3
4
Output DC
Input A.C.
Efficiency
Power (Watts) Power (Watts)
η%
45 V
56 V
70 V
87 V
160 V
160 V
160 V
160 V
Output DC Voltage
140
120
100
Load Regulation
80
60
40
20
0
0
0.2 0.4 0.6 0.8
1
1.2 1.4 1.6 1.8
Output DC Current
Output DC Current Vs Output DC Voltage
90
80
70
60
50
%Efficiency
40
30
20
10
0
0
50
100
150
200
250
300
Output Power
Output Power Vs Efficiency:
5. CONCLUSION
4. EXPERIMENTAL RESULTS
1
2
3
4
160
%Efficiency
Select a wire size from table, if the area is
not within 10% take the next smallest size.
Also record micro-ohms per centimeter from
column 4 and wire area with insulation Aw
from column 5.
AWG No. 14
Bare, Aw(B) = 0.02082 cm2
Insulated, Aw = 0.02295 cm2
μΩ/cm = 82.80
0.3 A
0.7 A
1.2 A
1.6 A
DC power converters have been described
which are capable of operating in the
hundreds of kilohertz range and which employ
a resonant circuit. The advantages of the series
resonant, parallel loaded DC-DC converters
which are verified by a developed setup as
follows• Low power MOSFET stresses due to
lossless snubber operation.
• Switching frequencies in the hundreds of
kilohertz range to reduce size, weight and
cost of the power supply.
• Switching losses i.e. turn-on & turn-off
losses; are reduced by zero voltage and
zero current switching.
• Ability to step the input voltage either up
or down.
REFERENCES
1. Robert L. Steigerwald, “High-frequency Resonant
Transistor DC-DC converters”, IEEE Tran. On
Industrial Electronics. Vol. IE-31, No.2, May 1984.
Volume 50, Number 2, 2009
2. Pravin K. Jain, Wen Kang, Harry Soin, and Youhao
Xi, “Analysis and Design Consideration of a Load
and Line Independent Zero Voltage Switching Full
Bridge DC/DC converter Topology”, IEEE Tran.
On Power Electronics. Vol.17, No.5 September
2002.
3. Robert L Steigerwald, ”A Comparision of Halfbridge Resonant Converter Topologies”, IEEE
Tran. On Power Electronics. Vol.3, No.2, April
1988.
4. Ashoka K.S. Bhat, “Analysis and Design of a
Series-Parallel Resonant Converter with Capacitive
Output Filter”, IEEE Tran. On Industrial
Applications. Vol.27, No.3, May/June 1991.
5. N. Mohan, J. Undeland, W. Rabbins, “Power
Electronics Converters, Applications and Design”,
2nd Edition, John Wiley and Sons, Singapur 1995.
151
Assist.Prof. T.T. Waghmare, Member IEEE
Department of Electrical Engineering
Modern College of Engineering
Pune, Maharashtra, INDIA
E-mail: wtushar123@gmail.com
Prof. V.B. Waghmare, Member IEEE
Department of Electrical Engineering
Government College of Engineering
Amravati, Maharashtra INDIA
E-mail: vbwaghmare@rediffmail.com
Assist.Prof. Vitthal Bandal, Member IEEE
Department of Electrical Engineering
Government College of Engineering
Pune, Maharashtra, INDIA
E-mail: wtushar123@yahoo.co.in