DISS. ETH No. 18472
A Low-Noise-Figure 35-GHz Receiver with
Beamforming Capability based on
Injection-Locked Local Oscillators
A dissertation submitted to the
SWISS FEDERAL INSTITUTE OF TECHNOLOGY ZURICH
for the degree of
Doctor of Sciences
presented by
HANNES GRUBINGER
Dipl.-Ing., Technische Universität München (TUM), Germany
born September 2, 1978
citizen of Austria
accepted on the recommendation of
Prof. Dr. Christian Hafner, examiner
Prof.em. Dr. Werner Bächtold, co-examiner
Prof. Dr. Stefan Heinen, co-examiner
Prof. Dr. Rüdiger Vahldieck, co-examiner
2009
DISS. ETH No. 18472
A Low-Noise-Figure 35-GHz Receiver with
Beamforming Capability based on
Injection-Locked Local Oscillators
A dissertation submitted to the
SWISS FEDERAL INSTITUTE OF TECHNOLOGY ZURICH
for the degree of
Doctor of Sciences
presented by
HANNES GRUBINGER
Dipl.-Ing., Technische Universität München (TUM), Germany
born September 2, 1978
citizen of Austria
accepted on the recommendation of
Prof. Dr. Christian Hafner, examiner
Prof.em. Dr. Werner Bächtold, co-examiner
Prof. Dr. Stefan Heinen, co-examiner
Prof. Dr. Rüdiger Vahldieck, co-examiner
2009
Acknowledgment
The work behind this dissertation was carried out at the Laboratory for Electromagnetic
Fields and Microwave Electronics (IFH), Swiss Federal Institute of Technology, Zurich,
Switzerland. The research project was supported by armasuisse.
First of all, I would like to thank Prof. Rüdiger Vahldieck, who granted me the opportunity to carry out my Ph.D. thesis in his research group. He arranged the project
financing and offered resources, laboratory environment and the support necessary for
my work. Despite his busy schedule, he was accessible for discussions about my thesis
and for other research, teaching and institute-related topics throughout. He was always
willing to contribute his experience and ideas to my research. Implementing and testing
these ideas often improved the results significantly. He encouraged me to publish the
results, helped improving the manuscripts and provided the necessary financial support
for publishing journal papers and traveling to conferences.
I want to thank my supervisor Dr. Helmut Barth for his guidance during my Ph.D.
thesis. It was he who suggested the construction of a receiver and the use of an injectionlocked VCO as LO. His experience and enormous knowledge were a great support in
getting me started with this research project. He was always willing to share his ideas
with me and I greatly appreciate his contribution in many long discussions. The results
of which were helpful and contributed to the results presented in this dissertation.
I owe sincere thanks to my co-examiners Prof.em. Dr. Werner Bächtold from ETH
Zurich and Prof. Dr. Stefan Heinen from RWTH Aachen, Germany for the thorough
reviews of the dissertation, their very constructive comments, and their personal commitment. Furthermore, I want to thank Prof. Dr. Christian Hafner for agreeing to be my
examiner. His constructive comments were very helpful for improving this manuscript.
I would like to thank Dr. Jan Hesselbarth for reading my dissertation, his helpful
comments enabled me to make a number of improvements to the manuscript. A special
thank you to Dr. Martin Gimersky for helping me refine many of my papers. He gave me
valuable answers to all my various engineering and language-related questions. I really
appreciated that, however close a deadline was, Martin was always willing to spend his
time assisting me with my work.
A big “Merci vielmoooool” goes to Hansruedi Benedickter for instructing me in the
use of many measurement devices and methods, and aiding me in their use. Frequently,
the day became too short yet he stayed with me in the laboratory until late at night in
order to complete a measurement. Furthermore, I want to thank Thomas Kleier from
the Integrated Systems Laboratory for instructing me in the use of the phase-noisemeasurement equipment.
I want to thank Aldo Rossi for supporting me with electronic components and helping
setting-up the clean-up shunt. He complied with every request I had to the computer infrastructure. Many thanks go also to Martin Lanz, Claudio Maccio and Stephen Wheeler
ii
ACKNOWLEDGMENTS
for fabricating the hardware. I appreciate that all my designs were produced although
many of them required manufacturing with very low tolerances.
Finally, I want to thank my parents and my my girlfriend Nadine Künzler for their
patience and support. Furthermore, I want to express my gratitude to all the people
around me for long discussions and motivating talks, but also for long Nelson nights and
aggressive ski sessions. Last, but not least, I want to express my grateful thanks to the
Alps, which always gave me the opportunity to reinvigorate myself.
Hannes Grubinger
Zurich, March 18, 2009
Auch aus Steinen,
die einem in den Weg gelegt werden,
kann man Schönes bauen.
J.W. Goethe
Abstract
Radiometers are traditionally built in waveguide technology to keep their noise figure
low. The disadvantages of this technology are the large size and the bulky handling. In
contrast, a compact receiver for radiometry is discussed in this dissertation to overcome
these disadvantages. The noise figure is minimized by using an LNA and an image
rejection filter. The mixer is driven by the frequency doubled output signal of an injection
locked VCO. The injection locked VCO is operated as phase shifter.
Control of the injection-locked VCO phase is achieved by changing the VCO freerunning frequency. Applying high locking power leads to a phase shift of more than
180◦ . Furthermore, it has been shown that the phase can be changed over the whole
tuning range only if the locking range is smaller than the free-running frequency tuning
range. Therefore, a VCO with low Q-factor is used in all the models and prototypes
shown to achieve good controllability and a wide phase-tuning range.
The second advantageous attribute of injection-locked oscillators is their phase noise.
Although the low-Q VCOs exhibit a bad phase-noise behavior, the phase noise when
injection-locked is mainly dependent on the reference oscillator. Only at higher offset
frequencies does the phase noise of the injection locked VCO exceed the reference oscillator phase noise. For this reason, a reference oscillator, based on a 15-GHz LNA fed
back by a high-Q cavity resonator, has been developed.
Two different phase shifters are tested. The first phase shifter type is based on an
unbuffered chip-VCO connected to a T-junction and input and output amplifiers. As
the VCO is based on a bipolar transistor and has two bias voltages – a collector and a
base voltage – the VCO operating points can be set such that the free-running frequency
tuning range is symmetrical to the reference oscillator signal. The reference oscillator
signal is used as input signal. A phase-tuning range of ∆φ > 200◦ and excellent phase
noise of the phase shifter output signal have been measured.
In addition to the chip-VCO phase shifter, an alternative phase shifter type is demonstrated. The architecture of this phase shifter is similar to the reference oscillator, but
a tunable low-Q resonator is used in the feed-back path instead of the high-Q cavity.
Input and output amplifiers are directly connected to the oscillator loop. As the phasetuning and phase-noise behavior of this phase shifter type is similar to the behavior of
the chip-VCO phase shifter, it represents a cheap alternative when low quantities are
required. However, this phase shifter type is not suitable for larger quantities, as the
production is very complex. An additional disadvantage of this phase-shifter type is the
larger size in comparison to the chip-VCO phase shifter.
The chip-VCO phase shifter is used in a receiver implementation. The receivers are
designed such that they can be replicated and be used in an array. In order to test
the noise and phase behavior of a receiver array, two parallel receivers have been implemented. The small receiver-to-receiver distance and the employed small connectors
v
vi
ABSTRACT
simplify the connectivity to an antenna array. A noise figure of only 2.5 dB and a phase
tuning range of ∆φ > 400◦ are measured with this experimental set-up.
Zusammenfassung
Um die Rauschzahl gering zu halten werden Radiometer typischerweise in Hohlleitertechnologie gefertigt. Die Nachteile dieser Technologie sind die Grösse und die sperrige Handhabung. Im Kontrast zu hohlleiterbasierten Radiometer, wird in dieser Dissertation eine kompakte Realisierung eines Empfängers für Radiometrie diskutiert. Die
Empfängerrauschzahl wird durch die Verwendung eines LNAs am Eingang und eines
Eingangsfilters reduziert. Der Mischer wird durch das frequenzverdoppelte Ausgangssignal eines injektionssynchronisiert VCOs gespeist. Der synchronisierte VCO wird dabei
als Phasenschieber verwendet.
Die Steuerbarkeit der VCO Phase wird durch die Veränderung der Freilauffrequenz
erreicht. Es wird gezeigt dass ein Steuerbereich grösser als 180◦ erreicht werden kann
indem man die Leistung des Referenzoszillators erhöht. Eine Steuerung der Phase über
den gesamten Phasensteuerungsbereich ist nur dann möglich, wenn der synchronisierte
Frequenzbereich des Oszillators kleiner als der Frequenzsteuerbereich ist. Letzteres wird
durch die Verwendung eines VCOs mit geringer Güte erreicht.
Der zweite Vorteil von injektionssynchronisierte Oszillatoren ist deren tiefes Phasenrauschen. Obwohl der verwendete, freilaufende VCO ein sehr hohes Phasenrauschen aufweist, wird im synchronisierten Zustand das Phasenrauschen vor allem durch den Referenzoszillator bestimmt. Aus diesem Grund wurde ein Referenzoszillator mit geringem
Phasenrauschen entwickelt. Dieser Oszillator beruht auf einem durch eine hochgütige
Kavität rückgekoppelten 15-GHz Verstärker.
Auf synchronisierten Oszillatoren basierend wurden zwei verschiedene Phasenschieber
entwickelt. Der erste besteht aus einem Chip-VCO verbunden mit einer T-Verzweigung,
einem Eingangs- und einen Ausgangsverstärker. Der VCO hat zwei Versorgungsspannungen: Eine Kollektor- und eine Basispannung. Dies ist in der gezeigten Anwendung
ein Vorteil, da der Frequenzbereich des VCOs mit Hilfe dieser beiden Spannungen so
eingestellt werden kann, dass er symmetrisch zum Referenzoszillatorsignal ist. Ein grosser Phasenbereich von ∆φ > 200◦ und exzellentes Phasenrauschen wurden mit Hilfe des
Prototypen gemessen.
Als Alternative zum vergängig beschriebener Phasenschieber wurde ein zweiter Phasenschieber entwickelt. Das Konzept dieses Phasenschiebers ist ähnlich wie das des Referenzoszillators. Der Unterschied besteht im Resonator: Der Phasenschieber verwenden
einen steuerbaren Resonator niederer Güte anstatt der hochgütigen Kavität im Referenzoszillator. Ferner besitzt der Phasenschieber auch einen Eingang mit Verstärker, welcher
Injektionssynchronisierung ermöglicht. Wenn kleine Stückzahlen benötigt werden, bietet
dieser Phasenschiebertyp eine gute Alternative zum Chip-VCO Phasenschieber, da die
gemessene Phasenbereich und das Phasenrauschen der beiden Phasenschieber ähnlich
sind. Bei grösseren Stückzahlen ist der zweite Typ allerdings aufgrund der aufwändigen
Produktion nachteilhaft. Ferner ist der zweite Phasenschieber wesentlich grösser als der
erste.
vii
viii
ZUSAMMENFASSUNG
Der Phasenschieber mit dem Chip-VCO wurde anschliessend im Empfängerprototyp
verwendet. Der Empfänger ist so entwickelt, dass er vervielfacht und in einer Zeile verwendet werden kann. Die kleine Empfängerbreite und die verwendeten schmalen Stecker
erleichtern die Verbindung mit einer Antennenzeile. Um die Rauschzahl und die Phasensteuerung der Empfänger zu testen, wurden zwei nebeneinander liegende Empfänger
produziert. Die Messung ergibt eine Rauschzahl von nur 2.5 dB und einen steuerbaren
Phasenbereich von ∆φ > 400◦ .
Contents
Acknowledgments
i
Abstract
v
Zusammenfassung
vii
Acronyms and Abbreviations
xix
Symbols
xxi
1 Introduction
1.1 Motivation . . . . . . . . . . . . .
1.2 Electronic Beamforming Networks
1.3 Low-Noise Receiver . . . . . . . . .
1.4 Array of Receivers . . . . . . . . .
1.5 Objective . . . . . . . . . . . . . .
1.6 Outline . . . . . . . . . . . . . . .
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1
1
1
2
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2 Injection-Locked VCO
2.1 Principle of Injection Locking . . . . . . . . .
2.2 Free-Running Frequency-to-Phase Relation .
2.3 Output Signal of Injection-Locked Oscillator .
2.4 Comparison of VCO-Tree and VCO-Chain . .
2.5 Conclusion . . . . . . . . . . . . . . . . . . .
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9
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and Frequency Fine-Adjustments
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21
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33
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3 Reference Oscillator
3.1 Introduction . . . . .
3.2 Cavity Design . . . .
3.3 Oscillator . . . . . .
3.4 Measurement Results
3.5 Conclusion . . . . .
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4 Phase Shifter
4.1 VCO Requirements . . . . . . . . . . . . . . .
4.2 Chip-VCO based Phase Shifters . . . . . . . .
4.2.1 VCOs . . . . . . . . . . . . . . . . . .
4.2.2 VCO Load . . . . . . . . . . . . . . .
4.2.3 Phase Shifter Circuits . . . . . . . . .
4.2.4 Free-Running Frequency and Pushing
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ix
x
CONTENTS
4.3
4.4
4.2.5 Maximized Locking Range and Phase-Tuning Behavior .
4.2.6 Phase Noise . . . . . . . . . . . . . . . . . . . . . . . . .
Feed-Back Loop based VCO used as Phase Shifter . . . . . . .
4.3.1 VCO Architecture . . . . . . . . . . . . . . . . . . . . .
4.3.2 VCO Resonator . . . . . . . . . . . . . . . . . . . . . . .
4.3.3 VCO Circuit and Free-Running Frequency . . . . . . . .
4.3.4 Phase-Tuning Behavior . . . . . . . . . . . . . . . . . .
4.3.5 Phase Noise . . . . . . . . . . . . . . . . . . . . . . . . .
Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
5 Receiver
5.1 Receiver Architecture . . . . . . . . . . . . . . . . .
5.2 Low-Noise Down-Converter . . . . . . . . . . . . . .
5.2.1 Input LNA and Lower Sideband Suppression
5.2.2 Filter Concepts . . . . . . . . . . . . . . . . .
5.3 Filter Simulation and Optimization Method . . . . .
5.4 Broadband Microstrip Filter . . . . . . . . . . . . . .
5.4.1 Filter Topology . . . . . . . . . . . . . . . . .
5.4.2 Simulation and Optimization . . . . . . . . .
5.4.3 Realization and Measurement . . . . . . . . .
5.5 Narrowband SIW Filter in LTCC . . . . . . . . . . .
5.5.1 Filter Topology . . . . . . . . . . . . . . . . .
5.5.2 Simulation and Optimization . . . . . . . . .
5.5.3 Prototype and Measurements . . . . . . . . .
5.6 Noise Figure . . . . . . . . . . . . . . . . . . . . . .
5.6.1 Testing Device . . . . . . . . . . . . . . . . .
5.6.2 Measurements . . . . . . . . . . . . . . . . .
5.7 Prototype of Receiver with Beamforming Capability
5.7.1 System Layout . . . . . . . . . . . . . . . . .
5.7.2 Receiver Tests . . . . . . . . . . . . . . . . .
5.7.3 Measured Phase Shift . . . . . . . . . . . . .
5.7.4 Noise Figure Measurement . . . . . . . . . .
5.8 Conclusion . . . . . . . . . . . . . . . . . . . . . . .
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59
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6 Conclusion
89
6.1 Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89
6.2 Future Work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 90
A Photographs of Measurement Set-Ups
A.1 Introduction . . . . . . . . . . . . .
A.2 Wiltron Universal Test Fixture . .
A.3 Phase Shifter . . . . . . . . . . . .
A.4 Receiver . . . . . . . . . . . . . . .
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91
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94
CONTENTS
xi
B LTTC Receiver Design
97
B.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 97
B.2 Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 97
B.3 Complexity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102
Bibliography
103
List of Publications
111
Curriculum Vitae
113
xii
List of Figures
1.1
1.2
1.3
Schematic diagram of receiver in proposed receiver-array. . . . . . . . . .
Schematic diagram of overall system. . . . . . . . . . . . . . . . . . . . . .
Simulated normalized array factor of a 8 × 8-Butler matrix. . . . . . . . .
2
4
5
2.1
2.2
2.3
2.4
2.5
2.6
Equivalent circuit of VCO with additional admittance. . . . . . . . . . .
Derived phase shift of injection-locked VCO. . . . . . . . . . . . . . . . . .
Derived single-sideband phase noise of injection-locked VCO . . . . . . . .
Schematic diagram of VCO-chain. . . . . . . . . . . . . . . . . . . . . . .
Derived single-sideband phase noise of injection-locked VCOs in VCO-chain.
Schematic diagram of VCO-tree. . . . . . . . . . . . . . . . . . . . . . . .
11
13
15
16
17
18
3.1
3.2
3.3
3.4
3.5
3.6
3.7
3.8
3.9
3.10
Schematic of reference oscillator. . . . . . . . . . . . . . . . . . . . . . . .
Cross section and photograph of cavity resonator. . . . . . . . . . . . . . .
Simulated cavity insertion loss. . . . . . . . . . . . . . . . . . . . . . . . .
Measured insertion loss of resonator. . . . . . . . . . . . . . . . . . . . . .
Photograph of coupling post. . . . . . . . . . . . . . . . . . . . . . . . . .
Photograph of manufactured oscillator. . . . . . . . . . . . . . . . . . . . .
Measured spectrum of oscillator. . . . . . . . . . . . . . . . . . . . . . . .
Transmission parameter S21 of frequency-adjusted cavity. . . . . . . . . .
Photograph of resonator cavity. . . . . . . . . . . . . . . . . . . . . . . . .
Measured single-sideband phase noise power density of reference oscillator.
21
24
25
26
26
27
28
28
29
29
4.1
4.2
4.3
4.4
4.5
4.6
4.7
4.8
4.9
4.10
4.11
4.12
4.13
4.14
4.15
4.16
Schematic diagram of Phase shifters based on injection-locked VCOs. .
Schematic diagram of used VCOs. . . . . . . . . . . . . . . . . . . . .
Measured frequency-tuning behavior of VCO with 6HP BJT. . . . . .
Schematic of possible phase shifter architectures. . . . . . . . . . . . .
Simulated return loss of T-junction and coupler. . . . . . . . . . . . .
Dimensions of tested locking networks. . . . . . . . . . . . . . . . . . .
Photograph of tested locking networks. . . . . . . . . . . . . . . . . . .
Insertion loss and return loss of measured coupler. . . . . . . . . . . .
Measured free-running frequency of VCO with T-junction and coupler.
Influence of bias voltages to VCO frequency. . . . . . . . . . . . . . . .
VCO tuning and locking range. . . . . . . . . . . . . . . . . . . . . .
Measurement setup for phase-tuning measurements. . . . . . . . . . .
Oscilloscope plot of detector and ramp signal. . . . . . . . . . . . . . .
Measured phase-tuning curve of phase shifter. . . . . . . . . . . . . . .
Phase noise measurement set-up. . . . . . . . . . . . . . . . . . . . .
Measured phase noise of chip-based VCO. . . . . . . . . . . . . . . . .
33
35
36
37
38
39
40
41
41
42
42
43
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xiii
xiv
LIST OF FIGURES
4.17
4.18
4.19
4.20
4.21
4.22
4.23
Feed-back loop based VCO architecture. . . . . . . .
Photographs of tested resonators. . . . . . . . . . . .
Resonator insertion loss. . . . . . . . . . . . . . . . .
Photograph of feed-back loop based VCO. . . . . .
Measured frequency tuning range. . . . . . . . . . .
Measured phase-tuning range. . . . . . . . . . . . . .
Measured phase noise of feed-back loop based VCO.
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50
53
54
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56
5.1
5.2
5.3
5.4
5.5
5.6
5.7
5.8
5.9
5.10
5.11
5.12
5.13
5.14
5.15
5.16
5.17
5.18
5.19
5.20
5.21
5.22
Block diagram of receiver input. . . . . . . . . . . . . . . . . . . . . . .
Schematics of receivers with broadband and narrowband filters. . . . . .
Cross section of Shunt-inductance-coupled waveguide filter. . . . . . . .
Cross section of microstrip filter and amplifiers. . . . . . . . . . . . . .
Sketch of geometries used for optimizing microstrip-based filter . . . . .
Geometry and photograph of 35-GHz microstrip-based filter. . . . . . .
Simulated and measured insertion loss and return loss. . . . . . . . . . .
Cross section of LTCC module with SIW filter. . . . . . . . . . . . . . .
Cross section and 3D-drawing of SIW filter. . . . . . . . . . . . . . . . .
Geometries of simulation models used for optimizing SIW-based filter. .
Simulated filter insertion and return loss. . . . . . . . . . . . . . . . . .
Photograph (top-view) of manufactured SIW filter. . . . . . . . . . . .
Measured and simulated filter insertion and return loss. . . . . . . . . .
Photograph of test structure enclosure. . . . . . . . . . . . . . . . . . .
Photograph of receiver input. . . . . . . . . . . . . . . . . . . . . . . . .
Measured noise figure of receiver. . . . . . . . . . . . . . . . . . . . . .
Schematic of realized receivers. . . . . . . . . . . . . . . . . . . . . . . .
Photograph of implementation before placing active components. . . . .
Photograph of two-receiver module. . . . . . . . . . . . . . . . . . . . .
Photograph (side-view) of open and closed two-receiver module. . . . . .
Measured output frequency of receiver with not injection locked VCO. .
Schematic of measurement set-up used for measuring phase tuning behavior of receiver. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Oscilloscope plot of detector and ramp signals. . . . . . . . . . . . . . .
Receiver phase shift vs. VCO tuning voltage. . . . . . . . . . . . . . . .
Measured receiver noise figure. . . . . . . . . . . . . . . . . . . . . . . .
Photograph of antenna array enabled for connecting receiver. . . . . . .
Measured receiver gain. . . . . . . . . . . . . . . . . . . . . . . . . . . .
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85
5.23
5.24
5.25
5.26
5.27
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A.1 Photograph of Wiltron universal test fixture. . . . . . . . . . . . . . . .
A.2 Photograph of measurement showing batteries used for biasing. . . . . .
A.3 Photograph of measurement set-up for measuring phase tuning behavior
of phase-shifter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
A.4 Photograph of receiver measurement set-up. . . . . . . . . . . . . . . . .
A.5 Photograph of receiver in measurement fixture. . . . . . . . . . . . . . .
. 91
. 92
. 93
. 94
. 95
B.1 Cross-section of LTCC module. . . . . . . . . . . . . . . . . . . . . . . . . 98
LIST OF FIGURES
xv
B.2 Layer and microstrip-to-stripline transitions. . . . . . . . . . . . . . . . . . 99
B.3 Top-view of components and transmission lines in LTCC module. . . . . . 101
xvi
List of Tables
1.1
Frequencies of receiver components.
3.1
3.2
3.3
3.4
Derived resonator resonance frequencies and Q-factors. .
Calculated Q-factor when varying length coupling post.
Measured parameters of reference oscillator. . . . . . .
Used components in reference oscillator. . . . . . . . . .
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23
25
30
30
4.1
4.2
4.3
4.4
4.5
4.6
4.7
Measured phase-tuning range with T-junction and coupler.
Bias-voltage-to-phase sensitivity.. . . . . . . . . . . . . . . .
Measured parameters of chip VCO-based phase shifter. . .
Components employed in chip-VCO based phase shifter. . .
Comparison of tested microstrip-based resonators. . . . . .
Measured parameters of feed-back loop based phase shifter.
Components used in feed-back loop based phase shifter. . .
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45
46
48
49
51
57
58
5.1
5.2
Gain, loss and noise figure of LNA, mixer and filter. . . . . . . . . . . .
Derived Chebyshev parameters and resonator insertion losses for 35-GHz
microstrip filter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Derived Chebyshev parameters and resonator insertion losses for 35-GHz
SIW filter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Dimensions of manufactured filter. . . . . . . . . . . . . . . . . . . . . .
Comparison between calculated and measured noise figure. . . . . . . .
Components used in Receiver. . . . . . . . . . . . . . . . . . . . . . . .
Measured receiver parameters. . . . . . . . . . . . . . . . . . . . . . . .
5.3
5.4
5.5
5.6
5.7
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69
70
75
86
87
xvii
xviii
List of Acronyms and Abbreviations
Signals and Measurement Values
ENR . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Excessive noise ratio
IF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Intermediate frequency
LO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Local oscillator
RF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Radio frequency
AC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Alternating current
DC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Direct current
NF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Noise figure
Q-factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Quality factor
Technology, Packaging and Components
BJT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Bipolar junction transistor
LNA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Low-noise amplifier
LTCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Low-temperature co-fired ceramic
MEMS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Micro-electro-mechanical systems
MMIC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Monolithic microwave integrated circuit
PLL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Phase-locked loop
SIW . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Substrate-integrated waveguide
SMA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Sub-miniature version A (connector)
VCO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Voltage-controlled oscillator
VNA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Vector network analyzer
xix
xx
List of Symbols
f
frequency
f0
reference oscillator frequency
ffr
VCO free-running frequency
fl,m,n
resonance frequency of a resonator TE/TMl,m,n mode
fl , fu
lower and upper locking limit
fmin , fmax
VCO tuning limits
v
free-running / reference frequency ratio
B
bandwidth
ε0
dielectric permittivity constant
µ0
magnetic permeability constant
εr
relative dielectric permittivity
µr
relative magnetic permeability
F
noise figure
Famp
noise figure of amplifier
G
gain
Gamp
amplifier gain
Gloop
gain of oscillator loop
L
loss
LTL
loss of a transmission line
Lmx
conversion loss of a mixer
xxi
xxii
MATHEMATICAL NOTATION
Lfilter
insertion loss of a filter
S
signal power
N
noise power
L
single-sideband phase noise
λl,m,n
resonance free-space wavelength of a resonator TE/TMl,m,n -mode
Pin , Pout
input / output power of device
PVCO,in , PVCO,out input / output power of VCO
p
locking gain
φ0
constant phase
φloop
phase of oscillator loop
φvariable
phase of variable phase shifter
∆φ
controllable phase shift
∆φmax
maximal phase shift
Q
unloaded Q-factor of resonator
Qext
external Q-factor of oscillator
S
scattering matrix
Sm,n
element (m, n) in scattering matrix
Vbase
base voltage of VCO
Vdd
bias voltage of device
Vdetector
output voltage of detector
V in , V out
input / output voltage of component, device
MATHEMATICAL NOTATION
Vtune
tuning voltage of VCO
V VCO,in , V VCO,out input / output voltage of VCO
Z0
reference impedance, usually 50 Ω
Zw
characteristic transmission-line impedance
xxiii
xxiv
1 Introduction
1.1 Motivation
Traditionally radiometers are built in waveguide technology (e.g. [1, 2]) in conjunction
with horn antennas (e.g. [3]). The reason for choosing this bulky and expensive technology is the achieved low loss. The disadvantage of this technology is that beam steering
is only possible by mechanically moving the antenna. Both, the bulky structure and the
mechanical beam steering system lead to high space requirements.
In contrast, present-day requirements call for light and highly-compact systems. Small
low-loss receivers can be used in new application fields, such as security screening in
airports or at check points. Furthermore, small size implementations might be suitable
for mobile use. The use of electronic beamforming instead of mechanically moved parts
not only reduces the size of the receiver, but also simplifies the receiver configuration.
Planar packaging technologies reduce the size of the receiver further.
1.2 Electronic Beamforming Networks
Literature shows various possibilities to form beams of an antenna array. One of the
oldest techniques is employing switches for changing the phase relation between the
respective antennas in an array. An example of diode phase shifters is shown in [4].
Building switches by using Micro-Electro-Mechanical Systems (MEMS) is a promising
method (e.g. [5, 6]). However, switches allow only switching between certain phase
relations. Thus, steering the beam continuously in space is not possible.
Another common method for beamforming is employing phase shifters based on phaselocked loops (PLLs). Applications are shown in [7, 8]. PLLs require a certain settling
time. Thus, the scanning speed is limited in PLL-based beamforming networks.
An alternative way of changing the phase between antenna elements is to use a resonant
structure whose properties are changed by varactor diodes. An example, where the
properties of a reflector are changed is proposed in [9]. However, the use of varactor
diodes in the radiating part of an antenna increases losses. These additional losses make
this method unattractive for low-noise applications.
Injection-locking a local oscillator by a reference oscillator leads to a phase difference
between the two oscillator signals [10]. Therefore, also this method can be used for beamforming. The phase difference is defined by the oscillator properties and their operating
points. Voltage-controlled oscillators (VCOs) allow controlling this phase difference (Section 2.2). The phase can be changed continuously between the locking limits by simply
changing the DC-voltage. Furthermore, an additional advantage of this phase-tuning
method is that the phase change takes effect in a very short settling time.
Normally, the locked VCO has a low Q-factor, whereas a reference oscillator with high
Q-factor is used for locking. Since the VCO is operated only in its locked state, in which
1
2
1 INTRODUCTION
Figure 1.1: Schematic diagram of one receiver in proposed receiver-array (dashed rectangular).
Reference oscillator is used as reference for all receivers in the array.
it adopts the noise properties of the locking signal, the output signals of the injectionlocked VCO and the reference oscillator are similar (Section 2.3). This shows that this
method is suited for low-noise applications. Placing VCOs in an array allows controlling
the phase of every antenna in an antenna array independently (Section 2.4). The VCO
array can be used as beamforming network.
1.3 Low-Noise Receiver
Most systems reported in literature use the transmitted signal as reference signal for
injection locking the VCO [13, 14]. Only signal transmission, but not reception, is
possible with such a system architecture. In contrast to those concepts, the system
introduced in this dissertation uses a high-Q single-tone reference oscillator for injection
locking VCOs. Figure 1.1 shows the schematic of one receiver. The reference oscillator
signal is used for injection locking the VCOs in every receiver. Attenuators are used
for adjusting the power level in order to get an optimal locking range (the power level,
marked with * in Figure 1.1 depends on the VCO type). The phase-controlled output
signal of the VCO is frequency doubled and used for driving a mixer. Since the mixer
can be employed for down- and up-conversion, the concept can generally be used for
receivers and transmitters.
1.4 ARRAY OF RECEIVERS
3
In the following, a receiver with an input frequency of 35 GHz is discussed. The
frequency has been chosen, since the receiver is intended to be used as radiometer, and
35 GHz is the first transmission window of the earth’s atmosphere [15]. The output
frequency is set to 5 GHz to allow combining a receiver array with a Butler matrix.
The implementing of the reference oscillator and phase shifters at the lower frequency
of 15 GHz instead of 30 GHz and frequency doubling the signal has the advantage of
also doubling the phase tuning range [16]. Additionally, the component availability and
the relative tolerances in the manufacturing processes are better. This reduces also the
costs.
The receiver should be designed such that it can be used as radiometer with a thermal
resolution of ∆Tsys = 0.5 K. A bandwidth of B = 3 GHz is chosen as a tradeoff between
high thermal (requires a high bandwidth) and high spacial (requires a low bandwidth)
resolution. To allow fast scanning, the integration time should not exceed τ = 1 ms. The
radiometer formula [15]
Tsys
(1.1)
∆Tsys = √
B·τ
allows deriving the required system temperature of the radiometer system Tsys = 860 K.
If considering a antenna temperature [17, 18, 19] of up to 400 K, the receiver temperature
should not exceed 460 K. The considered antenna temperature represents a high value
in radiometry, however, antenna temperatures in active systems might be much higher.
Since cooling the receiver to cryogenic temperatures is not intended, the receiver noise
figure should be F ≤ 4 dB. A higher value would still lead to an operating radiometer,
but the thermal resolution would be worsened (e.g. [20] reports a radiometer noise figure
of around 8 dB.)
The input signal of the receiver is amplified by a low-noise amplifier (LNA) and filtered
by an image rejection filter. The power levels given in Figure 1.1 are calculated using
the values from the MMICs data sheets and assuming a filter insertion loss of 2.5 dB.
The calculated noise figure of the receiver is F = 2.8 dB.
1.4 Array of Receivers
The proposed system should allow building an array of equal receivers, where every
receiver is connected to an antenna in the antenna array. Each receiver has its own VCO
which is injection-locked to one reference oscillator. The reference oscillator acts also as
the phase reference for all the receivers.
Figure 1.2 shows a system including the introduced receiver array. The antenna array,
the receiver array and the Butler matrix are realized as system blocks, linked together
by connectors. The receiver array consists of modules, with two receivers per module.
The input of the receiver array is connected to the antenna array. To overcome the losses
of lines and connectors, which are used to connect the receivers with the antennas, an
additional LNA will be placed next to each antenna. This LNA increases the gain of the
radiometer and reduces the noise figure additionally. Also the antenna elements should
be low noise
4
1 INTRODUCTION
Antennas with
input amplifiers
Butler-matrix
Receiver with
beamforming capability
To
reference
To
reference
To
reference
To
reference
Antenna
beam 1
Antenna
beam 2
Antenna
beam 3
Antenna
beam 4
Antenna
beam n
Figure 1.2: Schematic diagram of a system with antenna, receiver array and Butler matrix.
For a system with beamforming capability, it would be sufficient to connect the output
ports with a power combiner. Alternatively, a passive distribution system, such as a
Butler matrix [21] or Rotman lens [22], can be used instead of the power combiner.
This combination has the advantage of having several parallel output ports. In such
a system, the integration time can be increased by keeping scanning speed constant.
Thus, a higher thermal resolution can be achieved. In noise sensitive applications, the
use of a passive distribution system at the radio frequency (RF) would not be possible
due to their high losses. Since the losses of distribution systems is much lower at the
IF, and the input signal is amplified by the receiver gain, the noise figure degradation
caused by the IF distribution system is minimal. Additionally, the disadvantage of
the Butler matrix of having fixed beam direction is solved by this combined system,
since the beams formed by the Butler matrix can be moved by the receivers. This
movement is demonstrated by the simulated array factor of an 8 × 8-Butler matrix,
which is represented by the solid and dash-dot-doted lines in Figure 1.3. The Butler
matrix port-to-port phase of [−157.5◦ , −112.5◦ , −67.5◦ , −22.5◦ , 22.5◦ , 97.5◦ , 122.5◦ ,
1.5 OBJECTIVE
5
Figure 1.3: Simulated normalized array factor of a 8×8-Butler matrix (solid and dash-dot-dotted
lines).
157.5◦ ] at the eight respective output ports, is changed to [−157.5◦ + φrec , −112.5◦ +
φrec , −67.5◦ + φrec , −22.5◦ + φrec , 22.5◦ + φrec , 97.5◦ + φrec , 122.5◦ + φrec , 157.5◦ + φrec ]
by the receivers. φrec is the controllable receiver-to-receiver phase shift provided by the
injection-locked VCOs. Thus, the receivers allow moving all the beams in parallel. The
whole space can be scanned, if the scanning range is large enough to move the beams
into the gaps between the solid lines in Figure 1.3. The scanning range of the dash-dotdotted beam is marked by the dashed lines. In the presented example, every receiver
has to provide a phase-tuning range of −22.5◦ ≤ φrec ≤ 22.5◦ .
1.5 Objective
The focus in this dissertation is on the realization of a receiver with low noise figure
and beamforming capability. The receiver-to-receiver distance should match with the
distance between antenna elements. Therefore, the receiver width is limited to 5 mm.
A planar packaging technology is used in order to achieve this small receiver width. A
prototype on a Rogers RT/duroidr substrate is designed such that it can be modified
to a later low-temperature co-fired ceramic (LTCC) implementation. LTCC allows for
an additional size reduction. The LTCC-prototype has not been manufactured since no
in-house LTCC process was available and the foundry costs are significantly higher when
compared to the costs of Rogers RT/duroidr . However, the test of subcomponents on
LTCC at Ka -band frequencies is part of this dissertation.
With the exception of the VCO, all active components are commercially available
monolithic microwave integrated circuits (MMICs). The operating frequencies of the
components are shown in Table 1.1. Since a VCO without buffer is required, the em-
6
1 INTRODUCTION
Element
Frequency f
Bandwidth B
Input amplifier
35 GHz
±1.5 GHz
Output amplifier
5 GHz
±1.5 GHz
Reference oscillator
15 GHz
single-tone signal
Injection-locked VCO
15 GHz
single-tone signal
Amplifiers in locking network
15 GHz
single-tone signal
Frequency doubler output (=LO)
30 GHz
single-tone signal
Table 1.1: Frequencies of receiver components when injection locked.
ployed VCO is a costumer tailored MMIC. As an alternative to this VCO, a hybrid VCO
realized on a Rogers RT/duroidr substrate is presented in this work. This VCO is based
on standard MMICs. Therefore, it represents a much cheaper implementation if small
quantities are required. However, this VCO is much larger than the chip-VCO.
The system bandwidth is not limited by the phase shifters, because of the operation
with a single tone signal. As mentioned before, a limitation is required to get a certain
spacial resolution of the antenna beam. Two different implementations for limiting the
bandwidth to B = 3 GHz are discussed: One with a broadband filter, and one with
a narrowband input filter. The implementation with the broadband filter requires an
additional IF-filter, whereas the implementation with the narrowband filter requires no
further IF-filtering.
The reference oscillator is also implemented as part of this work. A 15-GHz LNA has
been fed back by a cavity resonator. The high Q-factor of the cavity defines the oscillator
Q-factor and the operating frequency. The frequency of the reference oscillator is fineadjusted by a piece of Rogers RT/duroidr put into the cavity.
1.6 Outline
Chapter 2 reviews the fundamentals of injection-locked oscillators. It is explained, why
the phase noise of the injection-locked low-Q VCO is mainly defined by the reference
oscillator. Furthermore, also a model for deriving the resulting phase shift is given. This
model explains, why in this work a phase shift bigger than 180◦ (±90◦ ) and 360◦ (±180◦ )
at the respective fundamental and doubled frequency is possible.
Chapter 3 shows the concept of the reference oscillator. The implementation is based
on a cavity in a copper block and a feedback loop on the Rogers RT/duroidr 6006
substrate. A chip-LNA is used as the active element. The implementation, spectrum
and phase noise measurements are described. Finally, a method for fine-adjusting the
1.6 OUTLINE
7
reference oscillator frequency is shown.
The chip-VCO based phase shifter (VCO with two 15-GHz LNAs) is described in
Chapter 4. The first model is realized on Rogers RT/duroidr 6010 (εr = 10.2) for testing.
This test structure is used for measuring the free-running behavior of the VCO. The
manufactured prototype with the input port connected to the reference oscillator acts as
phase shifter. Therefore, the phase-tuning curve as well as the phase-noise behavior of
the injection-locked VCO at 15 GHz are discussed in this chapter. The influence of the
reference oscillator power level on the VCO phase noise is also measured. Furthermore,
the operation of a second VCO, also realized on a Rogers RT/duroidr substrate using
chip-LNAs, is shown in this chapter. Frequency, phase and phase-noise measurements
are also performed with this VCO.
A test structure with the 35-GHz input LNA, the filter and the mixer is used for testing
the influence of shielding to the noise figure. This test module, described in Chapter
5, is used for experimentally minimizing the receiver noise figure. Two filters – one
realized on Rogers RT/duroidr and one on LTCC – have been simulated, implemented
and measured (Sections 5.4 and 5.5). The results from all previous measurements are
used for designing the receiver on Rogers RT/duroidr 6010 (εr = 10.2). Two receivers
are built as one module. These two receivers are required in order to measure the
receiver-to-receiver phase shift, which demonstrates the beamforming capability of a
receiver array. Also the noise figure is measured with this prototype.
Chapter 6 summarizes the findings of this dissertation and gives an outlook.
8
2 Injection-Locked VCO
Abstract — The fundamental properties of injection-locked oscillators are discussed in this chapter.
It is shown that a phase-tuning range of φ > ±90◦ can be achieved by injection locking a VCO with
high power. Furthermore, it is shown that the higher locking power is also advantageous for the
phase noise behavior. An operation point with wide phase-tuning range and low phase noise can
only be achieved if the injection-locked VCO has a low Q-factor. Finally, the phase noise of VCOs
in an array is discussed. It is shown that the VCO-tree leads to a better phase noise performance as
the VCO-chain.
2.1 Principle of Injection Locking
The principles of nonlinear theory were explained by Van der Pol in the 1930s [23].
These principles include the description of oscillators using models with negative resistances. The operation point of the oscillators is derived by solving 2nd -order differential
equations. The effect of injection locking can be modeled by adding a resistance to an
existing oscillator circuit. Several publications from the late 1940s to the 1970s deal
with injection-locked oscillators [24, 25, 26]. The behavior of noise in synchronized oscillators has also been analyzed [27]. The principle of injection locking is also called
synchronization, phase locking, frequency entrainment, or forced oscillation [28].
Based on this theoretical work, the most important properties of injection-locked oscillators can be summarized as follows.
• Frequency: The output frequency of an injection-locked oscillator is mainly defined by the reference oscillator.
• Phase noise: The oscillator which is used for locking normally has a lower Q-factor
than the reference oscillator. Thus, its phase noise is higher than the reference
oscillator phase noise. When injection locked, the phase noise of the injectionlocked oscillator at low offset frequencies is as good as the phase noise of the
reference oscillator.
• Amplitude: The amplitude of the injection-locked oscillator signal is mainly defined by the oscillator itself. Since the reference oscillator amplitude is normally
much smaller, injection locking can be used for amplifying the reference signal.
When a high-power reference oscillator signal is used, injection locking can also be
used for limiting the amplitude.
• Modulation: When the frequency difference between the two oscillators is too
big, then the oscillator is no longer injection locked. In such an operation point,
the output signal is a modulated signal defined by the signals of the two oscillators.
9
10
2 INJECTION-LOCKED VCO
• Phase difference: A phase difference between the input and output signals of the
injection-locked oscillator appears. This phase shift is dependent on the difference
between the reference oscillator frequency and the free-running frequency of the
injection-locked oscillator.
The locking effects are often unwanted. Oscillators, for example on integrated circuits,
can be locked accidentally. When the Q-factor of the oscillator is low, signals with very
low power are sufficient for locking. To avoid locking, oscillators are made more robust.
Isolating oscillator tanks is important.
Contrary, the described effects are employed for numerous applications: Injection
locking is used for amplifying signals [29, 30], and limiting signals [31, 32]. These applications take advantage of the output power defined by the injection-locked oscillator.
Also systems used for modulating and demodulating signals have been reported [33, 34].
When employing a voltage-controlled oscillator (VCO) for injection locking, the difference in frequency between reference and free-running frequencies can be controlled.
Consequently, phase shifters can be realized based on this principle. As described in the
introduction, phase shifters based on injection-locked VCOs are used in the discussed
receiver. The VCOs are injection locked by a single-tone low-phase-noise signal. The
VCOs output signals are used for driving mixers. The most important attributes for
the implemented phase shifters are described in the following: Section 2.2 gives an estimate for the achieved tunable phase. Section 2.3 shows a literature review and gives an
analytical description of the phase noise of the injection-locked VCO output signal.
2.2 Free-Running Frequency-to-Phase Relation
The injection-locked VCO can be modeled by using a negative-resistance model [28].
The components, which are plotted with solid lines in Figure 2.1, represent the model,
which is used for describing the VCO properties in the following: The capacitor C, the
inductance L, and the admittance GL represent the passive elements. C, L and GL are
used instead of a complex YL , to allow also a description of the Q-factor of the VCO. The
negative resistance −GN represents the active component of the VCO. The free-running
operation point of the VCO is defined by:
− GN = YL (ffr ),
(2.1)
where YL (ffr ) = GL − j/(2πffr L) + j2πffr C. Based
on this equation, the VCO free√
running frequency can be derived to: ffr = 1/(2π LC).
When injection locked, the VCO frequency is forced to be equal to the reference
oscillator frequency f0 . The frequency change of the VCO can be modeled by adding an
additional admittance Y = G + jB in parallel to the VCO equivalent circuit [36]. This
admittance is plotted with dotted lines in Figure 2.1. The forced frequency change can
be expressed by changing the operation point given in Equation (2.1) to:
− GN = Ytot (f0 ),
(2.2)
where Ytot (f ) = YL (f0 )+Y . f0 is the operating frequency of the VCO with the additional
admittance Y .
2.2 FREE-RUNNING FREQUENCY-TO-PHASE RELATION
11
Figure 2.1: Equivalent circuit of VCO with additional admittance.
p
In the following, the external Q-factor Qext = ( C/L)/GL is used for describing
the VCO properties. Furthermore, the frequency ratio v = f /ffr − ffr /f is utilized for
simplifying expressions. In contrast to the considerations in [36], the frequency difference
between the VCO free-running frequency and reference frequency ∆f = |f0 − ffr | is
considered as small. Using these assumptions, Ytot can be written as [37]:
Ytot = GL · (1 + jQext · v) + Y.
(2.3)
Since the absolute values of the admittances are not of interest, Equation (2.3) can be
normalized by using the normalization constant Y0 :
ytot = 1 + g + j(Qext · v + b).
(2.4)
ytot = Ytot /Y0 is the normalized total admittance; g and b are the normalized conductance g = G/Y0 and susceptance b = B/Y0 .
Assuming that the free-running VCO is perfectly matched when connected to a transmission line with the characteristic admittance Y0 , the additional admittance Y causes a
certain mismatch. This mismatch can be described by a reflection factor Γ, as shown in
Figure 2.1. The reflection factor, describing the relation between the output and input
amplitudes of the VCO, can be written by using the normalized parameters:
Γ
=
=
U VCO,out
U VCO,in
Y0 − Y
1 − g − jb
=
.
Y0 + Y
1 + g + jb
(2.5)
The complex reflection factor can also be expressed by amplitude and phase:
Γ = p · ej(φ0 +∆φ) .
(2.6)
In the following, p is called the locking gain and ∆φ is the controllable phase. φ0 is
the constant phase, examined when injection locking the VCO at the operation point
f0 = ffr .
12
2 INJECTION-LOCKED VCO
Equations (2.5) and (2.6) allow finding expressions for g and b:
g
=
b
=
−1 + p2
2p · cos(∆φ) + 1 + p2
−2p · sin(∆φ)
.
2p · cos(∆φ) + 1 + p2
(2.7)
(2.8)
Both, g and b, are periodic functions. Therefore, the extreme values are the lowest
and highest possible values of b. This limitation of b also limits the number of possible
operation points =(ytot ) = 0. In other words, the VCO can only be locked if the difference
between reference and VCO free-running frequencies ∆f = |f0 − ffr | is below a certain
limit. An approximate for an analytical formula can be found in [10]. Most of the
published results assume a small p. This assumption limits the controllable phase to the
range −90◦ ≤ ∆φ ≤ 90◦ [38, 39].
In contrast to the above results, the phase at the locking limits can be found by
deriving the extreme values of b in Equation (2.8), without assuming simplifications.
Since the first negative and first positive extreme values define the phase at the locking
limits, all other periodic extreme values are not of interest [37]:
∆φmax = ± arccos
−2 · p
.
1 + p2
(2.9)
The locking limits can be computed by inserting the phase at the locking limits into
Equation (2.8) and formulating the VCO oscillation condition =(ytot ) = 0. For small
∆f = |ffr − f0 | and v ≈ 2 · (ffr − f0 )/f0 , the locking range can be written as:
[fu − f0 , fl − f0 ]
= ±(ffr − f0 )max
f0
p
= ±
,
·
Qext |1 − p2 |
(2.10)
where fu and fl are the upper and lower locking limits of the injection-locked VCO,
respectively. Using this notation, the VCO is injection locked for free-running frequencies
in the range fl ≤ ffr ≤ fu .
The change of the VCO operation point (YL ) allows controlling the VCO free-running
frequency. Thus, as demonstrated by Equations (2.5) and (2.6), also the phase of the
injection-locked VCO can be controlled. Phase shifters utilizing injection-locked oscillators must be operated within the locking range, because outside the locking range the
output signal is not a single-tone signal but a frequency/amplitude modulated signal
with the modulation frequency fm = f0 − ffr . Signal plots of VCO output signals can
be found in [28, 34].
The maximum phase tuning range, achieved when tuning the VCO from the lower to
the upper locking limit, can be derived using Equation (2.9). Values of 182◦ , 202◦ and
226◦ can be computed using a locking gain of p = −40 dB, p = −20 dB and p = −10 dB,
respectively. These examples show that a phase shift higher than 180◦ can be achieved
by applying high reference power.
Since no simplifications in terms of locking gain p have been made, it is not possible to
find a simple expression for the output phase as a function of the free-running frequency.
2.2 FREE-RUNNING FREQUENCY-TO-PHASE RELATION
(a)
13
(b)
Figure 2.2: Derived phase shift of injection-locked VCO using an external quality factor Qext =
20, frequency f0 = 15 GHz, locking gain p = −40 dB (a) and locking gain p = −10 dB (b).
Therefore, the phase tuning function applying two different reference power levels have
been plotted using MapleTM . For both plots a Q-factor of Qext = 20 and a reference
frequency f0 = 15 GHz is assumed. Figures 2.2(a) and 2.2(b) show the phase vs. the
VCO free-running frequency when injection locked with a locking gain of −40 dB and
−10 dB, respectively.
Figure 2.2(a) shows that the phase tuning range is 182◦ and the locking range is only
2 · 7.5 MHz. This operation point has not only the disadvantage of a low phase tuning
range, but also the locking range is very small. A locking range much smaller than the
VCO frequency-tuning range limits controllability due to the then high sensibility of the
phase to the tuning voltage. Using a higher locking power solves both issues: As shown
in Figure 2.2(b), the locking range and the phase tuning range are increased significantly
when using higher locking power. A locking range of 2 · 0.24 GHz and phase tuning range
of 226◦ can be achieved.
These examples and Equation (2.10) show that increasing the reference signal power
also increases the locking range. Additionally, a low VCO Q-factor has been chosen in
this example to show that a reasonable locking range and a wide phase tuning range
can be achieved. A VCO with high Q-factor would decrease the locking range of the
first example (Figure 2.2(a)), whereas the phase tuning range would stay constant. The
same effect would apply to the second example (Figure 2.2(b)): the locking range would
be decreased and the high phase tuning range would stay constant. Since a certain
locking range is required in order to get a sufficient controllability, and the reference
signal power level in practical realizations can not be increased over a certain limit, a
VCO with sufficient small Q-factor has to be chosen.
14
2 INJECTION-LOCKED VCO
2.3 Output Signal of Injection-Locked Oscillator
It has been shown that the phase noise of an injection-locked oscillator can be derived
by superposing the models of a noisy oscillator injection locked with a noise-free cavity
oscillator and a noise-free oscillator injection locked with a noisy cavity oscillator [42].
The result can be modified to express the phase noise of the injection-locked VCO:
LVCO,l (fm ) = Lref (fm ) · S1 (fm , p, Qext ) + LVCO,fr (fm ) · S2 (fm , p, Qext ).
(2.11)
LVCO,l (fm ),LVCO,fr (fm ), Lref (fm ) are the single side-band phase noise power densities
of the locked and free-running VCO as well as the reference oscillator, respectively. S1
and S2 are the noise-suppression factors [42]:
S1 (fm , p, Qext )
S1 (fm , p, Qext )
1
=
1+
=
1
2
fm
f02
· Qext ·
(2.12)
1
p2
2
fm
· Qext · p12
f02
f2
+ fm2 · Qext · p12
0
.
(2.13)
fm and f0 are the offset frequencies from the carrier and the reference oscillators frequency, respectively. Qext is the external Q-factor of the VCO, and p is the locking gain
as used in the previous section. The
p locking gain can also be expressed by the VCO
input and output power levels: p = PVCO,in /PVCO,out . The VCO input signal is equal
to the reference oscillator signal at the VCO port.
From Equations (2.12) and (2.13) it can be observed that S1 ≈ 1 >> S2 ≈ 0 for small
fm . Hence, the phase noise of the locked VCO close to the carrier is mainly defined by the
reference source. However, S2 is increasing with higher offset frequencies. Consequently
the phase noise of the locked VCO at higher offset frequencies is defined by the phase
noise of the reference oscillator and the free-running VCO. This leads to a degradation
of the phase noise of the locked VCO. In a practical realization, this means that the
phase noise of the locked VCO is as good as that of the reference oscillator at low offset
frequencies and degrades with increasing distance from the carrier. Furthermore, it can
be seen that the noise-suppression factors are a function of the external quality factor
of the VCO and the locking gain. As a result, a low-Q VCO and a sufficient injection
power level are necessary to achieve a large locking range as well as to keep the phase
noise power density low.
To show this effect, the single-sideband phase noise power density of a reference oscillator and a free-running VCO are plotted in Figure 2.3. The phase noise behavior of
these two oscillators is chosen such that it is in the range of the VCOs used in Chapter
4. The noise-suppression factors have been used for calculating the phase noise of the
injection-locked VCO in MapleTM . For the calculation, an external Q-factor of Qext = 20
and a reference oscillator frequency of f0 = 15 GHz has been chosen. These values are
identical to the values used for deriving the phase shift in the previous section.
The curve calculated with the locking gain p = −60 dB shows a very poor phase noise
behavior: The phase noise is only identical to the reference oscillator phase noise up to a
2.4 COMPARISON OF VCO-TREE AND VCO-CHAIN
15
Figure 2.3: Assumed single-sideband phase noise L(fm ) of reference oscillator and VCO as well
as derived single-sideband phase noise of injection-locked VCO. Locking gain used for locking:
p = −60 dB, p = −40 dB and p = −10 dB
offset frequency of fm = 10 kHz. For offset frequencies fm > 1 MHz, the injection-locked
phase noise is close to the phase noise of the free-running VCO. The curve derived with
the locking gain p = −40 dB shows already a much better behavior: The phase noise is
identical to the reference oscillator phase noise up to a offset frequency of fm = 40 kHz.
At this offset frequency, the phase noise is about 20 dB lower than the phase noise of the
injection-locked VCO when locked with p = −60 dB.
The operating point used for plotting the curve with p = −40 dB is identical to the
one used for deriving the phase shift in the previous section. It has been shown that
a much higher locking gain of p = −10 dB leads to a wider locking range and also to
a bigger phase-tuning range. Therefore, also the phase noise has been plotted in the
operation point using the locking gain p = −10 dB. It can be seen that the phase noise
of the injection-locked VCO is as good as the reference oscillator phase noise up to an
offset-frequency of 2 MHz. This is an improvement of 40 dB (at an offset frequency of
1 MHz) compared to the operation point where injection locked with p = −60 dB.
2.4 Comparison of VCO-Tree and VCO-Chain
As mentioned in the introduction, there are basically two topologies of oscillator arrays.
The first one – as shown in the schematic diagram in Figure 2.4 – connects oscillators in
a chain. The diagram shows that the first VCO in the chain is injection locked by the
reference oscillator. The output signal of the first VCO signal is split into two paths,
16
2 INJECTION-LOCKED VCO
Figure 2.4: Schematic diagram of VCO-chain. Attenuators and amplifiers required for adjusting
locking power are not shown.
where the first path is used as output signal, and the second path is used for injection
locking the second VCO. The third VCO is injection locked by the signal of second VCO,
the forth by the third, and the nth by the (n − 1)st . All VCOs are injection locked with
the same locking gain. This is required in order to get an approximately equal phase
tuning behavior of every VCO in the array.
In the literature, many coupled oscillator arrays similar to the one shown in Figure 2.4
are reported (e.g. [43, 44, 45, 46]). The power splitters shown here might be replaced
by another kind of coupling network. The advantage of the chain topology is that the
phase of every VCO can be controlled in respect to the neighboring VCO. Therefore,
calibrating the oscillator array is relatively simple. Disadvantages are that the phase
error of the nth VCO output signal is equal to the sum of the phase errors of all VCOs
in the chain. This can lead to a noticeable error in big arrays. Furthermore, the failure
of a VCO makes also all VCOs after the failed one inoperative.
The phase noise of the first VCO in the chain can be calculated by using Equation
(2.11). However, for calculating the phase noise of the higher order VCOs, the phase
noise of the VCO used for locking has to be considered. Therefore, the phase noise of
the nth VCO LVCO,l,n (fm ) can be written as:
LVCO,l,n (fm )
= LVCO,l,(n−1) (fm ) · S1 (fm , p, Qext )
+ LVCO,fr,n (fm ) · S2 (fm , p, Qext ).
(2.14)
S1 (fm , p, Qext ) and S2 (fm , p, Qext ) are the noise-suppression factors. LVCO,l,(n−1) and
and LVCO,fr,n are the single-sideband phase noise of the injection-locked (n − 1)st and
the free-running nth VCO, respectively. The equation makes clear that the phase noise
is getting worse from VCO to VCO.
In order to show this effect, the phase noise has been derived with MapleTM and plotted
in Figure 2.5. The calculation has been performed using the same operation points as for
the phase and phase noise calculations in the previous sections. Figure 2.5(a) shows the
phase noise of the chain VCOs when injection locking with a locking gain of p = −40 dB.
The plotted phase noise of the reference oscillator, free-running and first injection-locked
VCOs are identical to the curves presented before. The phase noise of the higher order
VCOs are also plotted. Comparing the curve of the 1st VCO with the 11th VCO shows
2.4 COMPARISON OF VCO-TREE AND VCO-CHAIN
17
(a)
(b)
Figure 2.5: Supposed single-sideband phase noise L(fm ) of reference oscillator and free-running
VCO as well as derived single-sideband phase noise of first and higher order injection-locked
VCOs in VCO chain. VCO properties used for calculation: external Q-factor Qext = 20,
reference oscillator frequency f0 = 15 GHz, locking gain p = −40 dB (a) and p = −10 dB (b).
a degradation of the phase noise by 10 dB at an offset frequency of 100 kHz. The offset
18
2 INJECTION-LOCKED VCO
Figure 2.6: Schematic diagram of VCO-tree. Attenuators and amplifiers required for adjusting
locking power are not displayed.
frequency, where the phase noise of the injection-locked VCO is getting worse than the
reference oscillator, is lowered from 40 kHz to 10 kHz. For bigger arrays, the degradation
is even worse as shown by the plotted curve for the 101st VCO.
Figure 2.5(b) shows the derived phase noise when assuming a higher locking gain of
p = −10 dB. As expected, the phase noise is better. However, the effect of a worsened
phase-noise performance of the higher-order VCOs is also significant.
Since a phase-noise degradation of 10 dB and 20 dB for 10 and 100 VCO chains, respectively, is significant, and also the reliability of the VCO-chain is worse than the
reliability of the VCO-tree, a tree-architecture is used in the following implementation.
Figure 2.6 shows the schematic diagram of injection-locked VCOs which are all directly
injection locked by the reference oscillator [47, 48]. The realization of such a structure
is more complex due the necessity of a 1 : n power splitter. In this architecture the
reference oscillator signal also acts as a phase reference. The phase must be controlled
with respect to the global reference, and not to the neighbor VCO as it is the case in
the VCO-chain.
Since every VCO is directly injection locked by the reference oscillator, all VCOs have
the same low phase noise, as defined by Equation (2.11). Furthermore, the failure of one
VCO causes only the breakdown of one VCO, and all neighboring VCOs are functional.
Finally, the phase error of one VCO is defined by the VCO itself.
2.5 Conclusion
It has been shown that injection locking a VCO allows controlling the phase relation
between the input and output signal of the VCO. In the injection-locked state, the VCO
tuning voltage can be used for changing the phase. A high locking gain leads to a big
phase-tuning range. In comparison to other methods used for increasing the phase tuning
range [49, 50], the proposed method leads also to a wide locking range which leads to a
small phase sensitivity.
Furthermore, it has been shown that the phase noise of the injection-locked VCO can
be derived from the free-running VCO and reference oscillator phase noise. Numerical
experiments show that a phase noise performance almost as good as the reference oscillator phase noise can be achieved when the VCO is injection locked with high locking
2.5 CONCLUSION
19
gain.
The calculations assume a low external Q-factor of the VCO. A higher Q-factor reduces
the locking range. A locking range, much smaller than the free-running VCO frequency
tuning range, would limit controllability. Furthermore, the sensitivity to noise on the
tuning-DC-voltage would increase. Theoretically, a VCO with bigger Q-factor locked
with higher locking gain would lead to a similar locking range. The drawback of this
concept is that the reference signal power level at the phase shifter input is limited
by the saturation power level of the used amplifiers. Increasing the power levels by
using power amplifiers is not useful because of the high losses and heat dissipation.
Finally, operating the components close to their saturation points causes a non-linear
behavior. This operation point does not allow correct setting of the locking gain. It
can be concluded that a VCO with sufficiently low Q-factor is indispensable for the
phase-shifter implementations described in the following.
It has been shown that the VCO-tree is more reliable than the VCO-chain and exhibits
also a lower phase noise for all employed VCOs.
20
3 Reference Oscillator
Abstract — The implementation of the reference oscillator is discussed in this chapter. The oscillator
is realized based on an amplifier fed back by a high-Q cavity. The TM0,1,1 resonance mode of the
cavity is used as a compromise between high Q-factor and compact oscillator size. A comparison of
the measured oscillator noise to the phase noise of a commercial synthesizer demonstrates excellent
phase noise behavior of the oscillator. The low phase noise and the simple architecture is the reason
for employing this reference oscillator in the receiver.
3.1 Introduction
Since the reference oscillator is used for locking all receivers, it has been implemented
as a sperate part. As described in the previous chapter, the reference oscillator phase
noise determines also the phase noise of the injection-locked VCO. Thus, a 35-GHz
low-phase-noise oscillator is required for the operation of the receivers.
Since MMICs operating at Ku -band frequencies are commercially available, an oscillator based on MMIC amplifiers has been developed. The used amplifier is fed back by
a cavity resonator. The schematic of the oscillator is shown in Figure 3.1. The principle
of the oscillator can be understood by using the loop condition [23]. (The equivalence
between the loop condition and the device-load lines model can be found in [52].) The
shown circuit oscillates, if the complex voltages at an arbitrary point of the oscillation
loop are equal:
V in = V out .
(3.1)
Figure 3.1: Schematic of reference oscillator. Oscillator is pointed out by dashed rectangular.
Output amplifier used for buffering signal and isolating oscillator tank is also shown.
21
22
3 REFERENCE OSCILLATOR
V in and V out are the in- and output-voltages at the reference point. This condition can
be split into a gain and a phase condition:
Gloop (f0 ) > 0 dB
φloop (f0 ) = n · 2π
(3.2)
where Gloop (f0 ) is the gain and φloop (f0 ) is the phase of the oscillator loop at the oscillation frequency f0 . The gain of the loop is defined by the amplifier gain Gamp , the
transmission factor of the cavity resonator S21res , and the losses of the planar transmission lines LTL have influence on the gain:
Gloop (f ) = Gamp (f ) · S21res (f ) · 1/LTL (f ).
(3.3)
The design of the loop and the resonator has to guarantee that the oscillation conditions
are fulfilled for the operation frequency only. This guarantees that the oscillator has an
unique oscillating frequency.
3.2 Cavity Design
As a compromise between a high Q-factor and compact realization, a rectangular cavity
with the dimension H = 4 mm, L = 15.15 mm, B = 13 mm using the TM0,1,1 -mode has
been chosen. This choice can be shown by the normalized Q-factors and the resonance
frequencies of the different cavity modes in Table 3.1. Normalizing the Q-factor allows
a frequency independent comparison of the Q-factor. The Q-factors have been derived
on basis of [53]. It can be seen that, for example, the TM0,1,3 -mode would have led to
a much higher Q-factor, but building a cavity with this mode at a Frequency of 15 GHz
would also increase the cavity size significantly. The table shows further that within the
lower modes, the TM0,1,1 -mode is the one with the highest Q-factor. This is the reason
for using this mode in this oscillator. As apparent in Table 3.1, the resonance frequency
of the used mode is 15.193 GHz. This – in comparison to the targeted frequency of
15 GHz – higher value has been chosen in order to have some margins for later frequencyadjustments.
The cavity is milled into a block of copper (part 1) and closed by another copper
block (part 2). As shown in Figure 3.2(a), two holes are drilled into the cap in order
to insert two coupling posts. The insulation between the copper body and the coupling
post is glass. A 10 mil thick Rogers RT/duroidr 6006 (εr = 6.15) substrate is adhesively
bonded to the top of part 2. The MMICs are placed in mounting holes of the substrate,
on top of part 2. The microstrip lines are connected to the feedthroughs and amplifiers
by wire bonds. The photograph in Figure 3.2(b) shows the metallic block with the SMA
connectors on the side.
Due to the high current density on the coupling posts, their length also influences
the Q-factor and frequency of the resonator. Figure 3.3 shows the in Ansoft HFSSTM
simulated insertion loss of the cavity. The different plots represent simulations with
different length of the coupling post. It can be seen that the insertion loss of the resonance
is reduced for long posts. At the same time, the Q-factor of the cavity is getting smaller.
3.2 CAVITY DESIGN
TEl,m,n / TMl,m,n
modes
23
Normalized Q-factor
Q·
δ
λl,m,n
Resonance frequency
fl,m,n
TE0,1,0
0.115
9.894 GHz
TE0,0,1
0.134
11.530 GHz
TE0,1,1
0.078
15.193 GHz
TM0,1,1
0.177
TE0,2,1
0.120
TM0,2,1
0.267
TE0,1,2
0.128
TM0,1,2
0.292
TE0,1,3
0.184
TM0,1,3
0.420
TE1,0,0
0.118
37.473 GHz
TE1,1,1
0.213
40.435 GHz
22.902 GHz
25.093 GHz
35.977 GHz
Table 3.1: Derived resonator (H = 4 mm, L = 15.15 mm, B = 13 mm) resonance frequencies
and unloaded Q-factors [53]. δ is the skin depth of the metallization surrounding the cavity
and λl,m,n is the resonant free-space wavelength: λl,m,n = c0 /fl,m,n .
The derived Q-factor, shown in Table 3.2, shows that the dependence on the coupling
post length is significant.
For the oscillator, a cavity with a high Q-factor and sufficient low insertion loss is
required. In the practical realization, the resonator has been designed for an insertion
loss of S21 (f0 ) = −15 dB. An oscillator with this cavity and an amplifier with a gain
of Gamp = 22 dB fulfills the oscillation condition and allows for additional losses of the
used lines. Simulations (Figure 3.3) show that coupling posts with a length lcp ≈ 0.3 mm
lead to the targeted insertion loss.
To take a possible error of the simulated value into account, the coupling posts have
been shortened to a length of 0.5 mm. The measured insertion loss is plotted in Figure
24
3 REFERENCE OSCILLATOR
(a)
(b)
Figure 3.2: Cross section (a) and photograph (b) of cavity resonator.
3.4. Since the insertion loss is below 10 dB, the length of the coupling posts have been
shortened to 0.3 mm. A Q-factor of Q ≈ 3000 and a insertion loss of S21 = −15 dB have
been measured after shortening. Figure 3.5 shows a picture of a shortened coupling post.
3.3 Oscillator
A photograph of the manufactured oscillator can be seen in Figure 3.6. The two
feedthroughs, which are used for coupling to the cavity, are marked in the lower part
of the photograph. The marked measurement ports, which were used for performing
the S-parameter measurement of the cavity, are visible on the left and right side of the
3.3 OSCILLATOR
25
Figure 3.3: Simulated cavity insertion loss.
Length of coupling posts
Q-factor
lcp
Insertion loss
Resonance frequency
S21 (f0 )
f0
0 mm
6000
−33 dB
15.204 GHz
0.3 mm
3100
−16 dB
15.191 GHz
0.5 mm
2200
−10 dB
15.182 GHz
1 mm
840
−3 dB
15.138 GHz
2 mm
120
−0.6 dB
14.860 GHz
Table 3.2: From simulation data calculated Q-factor of resonator when varying length of coupling posts.
photograph. The feedthroughs are connected to the measurement ports by means of wire
bonds (points A and B). In order to put the oscillator into operation, the bond wires
connecting the measurement ports at points A and B are removed. New wire bonds are
set such that the feedthroughs are connected with the oscillator loop. A variable line is
visible in the left upper corner. This line can be used for changing the electrical length
26
3 REFERENCE OSCILLATOR
Figure 3.4: Measured insertion loss of resonator.
Figure 3.5: Photograph of coupling post.
of the oscillator loop experimentally, if the oscillation conditions of the oscillator are not
fulfilled in the set operation point.
As discussed earlier in this Chapter, the cavity TM0,1,1 -mode has been used. Since
the feedthroughs are in the center of the cavity width, only TEl,m,n and TMl,m,n with
odd n can be excited in the cavity. Table 3.1 shows that the only existing cavity modes
within the amplifiers operation band [54] are the TM0,1,1 and TM0,2,1 -modes. To prevent
3.4 MEASUREMENT RESULTS AND FREQUENCY FINE-ADJUSTMENTS
27
Figure 3.6: Photograph of manufactured oscillator.
the oscillator from oscillating at the not wanted frequencies, stubs are added to block
transmission in the unwanted frequency range.
3.4 Measurement Results and Frequency Fine-Adjustments
The spectrum has been measured after putting the oscillator into operation. Figure
3.7(a) shows a plot of one of the measurements performed with a span of 20 kHz. The
achieved reference oscillator frequency is f0 = 15.183 GHz. The oscillation frequency fits
very well to the measured resonance frequency of the cavity.
In order to move the oscillator frequency closer to the targeted 15 GHz, the oscillator
has been frequency-adjusted. Therefore, a 10 mil tick piece of of Rogers RT/duroidr 6010
(εr = 10.2) has been included into the cavity floor. The substrate is cut as rectangular;
the length of the substrate is equal to the cavity length, the width has been used for
adjusting the resonance frequency. Figure 3.8 shows the insertion loss of the cavity
TM0,1,1 mode simulated in Ansoft HFSSTM for different substrate widths.
Based on the simulation, a width of 2.6 mm has been chosen for the implementation.
Figure 3.9 shows the cavity floor with the bonded piece of substrate. Since the frequency
change is minor, also the phase-change in the oscillator loop is very small. Thus, a
modification of the variable line is not required.
The measured spectrum after inserting the substrate is plotted in Figure 3.7(b). The
achieved frequency is f0 = 14.975 GHz. This frequency is 0.17% smaller than the targeted frequency of 15 GHz. Despite of this small error, the frequency was not adjusted
further, although this would be possible by changing the size of the glued substrate. The
comparison of the signals in Figures 3.7(a) and 3.7(b) shows that the signal quality is
similar. Also the simulation of the cavity showed that the degradation of the Q-factor
28
3 REFERENCE OSCILLATOR
(a)
(b)
Figure 3.7: Measured spectrum of oscillator before (a) and after (b) frequency adjustment.
Spectrum measured with HP 8565E spectrum analyzer.
Figure 3.8: Transmission parameter S21 of frequency-adjusted cavity. Different plots are simulated with different width of substrate.
by inserting the substrate is not noticeable.
The measured single-sideband phase noise L(fm ) [55] is plotted in Figure 3.10. Ex-
3.4 MEASUREMENT RESULTS AND FREQUENCY FINE-ADJUSTMENTS
29
Figure 3.9: Photograph of resonator cavity with strip of substrate for frequency fine-adjusting.
Figure 3.10: Measured single-sideband phase noise power density L(fm ) of manufactured reference oscillator and Agilent E8267C synthesizer.
cellent phase-noise values of L = −88 dBc/Hz and L = −115 dBc/Hz at the respective
offset frequencies of fm = 10 kHz and fm = 100 kHz have been found. Also the comparison with the measured phase noise of the commercial Agilent E8267C synthesizer shows
that the achieved phase noise represents an excellent result. The synthesizer measurements show a shoulder in the measured curve around the offset frequency of 100 kHz.
This behavior is typical for PLLs due to their locking mechanism. Also low-phase-noise
PLLs (e.g. [56]), show such a behavior. The built reference oscillator however, exhibits
30
3 REFERENCE OSCILLATOR
Parameter
Value
Frequency f0
14.975177 GHz
Bias voltage
3V
Output power
+8 dBm
Single sideband phase noise
@ 10 kHz offset
−88 dBc/Hz
@ 100 kHz offset
−115 dBc/Hz
Q-factor of Cavity
3000
w/o freq. tuning dielectric
Table 3.3: Measured parameters of reference oscillator.
Part
Component
Oscillator amplifier
Hittite HMC 516
Output amplifier
Hittite HMC 516
Oscillator loop substrate
Rogers RT/duroidr 6006
Substrate on Cavity floor
Rogers RT/duroidr 6010
Table 3.4: Components used in reference oscillator:
no shoulder. For this reason, the synthesizer phase noise is lower close to the carrier,
whereas the reference oscillator exhibits better phase noise behavior at higher offset
frequencies.
Tables 3.3 and 3.4 give a summery of the measured parameters and the used components, respectively.
3.5 Conclusion
The implementation of the 15-GHz reference oscillator has been described in this chapter.
The oscillator is based on a high-Q cavity. The comparison of the measured phase noise
3.5 CONCLUSION
31
with the phase noise of a commercial synthesizer confirms the high quality of the output
signal. Additionally, a method for adjusting the output frequency has been introduced.
This method has been used to set the oscillators frequency close to the targeted 15 GHz.
32
4 Phase Shifter
Abstract — Two different implementations of phase shifters are shown in this chapter. Both are
based on injection-locked VCOs. The first implementation utilizes chip-based VCOs. Circuits with
input and output amplifiers are used to isolate the not buffered oscillator from the output port and
to allow injection locking. The second implementation is based on an amplifier fed back by a tunable
low-Q resonator. The second implementation represents a solution which only requires commercially
available components. The disadvantage of the second implementation is the in comparison to the
first implementation much bigger size. Both implementations lead to a phase tuning-range of about
200◦ . Due to the use of VCOs injection-locked to a reference oscillator with excellent phase noise
behavior, the phase shifter output signals exhibit also low phase noise.
4.1 VCO Requirements
Commercial VCOs have high isolation between the VCO output port and their oscillator
tanks. An isolation of 40 dB or more is common in monolithically integrated VCOs.
This isolation makes the oscillator independent of the load. Since overcoming the high
isolation is not possible, buffered VCOs can not be used for injection locking. Therefore,
oscillators without buffer are required (e.g. [58]).
In this dissertation, two different concepts of injection-locked VCO-based phase shifters
are tested. The first concept utilizes an unbuffered VCO developed in the Electronics
Laboratory at ETH Zurich. A network for separating the input and output signals of
(a)
(b)
Figure 4.1: Phase shifter based injection-locked VCO utilizing circulator (a) and VCO with
input port (b).
33
34
4 PHASE SHIFTER
the VCO is required. The schematic of such a network is shown in Figure 4.1(a). The
shown model has the advantage that the circulator provides perfect decoupling of the
VCO input and output signals (V VCO,in , V VCO,out ). Therefore, this circulator based
model is also used in theoretical concepts (e.g. [28]). The amplifiers provide additional
gain to the input and output signals and increase the output-to-VCO and VCO-to-input
isolations. The shown network acts as phase shifter, since the phase can be steered by
changing the VCO operating point.
Another possible phase shifter architecture is employing a VCO with input and output
amplifiers, as shown in Figure 4.1(b). In this model, both amplifiers are directly connected to the VCO; a decoupling network is not required. The output amplifier provides
gain to the output port and isolates the VCO tank from the output port. The input port
however, provides matching to the tank but also allows signal injection. Since such a
VCO can only be used for injection locking application, it is not available as off-the-shelf
chip.
Both approaches (injection locking a not buffered VCO and a VCO with input port)
are tested in this work. Section 4.2 shows the implementation of a phase shifter utilizing
an unbuffered VCO. Different networks for separating the input and output signal have
been tested and the size of the phase shifter is reduced to a minimum. Section 4.3
shows the implementation of a VCO with input port. The VCO circuit is similar to the
architecture of the reference oscillator (Chapter 3), but a low-Q resonator with varactor
tunability is used instead of the high-Q cavity.
4.2 Chip-VCO based Phase Shifters
4.2.1 VCOs
Figure 4.2(a) shows the schematic diagram of a single-ended VCO without output buffer
designed in the Electronics Laboratory of ETH Zurich. This design is similar to the
one introduced in [59]. The circuit has been manufactured on the IBM BiCMOS 6HP
process (0.25-µm BiCMOS technology [60]). The transistor base and collector require two
different bias voltages (Vbase , Vdd ). This kind of biasing is an advantage for the prototype
implementation, since changing the bias-voltage allows adjusting the frequency of the
VCO. However, a change of the biasing network is required in a later series-production
to reduce the frequency-to-bias-voltage dependence. The varactor diode and the tuning
port are placed in the emitter-circuit. Due to the orientation of the varactor diode Cvar ,
negative voltages are required for tuning the VCO.
An alternative VCO has been developed on basis of the IBM 7WL-process (0.18µm BiCMOS technology [61]). The schematic diagram is shown in Figure 4.2(b). The
comparison of the schematics shows that the biasing of the 6HP VCO is very problematic
since the bias lines are not decoupled from the VCO. This has been solved in the 7WL
VCO [62]. Another difference is the smaller tuning range, which is caused by the smaller
value of the varactor capacitance in the 7WL process.
4.2 CHIP-VCO BASED PHASE SHIFTERS
(a) 6HP-based
35
(b) 7WL-based
Figure 4.2: Schematic diagram of VCOs provided by the Electronics Laboratory.
4.2.2 VCO Load
Since the VCO has no output amplifier, the load impedance becomes a crucial part.
The negative resistance model [63] shows that the load line has to be loop-free to avoid
frequency jumps in the VCO tuning range. Although the exact value of the load line
is not known, from the schematic diagram can be seen that the load line can only be
loop-free if the load connected to the VCO is also loop-free.
Figure 4.3 shows the frequency tuning behavior measured on-chip using a wafer prober.
Two discontinuities can be observed. These discontinuities are caused by two loops in
the load line. Furthermore, it is visible that the frequency jumps take place at a different
point when lowering the tuning voltage and when increasing the tuning voltage Vtune .
The reason for this hysteresis can also be found in the loops.
The measurement plotted in Figure 4.3 has been performed with the 6HP-based VCO.
Although the used wafer probers and the connected cables provide good matching, the
high phase change over the frequency caused by the long lines and a minimal mismatch
lead to frequency jumps in the measurement.
4.2.3 Phase Shifter Circuits
Since a loop-free load line is required for continuous frequency tuning of the VCO, the
phase shifter network must provide a loop-free load to the VCO and must have not only
an output port but also an input port for injection locking. If these conditions are not
fulfilled, the phase shifter can not operate reliably.
As described earlier, a loss-less circulator provides a good possibility of decoupling the
VCO input and output signals. However, employing circulators (e.g. [64, 65]) leads to
36
4 PHASE SHIFTER
Figure 4.3: Measured frequency-tuning behavior of VCO with 6HP BJT. Measurement has been
performed using wafer probers.
large structures due to necessity of ferrites. Implementing them into an LTCC-module
is difficult. Additional disadvantages of circulators operating at Ku -band frequencies are
their losses.
Due to all these disadvantages, two much cheaper and smaller networks have been
tested for locking the VCO. The first tested network is a T-junction including input and
output amplifiers (Figure 4.4(a)). The input and output amplifiers provide a VCO-toinput and output-to-VCO isolation. The disadvantage of this structure is the crosstalk
from the input port to the output port. For this reason, the phase tuning range at the
output port of the network might be smaller as at the VCO output.
To overcome the disadvantage of the high crosstalk of a T-junction, a second approach
has been tested. This topology is based on a rat-race coupler [66]. To reduce the VCOto-output losses and the return from the output of the input amplifier, a 8-dB instead of
a standard 3-dB coupler has been realized (Figure 4.4(b)). The disadvantage of having a
8-dB attenuation of the input signal is overcome by the gain of the input amplifier. An
additional advantage of this structure is that the VCO can be matched. The forth port
of the coupler is terminated, and the the input port is isolated from the output port.
An Ansoft Nexximr model has been used for optimizing the T-junction and the coupler. Since the not perfectly matched amplifiers have influence on the return loss at
the VCO port (port 1) of the T-junction and the coupler, the measured amplifier data
has been included in this simulation. Figure 4.5(a) shows the schematic of the circuit
model. The T-junction and coupler is simulated as an Ansoft Designerr MoM model,
but parameter variations are controlled from the Ansoft Nexximr circuit simulator. The
4.2 CHIP-VCO BASED PHASE SHIFTERS
(a)
37
(b)
Figure 4.4: Schematic of T-junction (a) and coupler (b) for injection locking VCO.
geometries of the T-junction and the coupler have been tuned to get a loop-free S11 .
Figures 4.5(b) and 4.5(c) confirm the loop-free behavior of the simulated T-junction
and the coupler return loss in the desired frequency band between 14 GHz and 16 GHz.
At this stage, both structures are usable as phase shifter networks. As expected, the
T-junction does not provide good matching at any frequency point, but the return loss
variation over the frequency is relatively small. This small fluctuation is mainly caused
by the small size of the structure. In contrast, the coupler return loss includes a perfectly
matched frequency point, but the fluctuation of the return loss is much bigger. Also the
loop-free frequency band is smaller compared to the T-junction.
Both structures have been manufactured on an Rogers RT/duroidr 6010 substrate.
Figures 4.6(a) and 4.7(a) show the dimensions and a photograph the T-junction connected to the input and output amplifiers and the VCO. Both amplifiers are Hittite
HMC516-amplifiers [54]. The VCO connected to the T-junction can be seen below the
junction. All visible microstrip lines – including the T-junction – are 50 Ω-lines. The amplifier bias voltage is provided through a line visible on top of the photograph, whereas
the three DC voltages required for the VCO are provided by lines shown at the bottom
of the photograph. All DC-lines are connected to chip-capacitors by spiral inductors.
The DC-pads on the active chips are connected with the chip-capacitors by wire-bonds.
The inductors are of the Microwave Component 10-1847 GSA-type [67]. The inductors
together with the capacitors are used for reducing the noise to the DC-voltages.
Figures 4.6(b) and 4.7(b) show the dimensions and the photograph of the 8-dB-coupler.
The input amplifier and the VCO can be seen on the photograph, but the output amplifier is not visible due to the big distance between the coupler and output amplifier.
The respective thinner and thicker lines within the coupler are 80 Ω and 38 Ω lines. This
impedance system has been used, since a 60 µm thick line is the absolute thinnest structure which can be etched in the IFH in-house process. Due to this impedance-system
λ/4-transformers are used at the three used ports. The transformer is a 41 Ω line. The
terminated port is terminated with a SMD resistor and an open λ/4-line. A resistor
with the size of 0.6 × 0.3 mm and a λ/4-line instead of a short to ground have been used,
since they showed best performance in comparison measurements.
38
4 PHASE SHIFTER
(a)
(b)
(c)
Figure 4.5: Schematic of circuit simulator model (a) and simulated return loss (port 1) of
T-junction (b) and coupler (c).
Figure 4.8(a) shows the coupler insertion loss measured with a coupler identically
to the one shown in Figure 4.7(b). The measured coupler has no mounting holes for
the amplifiers and the VCO. Therefore, SMA connectors with integrated microstrip-toSMA transition are mounted to the three ports. The forth port is terminated. The
measurement results show that the desired 8 dB attenuation between input port and
VCO has been achieved in the operation frequency band between 14 GHz and 16 GHz.
Furthermore, the excellent value of < −25 dB for the isolation between input and output
port can be seen in the plot. The losses between VCO and output port are in the range
of 3 dB, and the return loss at every port (Figure 4.8(b)) is Sn,m < −17 dB.
4.2 CHIP-VCO BASED PHASE SHIFTERS
39
(a)
(b)
Figure 4.6: Dimensions of T-junction (a) and coupler (b).
4.2.4 Free-Running Frequency and Pushing
Both circuits have been used for measuring the free-running tuning-behavior of the
VCOs. The input port of the phase-shifter network is terminated with a 50 Ω-load
in these measurements. Figure 4.9(a) shows the measured free-running frequency vs.
the tuning voltage. The 6HP VCO has been measured in the operation point Vbase =
1.28 V, Vdd = 1.80 V and the 7WL VCO in the operation point Vbase = 0.98 V, Vdd =
2.72 V. The comparison between the two measured tuning curves shows that the 7WL
oscillator has a significant smaller tuning range than the 6HP VCO. Figure 4.9(b) shows
the frequency tuning behavior of the 6HP VCO connected to the coupler. The VCO
operation point in this measurement is Vbase = 1.54 V, Vdd = 1.11 V. The 7WL VCO
has not been measured with the coupler.
All three tuning curves show a continuous tuning behavior. The comparison between
these results and the previously shown wafer-prober measurements (Figure 4.3) shows
that the described requirements to the return loss are fulfilled for all three circuits (Tjunction with 6HP and 7WL VCO and coupler with 6HP VCO). However, the different
form of the 6HP VCO frequency tuning curve when connected to the T-junction and
40
4 PHASE SHIFTER
(a)
(b)
Figure 4.7: Photograph of phase shifters incorporating T-junction (a) and coupler (b) for injection locking VCO.
the coupler shows that the different network impedances change the frequency-tuning
behavior.
VCO pushing has also been tested. Figures 4.10(a) and 4.10(b) show the influence
of a collector and base voltage variation on the VCO frequency, respectively. The measurements have been performed with the 6HP VCO connected to the T-junction and a
tuning voltage Vtune = 2.5 V. For the measurements with the swept collector and base
voltages, the same base and bias voltages as used in the measurement plotted in Figure 4.9(a) have been used. The measurements show a strong influence of the collector
and the base voltages. For this reason, DC-sources with excellent voltage stability are
required.
4.2 CHIP-VCO BASED PHASE SHIFTERS
(a)
41
(b)
Figure 4.8: Insertion loss (a) and return loss (b) of measured rat-race coupler. Fourth port is
terminated with a 50 Ω resistor. Port notation is given in Figure 4.6(b).
(a)
(b)
Figure 4.9: Measured free-running frequency of VCO with T-junction (a) and coupler (b). Input
port is terminated with a 50 Ω-load. VCO operation points: 6HP VCO with T-junction:
Vbase = 1.28 V, Vdd = 1.80 V; 7WL VCO with T-junction: Vbase = 1.54 V, Vdd = 1.11 V; 6HP
VCO with coupler: Vbase = 0.895 V, Vdd = 0.646 V.
4.2.5 Maximized Locking Range and Phase-Tuning Behavior
Figure 4.11 shows the VCO locking lange and tuning range. When used for phase
shifting, the operation point of the VCO has to be chosen such that the VCO operates
within the locking range. The output frequency of the phase shifter is the reference
oscillator frequency f0 (Section 2.2) and the output phase is:
φphaseshifter = φ0 + ∆φ
(4.1)
42
4 PHASE SHIFTER
(a)
(b)
Figure 4.10: Oscillation frequency ffr vs. VCO collector voltage Vdd (a) and base voltage Vbase
(b). VCO operation points: tuning voltage Vtune = 2.5 V, base voltage Vbase = 1.28 V (a)
and collector voltage Vdd = 1.80 V (b).
Figure 4.11: VCO tuning and locking range.
where φphaseshifter is the phase of the phase-shifter output signal, φ0 is the constant phase
of the phase-shifter and ∆φ is the phase controlled by the VCO.
As shown in Figure 4.11, the locking range should not exceed the VCO free-running
tuning range, because a locking range bigger than the free-running frequency tuning
range would limit the phase tuning range. Additionally, a small phase-tuning sensitivity
(dφ/dVtune ) should be found to increase the controllability. These requirements can be
visualized by using the parameters of the VCOs: The free-running frequency tuning
range can be controlled from fmin to fmax by changing the tuning voltage between
4.2 CHIP-VCO BASED PHASE SHIFTERS
43
Figure 4.12: Measurement setup for optimizing locking range and phase-tuning measurements.
Vtune,max = 0 and Vtune,min = −5. If the locking range is set such that the lower
and the upper boundaries correspond to the tuning voltages of Vtune,l,max = −0.5 and
Vtune,l,min = −4.5, the phase-tuning sensitivity is dφ/dVtune ≈ 50 degree/V. When using
a tuning-voltage source with an accuracy of 1 mV, the phase accuracy is 0.05◦ . Reducing
the locking range also reduces the phase accuracy.
In order to find the operation range of the phase shifters, the measurement set-up of
Figure 4.12 has been used (Photographs of the measurements set-up can be found in
Appendix A.3). The reference signal output power level is set such that the 15-GHz
amplifier output signal is high enough for driving the mixer. Therefore, the attenuator
is required for adjusting the phase shifter input power level. A 10-kHz ramp signal is
used in this experiment for finding the operation range. The mixer IF signal and the
ramp signal are connected to an oscilloscope.
Figure 4.13 shows the ramp signal and the detector output (mixer IF) signal. The
detector output signal has been measured with four settings (0◦ , 90◦ , 180◦ , 270◦ ) of the
variable phase shifter. A periodic signal with a frequency equal to the ramp generator
frequency can be seen within the locking range. In contrast, the signal outside the locking
range is not a DC-voltage. The locking limits are clearly visible. The reference oscillator
power and the VCO bias voltages have been adjusted to get a wide and symmetric
locking range, respectively.
Phase shifter operation ranges with a wide symmetric locking range have been found
44
4 PHASE SHIFTER
Figure 4.13: Oscilloscope plot of detector (in four different operation points) and ramp signal.
Operation point of 6HP VCO: Vdd = 1.80 V, Vbase = 1.28 V. Reference oscillator power level:
Pin = −28 dBm.
for both networks, the VCO connected to the T-junction and to the coupler. Table 4.1
shows a summary of the VCO operation ranges, with the corresponding input power and
the phase-tuning range. As explained in the previous section, the coupler has been built
with the design goal of a lower crosstalk between the input and output ports. Therefore,
the coupler was expected of having a higher phase-tuning range.
In contrast to these estimations, the measurements showed a wider phase-tuning range
of the circuit with the T-junction. This is caused by the following reasons:
• The optimization of the phase shifter operating range showed that the VCO connected to the T-junction requires a reference oscillator power level of only Pin =
−28 dBm. Since this power level is so small, the power transmitted from the Tjunction input port to the output port is small too. Therefore, the T-junction
causes no measurable reduction of the VCO phase tuning range.
• The measured free-running frequency tuning curve of the VCO connected to the
coupler shows that the tuning sensitivity at the lower tuning range limit is getting
high (dffr /dVtune (Vtune = 0 V) ≈ 500 MHz/V). The high frequency sensitive of the
free-running VCO results in a high phase sensitive of the injection-locked VCO
(d∆φ/dVtune ≈ 80 degree/V). Small variations of the tuning voltage make the
4.2 CHIP-VCO BASED PHASE SHIFTERS
Network / VCO
Operation point
45
Input power
Phase-tuning range
Vdd [V]
Vbase [V]
Pin [dBm]
∆φmax [Degree]
T-junction / 6HP
1.80
1.28
−28
210◦
Coupler / 6HP
1.90
1.10
−32
160◦
T-junction / 7WL
2.72
0.98
−34
200◦
Table 4.1: Measured phase-tuning range with T-junction and coupler.
locking limits fuzzy and phase measurements close to the locking limits are instable.
Due to this reason, the phase tuning range appears smaller.
Since a nonlinear simulation of the system was not possible due to the lack of the VCO
parameters, this experimental result demonstrates the feasibility of using the T-junction
in the phase shifter. Although it has been expected that the coupler leads to a bigger
phase tuning range, the experiments show that the T-junction exhibits a wider tuning
range. Since the T-junction is also smaller in size, it has been used in all the following
implementations. The comparison of the VCOs with 6HP and 7WL BJTs shows similar
results. The VCO based on the 7WL BJT requires less power, because of the smaller
free-running frequency-tuning range.
The detector voltage Vdetector is proportional to the sine of the phase difference between
reference and measurement signal (sin(φreference /φmeasurement )). For this reason, the
detector has its highest accuracy around Vdetector = 0. To have the highest phasemeasurement accuracy, the ramp generator and the oscilloscope have been replaced by
a battery and a voltmeter. After this exchange, the measurement set-up acts as a
measurement bridge; the phase changes can be read from the scale of the variable phase
shifter. When a battery is used for the tuning voltage, the fluctuation of the detector
voltages are in the order of 1 mV or less. This is equivalent to a phase fluctuation of 0.1◦
or less.
Figure 4.14 shows the measured phase shift of the T-junction based phase shifter with
both VCOs. Both tuning curves exhibit the required continuous and wide scanning
range.
The high influence of the bias voltages on the free-running frequency has been shown in
Section 4.2.4. The phase change of the injection-locked VCO can be estimated using these
results. Table 4.2 shows that a collector voltage variation of Vdd ± 0.1 V leads to a phase
change of ±75◦ . This phase change is equal to a frequency change of ±0.15 GHz in the
free-running state. The base voltage leads to a higher frequency and phase dependence.
The change of the VCO operation point changes not only the phase, but also the
locking range. Thus, a bias voltage variation can move the locking range out of the VCO
tuning range, which leads to a reduction of the phase tuning range. To limit the phase
error caused by bias-voltage fluctuations to 2◦ , the voltages should vary less than 1 mV.
46
4 PHASE SHIFTER
Figure 4.14: Measured phase-tuning curve of phase shifter with T-junction connected to 6HP
VCO and 7WL VCO. Operation points: Vdd = 1.80 V, Vbase = 1.28 V, Pin = −28 dBm (6HP
VCO) and Vdd = 2.72 V, Vbase = 0.98 V, Pin = −34 dBm (7WL VCO).
Voltage
Frequency variation
Phase variation
Vdd ± 0.1 V
±0.15GHz
±75◦
Vbase ± 0.1 V
∓0.4GHz
∓200◦
Table 4.2: 6HP VCO bias-voltage-to-phase sensitivity.
4.2.6 Phase Noise
Since the frequency of the free-running VCO is fairly unstable, phase-locking the VCO
to another oscillator is not possible. For this reason, a delay line frequency discriminator
has been used for measuring the phase noise of the free-running and the injection-locked
VCO. The measurement set-up (Figure 4.15) includes the HP 5500. A 50 ns delay line
is selected, because this delay line provides accurate measurement results up to offset
frequencies of 4 MHz [68].
The measured phase noise of the free-running VCO is shown in Figure 4.16. The
phase noise is plotted for two operation points (Vdd = 1.886 V, Vbase = 1.101 V, Vtune =
0 V and Vtune = −2.9 V) to demonstrate that the change of the tuning voltage has no
influence on the phase noise. As expected, the free-running VCO exhibits high phase
noise. The measured reference oscillator phase noise is plotted for comparison. This
4.2 CHIP-VCO BASED PHASE SHIFTERS
47
Figure 4.15: Phase noise measurement set-up.
Figure 4.16: Measured single-sideband phase noise of free-running, and injection-locked VCO.
Used operation points: Vdd = 1.886 V, Vbase = 1.101 V.
reference oscillator has been used for injection locking the VCO with different reference
oscillator power levels. It can be seen that the phase noise of the injection-locked VCO
is similar to the reference oscillator phase noise for small offset frequencies, but gets
worse at higher offset frequencies. The offset frequency, from which the phase noise of
48
4 PHASE SHIFTER
Parameter
Value
Bias voltage Vdd
1.80 V
Base voltage Vbase
1.28 V
Tuning voltage Vtune
15-GHz buffer amplifier bias voltage
Free-running frequency ffr
−5 V – 0 V
3V
14.7 GHz – 15.2 GHz
Single sideband phase noise (free running)
@ 10 kHz offset
−45 dBc/Hz
@ 100 kHz offset
−70 dBc/Hz
Reference signal frequency f0
Reference signal power level f0
15 GHz
−28 dBm
(at phase shifter input port)
Phase tuning range ∆φmax
Output frequency when locked f0
210◦
15 GHz
Single sideband phase noise (injection locked)
@ 10 kHz offset
−88 dBc/Hz
@ 100 kHz offset
−115 dBc/Hz
Table 4.3: Measured parameters of chip VCO-based phase shifter.
the injection-locked VCO deviates from the phase noise of the reference oscillator, is
strongly dependent on the locking power, which can be seen from the measured curves.
To show the dependence of the phase noise to the reference oscillator power, low
reference oscillator power levels have been used for locking the VCO. The power levels,
which are used when operating the circuit as phase shifter, are in the range between
−28 dBm and −34 dBm. As it can be seen form the measured values, the phase noise
of the injection-locked VCO is similar to the reference oscillator phase noise up to the
4.3 FEED-BACK LOOP BASED VCO USED AS PHASE SHIFTER
Part
Component
All 15-GHz amplifiers
Hittite HMC 516
VCOs
Internally developed VCOs based on IBM 6HP [60]
49
and 7WL [61] technology
Substrate for networks
10 mil Rogers RT/duroidr 6010
Table 4.4: Components employed in chip-VCO based phase shifter.
Figure 4.17: Feed-back loop based VCO architecture.
offset frequency of 1 MHz or higher for the used power levels.
Table 4.3 shows the measured parameters of the phase shifter for the used operation
point. Table 4.4 gives a summary of the used components.
4.3 Feed-Back Loop based VCO used as Phase Shifter
4.3.1 VCO Architecture
The schematic of the feed-back loop VCO is shown in Figure 4.17. The main difference
to the reference oscillator is the resonator. Instead of the reference oscillator cavity,
a tunable low-Q resonator is used in the VCO. The schematic diagram shows also the
input amplifier. The input is used for injecting power to the VCO.
The operation of the VCO can be described using the amplitude and phase conditions
as used for the reference oscillator in Chapter 3 (Equation (3.2)). The VCO free-running
frequency ffr can be calculated similar as the reference oscillator frequency f0 .
50
4 PHASE SHIFTER
(a)
(b)
(c)
(d)
(e)
Figure 4.18: Photographs of tested resonators: Varactor diodes put in field zero of a λ/2 (a)
and λ (b) resonator, one (c) and two (d) varactor diodes in λ-resonator field maxima and
ring resonator (e). Resonators are realized on Rogers RT/duroidr 6010 . Tuning is enabled
using M/A-COM MA46H120 varactor diodes.
The required gain is provided by the amplifier and resonator. Tunability of the VCO
is provided by the varactor-tunable resonator. The input and output amplifiers are
used for isolating the VCO from the output port. Additionally, the amplifiers buffer
the signals. The amplifier mismatch and the lines connecting the amplifiers are taken
into consideration when finding the oscillator loop length to meet the oscillator phase
condition.
4.3.2 VCO Resonator
The resonator requirements can be derived from the oscillation conditions. An amplifier
gain of Gamp = 20 dB is used for the following considerations. The resonator insertion
loss has to be S21 (f = ffr ) > −Gamp at the VCO free-running frequency. The freerunning frequency is identical to the resonator frequency. The insertion loss at all other
frequencies has to be S21 (f 6= ffr ) < −20 dB in order to avoid frequency jumps of the
VCO. Also when tuning the resonator, parasitic resonances have to be blocked.
To find a resonator which fulfills these requirements and exhibits a frequency-tuning
range of about 1 GHz around the center frequency of 15 GHz, different resonator types
have been simulated in Ansoft Designerr and Ansoft HFSSTM . All resonators are
microstrip-based. The Rogers RT/duroidr 6010 substrate has been used because of
its high dielectric constant of εr = 10.2. Testing the VCO on a substrate with similar
properties as LTCC simplifies a later implementation on LTCC. In all tested resonators,
4.3 FEED-BACK LOOP BASED VCO USED AS PHASE SHIFTER
Resonator
Tuning range
51
Insertion loss
Length / number varactors /
lower freq.
upper freq.
S21 [dB]
varactor position
fl [GHz]
fu [GHz]
(f = fl )
(f = fu )
λ/2 / one varactor / field zero
11.6
15.1
−10
−7
(Figure 4.18(a))
(13.0)
(16.6)
(−3)
(−3)
λ / one varactor / field zero
14.5
16.1
−10
−12
(Figure 4.18(b))
(15.9)
(17.4)
(−4)
(−7)
λ / one varactor / field maxima
14.3
15.3
−14
−5
(Figure 4.18(c))
(15.1)
(16.6)
(−4)
(−2)
λ / two varactors / field maxima
13.20
13.25
−14
−12
(Figure 4.18(d))
(14.9)
(16.4)
(−5)
(−5)
Ring resonator
9.7
13.6
−22
−19
(Figure 4.18(e))
(12.5)
(14.4)
(−4)
(−6)
Table 4.5: Comparison of measured and simulated (in brackets) insertion loss and tuning range
of tested microstrip-based resonator.
tunability is achieved by changing the electrical length of the resonator using the M/ACOM MA46H120 varactor diode [69]. The range of the varactor diode capacitance is
0.14 pF − 1.1 pF. The varactor diodes are large (0.7 mm). For this reason, also the
diodes body is influencing the resonators. Measurements allow for a realistic comparison between the different models and the architecture with highest tuning range can be
chosen.
Photographs of the microstrip resonators are shown in Figure 4.18. Figures 4.18(a) and
4.18(b) show varactors in the field zeros of a λ/2 and λ resonator, respectively. Figures
4.18(c) and 4.18(d) show λ resonators with 1 and 2 varactors in the field maxima. The
second 2-varactor based resonator has the advantage of decoupling the varactor tuning
voltage from the resonator. Figure 4.18(e) shows the implementation of a varactor-tuned
ring resonator [70].
Table 4.5 shows the measured tuning range and the insertion loss at the resonance
frequencies. The simulated values are given in brackets. The insertion loss is given for
52
4 PHASE SHIFTER
the measurements made at the lower and upper frequency-tuning limits.
The measurement of the λ/2-resonator with the varactor in the location of the field zero
shows a tuning range from 11.6 GHz to 15.1 GHz. The comparison with the simulated
tuning range from 13 GHz to 16.6 GHz shows that the tuning range is similar, but the
tuning frequency of the measured resonator is about 1.5 GHz lower. The reason for this
frequency shift is the electrical length of the varactor diode which has not been taken
into account when simulating the structure. The variation of the insertion loss over the
tuning range is relatively low. Although this resonator has a wide tuning range, it can
not be employed due to the too low frequency. Additionally reducing the size would
worsen the couplings and make bonding the varactor diodes impossible. Due to this
reason, λ-resonators have been implemented. Both, the resonator with one varactor in
the field zero and one varactor in the field maximum exhibit good tuning behavior.
The ring resonator shows a wide tuning range. However, this resonator is too lossy to
fulfill the discussed oscillation conditions. Additionally, the measured frequency band is
much lower in comparison to the targeted band around 15 GHz. The losses are mainly
caused by the small size, which allows only lossy couplings. Stronger couplings are
not possible at the targeted size, since the used lithography process does not allow a
further reduction of the gap width. Similar to the λ/2-resonator, reducing the size of
the resonator is not possible due to the given size of the varactor diode. For these
reasons, also this ring resonator is not suitable for the VCO. Larger ring resonators
with a electrically longer ring have also been tested. These size-increased rings showed
additional parasitic resonances caused by the structure. These resonators have not been
tested further, since these resonances are changing the tuning behavior of the resonator
and may cause frequency jumps when used in the VCO.
Out of the tested λ-resonators, the λ-resonator with a varactor diode at the location of
the field zero exhibits a wide frequency tuning range and a small variation of the insertion
loss. For this reason, this resonator type has been chosen for the VCO realization.
Figures 4.19(a) and 4.19(b) show the simulated and measured insertion loss of the λresonator with a varactor diode in the field zero. The three curves are simulated with
varactor capacitances of Cvar = 1.1 pF, 0.29 pF and 0.14 pF for the voltages Vtune = 0 V,
5 V and 10 V. In this case, a frequency shift of about 1.5 GHz between the simulated
and measured values is observed.
4.3.3 VCO Circuit and Free-Running Frequency
As explained at the beginning of this section, the insertion loss at all frequencies f 6= ffr
has to be S21 < −20 dB. Stubs are used for suppressing the resonances at 8 GHz and
4.5 GHz. The measurement in Figure 4.19(b) shows that another stub for suppressing
the resonance at 19.5 GHz is necessary to avoid possible frequency jumps of the VCO.
Figure 4.20 shows the photograph of the realized VCO. The dimensions of the circuit
were derived on basis of the resonator measurement results. The oscillator amplifier can
be seen in the center of the photograph, the resonator with the varactor diode is visible
below the amplifier.
Similar to the free-running frequency measurements of the chip-VCO (Chapter 4.2.4),
the input port is terminated with a 50 Ω load when measuring the free-running frequency.
4.3 FEED-BACK LOOP BASED VCO USED AS PHASE SHIFTER
53
(a)
(b)
Figure 4.19: Simulated (a) and measured (b) insertion loss of λ-resonator with varactor diode
in field zero.
54
4 PHASE SHIFTER
Figure 4.20: Photograph of feed-back loop based VCO.
Figure 4.21: Measured free-running frequency at the 15-GHz output port.
All amplifiers are biased with a DC-voltage of 3 V. The measured tuning curve of the
free-running frequency is plotted in Figure 4.21. The VCO free-running frequency range
4.3 FEED-BACK LOOP BASED VCO USED AS PHASE SHIFTER
55
Figure 4.22: Measured phase-tuning range. VCO is locked to reference oscillator at the frequency f0 = 15.90 GHz with power level Pin = −9 dBm.
is from ffr = 15.4 GHz to ffr = 16.2 GHz.
The oscillator amplifier gain G3V varies by ±1 dB for a bias voltage between 2.5 V
and .5 V. Changing the bias voltage between 2 V and 4 V led to a frequency change of
10 MHz or less.
4.3.4 Phase-Tuning Behavior
The VCO input port is used for injection locking the VCO. The output signal phase
can be changed by changing the VCO varactor voltage Vtune . Similar to the findings
in Section 4.2.5, an optimal operation point with a locking range slightly smaller than
the VCO frequency tuning range has been found. In comparison to the chip-VCO, the
operating frequency of this VCO can not be changed by adjusting the bias voltage. For
this reason, the reference oscillator frequency has been adjusted in order to get a wide
locking range.
Figure 4.22 shows the measured phase tuning range of the injection-locked VCO.
The used measurement set-up is identical to the one used for measuring the chip-VCO
based phase shifter (shown in Figure 4.12 in Chapter 4.2.5). A reference signal with the
frequency of 15.9 GHz and input power of −9 dBm has been used for locking the VCO.
The measured phase tuning range is 200◦ .
56
4 PHASE SHIFTER
Figure 4.23: Measured single-sideband phase noise power density of feed-back loop based VCO.
Operation point: Vbias = 3.0 V, reference signal input power level Pin = −47 dB, Pin =
−31 dB and Pin = −21 dB.
4.3.5 Phase Noise
The measurement set-up used for measuring the phase noise of the feed-back loop based
VCO is identical to the one used for measuring the chip-based VCO (shown in Figure 4.15
in Chapter 4.2.6). The measued single-sideband phase noise of the reference oscillator,
the free-running and the injection-locked VCOs is plotted in Figure 4.23. The lowest
reference power used for injection locking this VCO is Pin = −47 dBm. Also in this curve
can be seen that the phase noise of the injection-locked VCO is getting much better when
increasing the reference oscillator power level. A reference power level of Pin = −9 dBm
has been found as an optimal operation point for phase shifting. With this power level,
the phase noise of the injection-locked VCO is almost equal to the phase noise of the
reference oscillator.
Table 4.6 gives a summery of the measured phase shifter parameters in the used
operation point and Table 4.7 shows a list of the used components.
4.4 Conclusion
Two phase shifters – both based on injection-locked VCOs – have been developed, implemented and tested. A not-buffered chip-VCO is used in the first phase shifter type. Since
this VCO is very sensitive to its load, input and output amplifiers have been added to the
VCO. Frequency jumps of the free-running VCO and phase jumps of the injection-locked
4.4 CONCLUSION
Parameter
Bias voltage
Tuning voltage Vtune
Input and output amplifier bias voltages
Free-running frequency ffr
57
Value
3V
0 V – 10 V
3V
14.7 GHz – 15.2 GHz
Single sideband phase noise (free running)
@ 10 kHz offset
−58 dBc/Hz
@ 100 kHz offset
−81 dBc/Hz
Reference signal frequency f0
Reference signal power level f0
15.90 GHz
−9 dBm
(at phase shifter input port)
Phase tuning range ∆φmax
Output frequency when locked f0
210◦
15.90 GHz
Single sideband phase noise (injection locked)
@ 10 kHz offset
−88 dBc/Hz
@ 100 kHz offset
−115 dBc/Hz
Table 4.6: Measured parameters of feed-back loop based phase shifter.
VCO are suppressed by these amplifiers. The chip VCOs allow adjusting their operation
point by changing the bias voltages. This is used for finding an operation point with a
wide locking range. Since the operation point is very sensitive to the bias voltages, a
voltage source with a highly constant output voltage (±1 mV) has to be used. Another
disadvantage of this phase shifter is the required unbuffered VCO. Since these VCOs
are not available as standard MMICs, customer specific foundry processes are required.
These processes makes the chips expensive.
The second phase shifter type employs a MMIC LNA which is fed back by a tunable
resonator. Two additional amplifiers are added to provide input and output buffering.
58
4 PHASE SHIFTER
Part
Component
Input, output and oscillator amplifiers
Hittite HMC 516
Varactor diode
M/A-COM MA46H120
Substrate
10 mil Rogers RT/duroidr 6010
Table 4.7: Components used in feed-back loop based phase shifter.
The input LNA is used for injection the reference signal, whereas the output LNA
provides buffering of the VCO signal. This phase shifter is bigger than the phase shifter
with the chip-VCO.
The comparison between the phase noise of the free-running VCOs (Figures 4.16 and
4.23) shows that the phase noise of the feed-back loop VCO is about 10 dB lower than
the phase noise of the chip-VCO. The better frequency stability allows measuring the
locking state with lower reference power. This has been shown in several experiments:
The lowest measured reference power used for locking the feed-back loop based VCO
is Pin = −47 dBm, whereas the lowest reference power used for locking the chip-based
VCO is Pin = −40 dBm.
The lower phase noise indicates a higher Q-factor. Therefore, an optimal operation
point requires a higher locking gain. The phase measurements have been performed
using a reference power level of Pin = −28 dBm and Pin = −9 dBm for injection locking
the feed-back VCO and the chip VCO, respectively. Since the VCO operation point is
set such that the locking range is slightly smaller than the free-running frequency tuning
range, a phase tuning range of about 200◦ has been measured with both VCO-types.
The locking ranges of the VCOs depend on the reference power, also the tuning-voltageto-phase relation is highly dependent on the reference power level. To guarantee a phase
accuracy of ±0.5◦ , the power variations have to be kept below ±0.1 dB after calibrating
the voltage-to-phase relation.
5 Receiver
Abstract — The implementation of a low-noise receiver with beamforming capability is shown in
this chapter. Two different image-rejection filters are presented: A broadband microstrip filter and
a narrowband SIW filter. The SIW filter utilizes the 3D-integration possibilities of LTCC. The influence of the filter on the receiver noise figure is demonstrated experimentally with a test module.
This test module has also been used for testing the employed MMIC components. Two receivers are
implemented on basis of the test module to demonstrate the phase-control properties of a receiver
array. The measured controllable phase and noise figure are presented.
5.1 Receiver Architecture
As described in the introduction, the mixer of a receiver is driven by the frequency
doubled signal of one of the introduced phase shifters. Every receiver in an array has
its phase shifter, but all phase shifters are injection locked by one reference oscillator.
This reference oscillator acts not only as the global phase reference, but also defines the
phase noise of the phase shifter output signal. Since the reference oscillator exhibits a
very low phase noise, the mixer is driven with a low phase noise signal.
The drawback of this concept is the loss of the mixer. Unfortunately, all 35-GHz
mixers have a relative high conversion loss and thus a high noise figure. The used Hittite
HMC329 mixer data sheet [72] gives a typical and maximum conversion loss of around
9 dB and 11 dB, respectively. The measured conversion loss of different mixer samples
showed values closer to the maximum value.
5.2 Low-Noise Down-Converter
5.2.1 Input LNA and Lower Sideband Suppression
The block diagram of the receiver is shown in Figure 5.1. The noise figure of the receiver
without filter is [73, 74]:
F =
1
Nin (fUSB ) · G(fUSB ) · F (fUSB ) + Nin (fLSB ) · G(fLSB ) · F (fLSB )
·
. (5.1)
G(fUSB )
Nin (fUSB )
Nin (fUSB ) and Nin (fLSB ) are the input noise power in the upper and lower side bands,
respectively.
G(fUSB,LSB ) = Gamp (fUSB,LSB )/Lmx (fUSB,LSB ) and
F (fUSB,LSB ) = Famp (fUSB,LSB ) + Fmx (fUSB,LSB )/Gamp (fUSB,LSB )
are the receiver gain and noise figure in one side band.
59
60
5 RECEIVER
Figure 5.1: Block diagram of receiver input.
The specifications for the filter can be derived from Equation (5.1): The insertion
loss in the upper side band should be as small as possible, and the insertion loss in the
lower side band should be bigger than the receiver gain. For this reason, a filter with an
insertion loss of < 2.5 dB in the upper side band and > 30 dB in the lower side band is
targeted. Therefore, Equation (5.1) can be simplified to:
F =
1
Nin (fUSB ) · G(fUSB ) · F (fUSB ) + Nin (fLSB )
·
≈ F (fUSB ).
G(fUSB )
Nin (fUSB )
(5.2)
The data sheet values used for calculating the receiver gain and noise figure are summarized in Table 5.1. Measurements of the mixer showed that the losses of the mixer
are close to the maximum value given in the data sheet. The measured values of the
LNA are close to the given typical values. For this reason, the typical values given in
the LNA data sheet and the maximum values given in the mixer data sheet have been
used in the following considerations. The mixer noise figure is assumed to be identical
to the mixer loss. The calculated gain and noise figure of the receiver are G = 5 dB and
F = 2.8 dB. For comparison, the noise figure without filter would be F = 9.2 dB.
The receiver gain of only G = 5 dB is sufficient, since the receiver will be connected
to an antenna module which includes the antenna and an additional LNA. The gain
and noise figure of a system consisting of the receiver and an additional LNA will be
G = 24 dB and F = 2.2 dB.
5.2.2 Filter Concepts
Beside blocking the lower side band and having a low insertion loss in in the frequency
range of the upper sideband, there are no further requirements to the filter for minimizing
the receiver noise figure. However, due to the frequency dependence of the antenna array
5.3 FILTER SIMULATION AND OPTIMIZATION METHOD
Component
LNA
Mixer
Filter
Attribute
61
Value @
Value @
35 GHz
25 GHz
Hittite
Gamp [dB] (typical value)∗
19
24
HMC263 [75]
Famp [dB] (typical value)∗
2.1
2.5
Famp [dB] (maximum value)
2.6
3.3
Hittite
Lmx [dB] (typical value)
8.6
9
HMC329 [72]
Lmx [dB] (maximum value)∗
11.5
11.5
Lfilter [dB]∗
2.5
> 30
Table 5.1: Gain, loss and noise figure (data sheet values) of LNA, mixer and filter. Values used
for noise-figure calculation are marked (∗ ).
factor, the bandwidth of the received signal should be limited further. As a trade-off
between high thermal resolution (requires a high bandwidth), and high spacial resolution of the electronic beamforming network (requires small bandwidth), a bandwidth of
around 3 GHz has been chosen.
Two different approaches are possible for implementing the low-loss receiver and limiting the band: The first approach uses a narrowband RF filter (Figure 5.2(a)), whereas
the second approach uses a broadband RF filter and an additional IF filter (Figure
5.2(b)). The approach with the narrowband filter requires a more complex filter which
might have a higher insertion loss compared to a broadband filter. On the other hand
side, the approach with the broadband RF filter requires the implementation of two
different filters.
Both approaches are addressed in this work: A microstrip-based filter with 3 resonators has been developed and tested (Section 5.4). This filter is used for measuring
the low receiver noise figure (Section 5.6). The microstrip-based receiver implementation
(Section 5.7) includes also this filter. The receiver is designed such that it can easily be
realized on LTCC-technology. Since LTCC allows for a more advanced filter design, a
substrate-integrated-waveguide (SIW) filter has been designed and tested (Section 5.5).
5.3 Filter Simulation and Optimization Method
The filters are modeled as shunt-inductance-coupled filters [82] (also called inductanceiris-coupled waveguide filters [83]). The cross section in Figure 5.3 shows the iris coupled resonator filter. The dimensions of the waveguide (width and height) are constant
through the filter and define the waveguide impedance Z0 . Irises, which are inserted
62
5 RECEIVER
(a)
(b)
Figure 5.2: Schematics of receivers with narrowband filter (a) and broadband filter with additional IF-filtering (b).
Figure 5.3: Cross section of Shunt-inductance-coupled waveguide filter showing electrical length
of resonators (φ) and normalized inductance of irises (x).
into the waveguide, separate the resonators. The irises are modeled by their inductances, which are used in the normalized form. The normalized inductance between the
xth and (x + 1)st can be expressed by the inductance and the waveguide impedance:
xk,k+1 = Xk,k+1 /Z0 . The electrical length of the k th resonator is then [82]:
1
φk = 180◦ − [arctan(2 · xk−1,k ) + arctan(2 · xk,k+1 )]
2
(5.3)
5.4 BROADBAND MICROSTRIP FILTER
63
Figure 5.4: Cross section of microstrip filter and amplifiers.
This equation shows that the resonator length is only dependent on the length of a
reference resonator at the filter resonance frequency and the two irises which are limiting
the resonator.
In order to make use of commercial field solvers, the normalized inductances can also
be used to express the insertion loss of every iris. The S-parameter between the k th and
(k + 1)st resonator can be written as:
s
4 · xk,k+1
Sk,k+1 =
(5.4)
1 + 4 · xk,k+1
Since these parameters show that the irises are independent from each other, they
can be simulated and optimized independently. Furthermore, it has been shown that
also the resonators are only dependent on the two adjacent irises. Therefore, also the
resonators can be optimized independently when knowing the parameters of the irises.
This way, optimization time can be shortened due to the much lower complexity of the
optimization models.
The calculation of the resonator insertion loss is done on basis of a shunt-inductancecoupled filter. The derived insertion loss provides a technology-independent way of describing a coupling. Therefore, every filter topology can be optimized with this method.
5.4 Broadband Microstrip Filter
5.4.1 Filter Topology
The filter is realized with parallel coupled microstrip resonators on a Rogers RT/duroidr
6010 substrate. This substrate is identical to the substrate used for connecting the
active components (e.g. for the T-junction in the phase shifters as shown in Chapter 4).
Therefore, integrating the filter into the receiver is simplified since only one substrate
type is required. The embedding of the filter into the receiver is shown in Figure 5.4.
The substrate is adhesively bonded to a metal block. This metal block is part of the
housing and acts also as heat sink for the active components. MMICs are placed in
substrate holes. The holes are milled into the substrate and the metal block. The depth
of the cavities is chosen such that the top of the MMIC is on the same height as the
metallization of the substrate. Wire bonds are used for connecting the microstrip lines
with the MMICs.
64
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Tschebychev parameters
Insertion loss
[dB]
T (0) = 0.306570
S1,0 = −1.092027
T (1) = 0.853467
S2,1 = −4.817070
T (2) = 1.103879
S3,2 = −4.817070
T (3) = 0.853467
S4,3 = −1.092027
T (4) = 0.306570
Table 5.2: Derived Chebyshev parameters and resonator insertion losses for 35-GHz microstrip
filter.
(a)
(b)
Figure 5.5: Sketch of geometries used for optimizing insertion loss (a) and length of resonators
(b).
5.4.2 Simulation and Optimization
The Chebyshev parameters have been calculated for a 3-resonator 3-GHz bandpass filter
[84]. The insertion loss of the couplings is derived (Table 5.2). In a next step, Ansoft
HFSSTM simulation models are set up to simulate the couplings and the resonators.
Figure 5.5(a) shows the geometries used for optimizing the couplings, and Figure 5.5(b)
shows the geometries used for optimizing the resonator lengths.
In a next step, the filter has been simulated as a whole. Due to the previous optimization, the result is already very close to the optimum. An optimization which includes
all parameters has been performed to improve the filter result additionally. In this simulation, small variations only of the parameters were allowed. The simulated insertion
loss and return loss are plotted in Figure 5.7 by the dashed and solid lines, respectively.
5.4 BROADBAND MICROSTRIP FILTER
65
(a)
(b)
Figure 5.6: Geometry (a) and photograph (b) of 35-GHz microstrip-based filter.
5.4.3 Realization and Measurement
The filter outline and a photograph of the realized filter are shown in Figures 5.6(a) and
5.6(b).
The measured insertion loss and return loss of the filter are plotted in Figure 5.7.
Measured and simulated parameters show a good agreement. The measured insertion
loss in the passband is around 0.5 dB higher than in the simulation. The two peaks which
can be seen in the measured return loss within the passband are caused by detuned
resonators. The reason for it can be found in manufacturing tolerances. The tolerances
of etched metal lines with a width of around 50 µm and a thickness of 17 µm are in the
range of 10%.
When comparing the measured filter with the expected values for the return loss in
the LSB and USB, then it is apparent that the measured filter parameters fulfill the
requirements: the lower sideband is suppressed by > 30 dB, whereas the insertion loss
in the upper side band exhibits a value of about 3 dB. The measured insertion loss of
3 dB is slightly higher as the targeted 2.5 dB. This higher insertion loss will lead to an
increase of the noise figure of 0.04 dB or less.
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Figure 5.7: Simulated and measured insertion loss (solid lines) and return loss (dashed lines).
5.5 Narrowband SIW Filter in LTCC
5.5.1 Filter Topology
Substrate-integrated waveguides (SIWs) [85] have been used for building compact waveguide structures. In this technology, structures similar to waveguides are realized in planar technologies. Commonly, the lower and upper side walls are metallizations, and
the vertical boundaries are via fences. This leads to a much smaller size in comparison to traditional waveguide structures, and to lower losses than microstrip or stripline
structures. Examples of SIW devices are a 180◦ 3-dB directional coupler at Ka -band
frequencies [86], waveguide-to-microstrip transitions at 6 GHz [87], and antennas at frequencies up to 10 GHz [88, 89]. Also SIW-filter for frequencies around 10 GHz [90, 91]
have been demonstrated. LTCC-implementations of 10-GHz filters [92] have also been
demonstrated.
The implementation of a SIW-based filter has not only the advantage of low loss, but
the SIW-filter together with the active elements on one LTCC module leads to a very
high packaging density. This is shown in the drawing in Figure 5.8: The amplifiers are
placed in cavities. Thermal vias between the amplifier bottom and the bottom of the
LTCC module act as heat sink. The SIW-filter (dark gray block in Figure 5.8) is placed
underneath the microstrip lines.
The drawing in Figure 5.9(a) shows a cross section of the microstrip-to-SIW transition.
The SIW is placed in the lower two layers (layer 1 and layer 2). The metallization below
layer 1 and above layer 2 as well as the SIW wall vias are the waveguide walls. The 3D-
5.5 NARROWBAND SIW FILTER IN LTCC
67
Figure 5.8: Cross section of LTCC module with integrated narrowband SIW filter.
drawing of the first resonator in Figure 5.9(b) shows that the SIW wall vias are placed in
two rows to provide a good isolation. The catch pad between layer 1 and layer 2 (catch
pad of the wall vias) is realized as a big metalization, as shown in the cross section in
Figure 5.9(c).
The microstrip-to-SIW transition is realized by a via, as shown in the vertical cross
section in Figure 5.9(a). It is connected to the microstrip line on top of layer 3, to
the catch pad between layers 2 and 3, and to the coupling pad between layers 1 and
2. As visible in Figure 5.9(b), this microstrip-to-SIW transition is the first coupling
of the shunt-inductance-coupled filter. This way, the first coupling can be adjusted by
changing the length of the microstrip stub lstub , as shown in Figure 5.9(c). Since the
layer transition is part of the filter, the filter size and losses are reduced.
All the other couplings are irises, which are realized by rows of vias. The insertion
loss of the couplings is determined by the openings between those rows of vias (d1 and
d2 ). The lengths of the resonators are marked with l1 , l2 and l3 in Figure 5.9(c).
5.5.2 Simulation and Optimization
The simulation and optimization processes are similar to the processes used for optimizing the microstrip filter. Table 5.3 gives an overview of the filter parameters. The
couplings have been optimized such that their insertion loss is equal to the insertion
loss derived from the filter parameters (Table 5.3). Figures 5.10(a) and 5.10(b) show
structures of the simulation geometries used for optimizing the inner couplings (irises)
and the first coupling (microstrip-to-SIW transition). The iris openings d1 and d2 are
used as variables for optimizing the insertion loss of the inner couplings, and the length
of the microstrip stub is used as variable for optimizing the insertion loss of the first
coupling.
The geometries shown in Figures 5.10(c) and 5.10(d) are used for optimizing the
resonator lengths. Also here, two different models are required because of the different
first coupling. The length l2 and l3 are found by optimizing the structure shown in
Figure 5.10(c), whereas the length l1 is found by optimizing the structure shown in
Figure 5.10(d).
In a final step, the complete filter has been simulated. An optimization, in which
small variations only of the six parameters were allowed, has been performed to improve
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(a)
(b)
(c)
Figure 5.9: Vertical cross section of LTCC module (a), 3D-drawing of first filter resonator (b)
and horizontal cross section of realized SIW filter (c).
the filter parameters further. The simulated insertion loss and return loss are plotted in
Figure 5.11 with the solid and dashed lines, respectively. Since the foundry allowed only
manufacturing with the Heraeus HeraLockr HL2000 tape [93], the nominal values of
the tape for the dielectric constant and the loss-tangent (εr = 7.3, tan δ = 0.0026) have
been used for all simulations.
The advantage of the Heraeus HeraLockr HL2000 tape is the zero-shrinkage. Therefore, the filter can be realized with very low tolerances. The data sheet specifies the
loss-tangent at a frequency of 2.5 GHz. A much higher loss-tangent can be expected
at the frequency of 35 GHz. Since neither the tape data sheet nor the foundry give
an estimate about the loss-tangent at the targeted frequency, simulations with different
loss-tangent have been performed. The results of these simulations are also plotted in
Figure 5.11. The solid and dashed lines represent the insertion loss and return loss,
5.5 NARROWBAND SIW FILTER IN LTCC
(a)
(c)
69
(b)
(d)
Figure 5.10: Geometries of simulation models used for optimizing insertion loss of irises (a,b)
and length of resonators (c,d).
Tschebychev parameters
Insertion loss
[dB]
T (0) = 0.136591
S1,0 = −3.648076
T (1) = 0.973228
S2,1 = −12.648025
T (2) = 1.372278
S3,2 = −15.271038
T (3) = 1.803190
S4,3 = −15.271038
T (4) = 1.372278
S5,4 = −12.648025
T (5) = 0.973228
S6,5 = −3.648076
T (6) = 0.136591
Table 5.3: Derived Chebyshev parameters and resonator insertion losses for 35-GHz SIW filter.
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Figure 5.11: Simulated filter insertion loss (solid lines) and return loss (dashed lines). Simulation
performed with different tape tan δ (indicated by different colors).
Parameter
Dimension
Parameter
Dimension
d1
1.041 mm
l1
1.946 mm
d2
0.863 mm
l2
1.778 mm
lstub
1.833 mm
l3
1.821 mm
dvia
120 µm
dvia−via
300 µm
Table 5.4: Dimensions of manufactured filter.
respectively.
5.5.3 Prototype and Measurements
The dimensions of the manufactured filter are given in Table 5.4. The symbols for lengths
and openings are given in Figure 5.9(c), dvia is the diameter of the vias and dvia−via is
the distance between two vias. The filter has been manufactured with 3.6 mil thick
Heraeus HeraLockr HL2000 tapes. Since a minimum of 4 layers was required by the
foundry, an additional layer has been added at the bottom of the LTCC module. Gold
ink is used only at the top, whereas the inner layers are realized with silver ink. This
5.5 NARROWBAND SIW FILTER IN LTCC
71
Figure 5.12: Photograph (top-view) of manufactured SIW filter.
mixed system has several advantages: Wire bonding is simplified by the much harder
gold on the top and the much lower resistivity of the inner layers used silver improves
the filter properties (The gold ink has a resistivity of < 10 mΩ/, and the silver ink has
a resistivity of ≤ 3 mΩ/). Additionally, the filter with mixed metallization is only half
the price of the pure gold realization.
Figure 5.12 shows a photograph of the realization. The microstrip lines can be seen on
the right and left side of the substrate. The SIW wall vias are carried to the top of the
module to indicate the filter size and location. The length of the filter is about 9.2 mm.
Microstrip resonators have been printed at the top layer of the LTCC-module in addition to the filter. The resonators are used for determining the tape loss tangent: Measurements with resonator frequencies between 30 GHz and 35 GHz reveal a loss-tangent
in the range between tan δ = 0.01 and tan δ = 0.015. The dielectric constant is in the
range εr = 7.3 ± 0.1, which is corresponds to the nominal value.
The measured filter insertion loss and return loss are plotted in Figure 5.13 by the
solid and dashed lines, respectively. For comparison, the simulated results with a tape
loss tangent of tan δ = 0.015 are plotted too. The simulated and measured bandwidth
are similar, but the frequency range of the measured filter is about 1 GHz lower than the
simulated filter. A small shrinkage (≤ 5%) in the manufacturing process and tolerances
of via positions (≤ 50µm) are the reason for this frequency shift. Since the measured
change of the dielectric constant is very small, it is very unlikely that the frequency
change is caused by a variation of the dielectric constant.
The nominal resistivity of the silver ink has been taken into account when simulating
the filter. This nominal value of 3 mΩ/ is around 2.5 times higher than the resistivity
of pure silver. Thus, losses caused by the increased resistivity of the metallizations are
taken into account when simulating the filter in Ansoft HFSSTM . The main reason for
the higher losses can be found in the increased loss-tangent. The measured insertion loss
of about S21 = −8 dB is in good agreement with the simulated value when using the
loss-tangent found in the previous measurement (tan δ = 0.015).
The insertion loss of around S21 = −8 dB is much worse than specified for the receiver
(S21 = −2.5 dB). Using this SIW filter in the receiver leads to a degradation of the noise
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Figure 5.13: Filter measured and simulated insertion (dashed lines) and return (solid lines) loss.
figure of 1 dB, when taking both LNAs (the LNA next to the antenna and the LNA at
the receiver input) into account. The noise figure of the receiver with 1 LNA only would
be much worse (around 8 dB), since the gain of the LNA is lower than the losses of the
mixer and filter.
An alternative solution would be the fabrication of the SIW filter on a low-loss LTCCtape. For example the DuPont 943 Low Loss Green TapeTM tape [94] has a loss-tangent
of tan δ = 0.002 at 35 GHz. The previously shown simulations show that this losstangent would lead to the much better insertion loss of S21 = −4 dB. A insertion loss
of S21 = −4 dB instead of S21 = −2.5 dB leads to a degradation of the noise figure
of 0.3 dB. However, the disadvantage of this tape is the higher tolerances in the firing
process.
The results show further that the filter suppresses not only the lower sideband of the
receiver (S21 (25 GHz << −40 dB), but also limits the band to the targeted 3 GHz.
5.6 Noise Figure
5.6.1 Testing Device
A module including the 35-GHz input LNA, the mixer, and the frequency doubler has
been manufactured. In order to demonstrate the noise figure enhancement caused by
the 35-GHz filter, the module has been manufactured in two different versions: The first
version without filter and the other one with the 35-GHz microstrip filter. This module
connected to one of the phase shifters shown in Chapter 4 represents one receiver of the
5.6 NOISE FIGURE
73
Figure 5.14: Photograph of test structure enclosure.
targeted receiver array.
The photograph in Figure 5.14 shows the module without filter in its measurement
fixture. The mixer, frequency doubler and 35-GHz LNA are marked. The LO and RF
ports are realized by 2.92-mm coaxial connectors, all other connectors are from the SMA
type.
Figures 5.15(a) and 5.15(b) show photographs of the modules without and with filter.
In both realizations, the Hittite HMC 263 [75] is used as 35-GHz LNA, the Hittie HMC
329 [72] as mixer, and the Hittite HMC 449 [95] as frequency doubler. The module is
realized on a 10 mil thick Rogers RT/duroidr 6010 substrate.
5.6.2 Measurements
For measuring the noise figure, the mixer is driven by the chip-based injection locked
phase shifter. The HP 8970B Noise Figure Meter [96] has been connected to the LO
port, and the noise source has been connected to the RF port. An isolator has been
added between ENR and RF port.
The dash-dotted and dashed graphs in Figure 5.16 show the measured noise figure
of the receiver without and with filter. The noise figure has been measured for RFfrequencies between 34 GHz and 36 GHz. The measured noise figure of the module
without filter is around 9 dB. This corresponds very well with the value calculated in
Section 5.2. The measured noise figure with filter is – as expected – much lower in
comparison to the structure without filter. However, the value of 5 dB is higher than the
expected value of 2.8 dB.
Due to this higher noise figure, additional experiments have been carried out. The
radiation of a microstrip line on a substrate with high dielectric constant is relatively
high. For this reason, also the losses and the noise figure are increased by the radiation.
To reduce the losses, the microstrip lines have been shielded using a copper foil. Figure
5.15(c) shows a photograph of the shielding structures. The solid line in Figure 5.16
shows the measured noise figure of the module with copper foil. The noise figure improves
by 2 dB.
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(a)
(b)
(c)
Figure 5.15: Photograph of receiver input without (a) and with (b) filter. Shielding of structure
is tested by adding copper-foil tunnel (c).
Table 5.5 gives a summary of the predicted and measured noise figures. It can be seen
that the derived and measured values match very well. The finally measured noise figure
is around 3 dB.
5.7 PROTOTYPE OF RECEIVER WITH BEAMFORMING CAPABILITY
75
Figure 5.16: Measured noise figure of receiver.
Measurement set-up
F [dB] predicted
F [dB] measured
LNA / Mixer
9.2
9
LNA / Filter / Mixer
2.8
5
LNA / Filter / Mixer / Shielding
2.8
3
Table 5.5: Comparison between calculated and measured noise figure.
5.7 Prototype of Receiver with Beamforming Capability
5.7.1 System Layout
To test the receiver and its beamforming capability, two identical receivers have been
manufactured. A schematic is shown in Figure 5.17. One 15-GHz reference oscillator is
used for injection locking the VCOs in both receivers. The reference oscillator signal is
divided by a power splitter and used for injection locking the VCOs in the phase shifters.
The chip-VCO based phase shifters with the 6HP VCOs are used in this implementation. The phase shifters are marked by the dash-dotted rectangles in the schematic. As
explained earlier, the phase shifters output signals are used for driving the mixers.
The implementation has been designed such that every receiver pair (dashed rectangle
in Figure 5.17) is only 10 mm wide. This narrow implementation allows realizing an array
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Figure 5.17: Schematic of realized receivers. Two receivers are placed on realized module
(dashed rectangular). Realized receiver modules include phase shifters (dash-dotted rectangular), but not reference oscillator.
by placing several of those implementations next to each other. A receiver-to-receiver
distance of only 5 mm simplifies connecting the antenna array.
The microstrip lines are realized on a Rogers RT/duroidr 6010 substrate. The bottom
of the substrate is adhesively bonded to a metallic block. Mounting holes for placing
MMICs are milled into the substrate. Furthermore, also empty surfaces on the substrate
are milled out.
Figure 5.18 shows the substrate before placing the MMICs and Figure 5.19 shows
a photograph of the module including all active components. The input and output
ports at the respective top and bottom of the photograph are realized by using compact
HUBER + SUHNERr MMPX connectors [97]. These connectors, which provide excellent
properties up to 60 GHz, are also important for the size-reduction, since other connector
types (such as 2.92-mm coaxial connectors) would required much more space. Employing
the MMPX connectors reduces not only the size of the receiver array, but makes the use
of the system also flexible. Antennas can easily be changed and also the number of
parallel receivers can be adjusted.
Flip chip attenuators can be added at the in Figure 5.19 marked position. Barry industry attenuators [98] have been tested for this purpose. Measurements showed that the
attenuators have reasonable properties up to 20 GHz, although the data sheet specifies a
5.7 PROTOTYPE OF RECEIVER WITH BEAMFORMING CAPABILITY
(a)
77
(b)
(c)
Figure 5.18: Photograph of implementation before placing connectors (a), phase shifters components (b) and mixer (c).
maximal frequency of 4 GHz. Attenuators with different nominal values between 0.5 dB
and 20 dB allow adjusting the reference signal power level for every VCO. Thus, differences between VCO samples can be compensated and the locking range and phase-tuning
range of every receiver can be optimized independently.
The DC lines, required for biasing and controlling the active components, are placed
in cavities within the metallic block below the substrate. The wires can be seen on the
top of the photograph in Figure 5.19. The shown feedthroughs are used for connecting
the wires in the metallic block with the upper side of the substrate. The tops of the
feedthroughs are wire bonded to lines on the substrate, which are connected to the
MMIC bias ports. Chip-capacitors are placed next to all MMICs. On the other end, the
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Figure 5.19: Photograph of two-receiver module.
5.7 PROTOTYPE OF RECEIVER WITH BEAMFORMING CAPABILITY
(a)
79
(b)
(c)
Figure 5.20: Photograph (side-view) of open (a) and closed (b) two-receiver module and photograph of top cap (c).
wires are connected to coaxial cables. The transition from the wire to coaxial cable is
made at the point where all wires exit the metallic block. Thus, also the DC-lines are
shielded over their full length.
Figure 5.20(a) shows the side view of the module. It can be seen that the metallic
block is thick enough to not only provide housing for the DC-wires, but also allow good
dissipation of the heat caused by the MMICs. Figure 5.20(b) shows the side view of the
module with the top cap. The cap is attached to the metallic block in the substrate
cavities and fixed with screws. The photograph of the top cap in Figure 5.20(c) shows
the interface required to attach the cap in the substrate cavities. Figure 5.20(b) shows
further that the vertical MMPX connector is placed in a hole of the top cap.
5.7.2 Receiver Tests
In order to prove the functionality of the used VCO, the free-running frequency is measured. Since the VCO is not accessible, a 35-GHz single-tone signal is used as input
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Figure 5.21: Measured output frequency of receiver with not injection locked VCO. VCO operation point is Vdd = 1.77 V and Vbase = 1.36 V
frequency. The measured output frequency is fIF = 35 GHz − 2 · ffr (Vtune ). The measured output frequency is plotted in Figure 5.21. The graph shows that the tuning
behavior is continuous. The tuning behavior has been measured while increasing and
degreasing the tuning voltage to identify a possible hysteresis. In this measurement,
both curves are identical (plotted as solid and dash-dotted lines). The used operation
point (Vdd = 1.77 V, Vbase = 1.36 V) is similar to the operation point used for phase
tuning later in this section.
Since every phase shifter differs slightly from the others, it is not possible to use the
operation points defined in Chapter 4. The differences in the behavior are caused by
manufacturing tolerances of the T-junction and the bond wires, but also by the variation
of the LNA and VCO properties. For this reason, an optimal operation point for every
phase shifter has to be found. This process is similar to the one in Section 4.2.5. Also
here, the VCO free-running frequency range and the locking gain can be adjusted by
changing the VCO bias voltages and the reference oscillator power level, respectively.
As described in the previous section, attenuation pads can be placed in every channel to allow adjusting the reference power level for every receiver independently. This
attenuators are required when using several modules in an array to adjust the different
required power levels. The chip attenuators are not used in the following experiments,
since only one module with two receivers has been manufactured. The power level of the
reference signal in the measured receiver has been adjusted externally.
The down converter used for the reference signal – which is required for measuring
the phase by using a mixer – has to have the same properties as the measured receiver.
Therefore, both receivers on the manufactured module are required. The first receiver
will be the measurement target, whereas the second acts as a reference receiver. Figure
5.22 shows the schematic of the measurement set-up (Photographs of the set-up can be
found in Appendix A.4). All LNAs and frequency doubler have been biased with 3 V
and 5 V battery voltage. The phase shifters in both receivers are injection locked using
the 15-GHz reference oscillator.
Since the phase detector requires a LO and RF signal with equal frequency, a 35-
5.7 PROTOTYPE OF RECEIVER WITH BEAMFORMING CAPABILITY
81
Figure 5.22: Schematic of measurement set-up used for measuring phase tuning behavior of
receiver.
GHz single-tone signal is used as receiver input signal. The 35-GHz signal is split by
using a 10-dB coupler. The attenuated output is connected to a 35-GHz LNA and to
attenuators. The attenuators are set such that the input power levels of both receivers
are equal. The output ports of both receivers are connected to the phase detector: The
measured receiver (receiver 1) to the RF port of the phase detector and the reference
receiver (receiver 2) to the LO port of the phase detector. A 5-GHz variable phase
shifter is added between reference receiver output and phase detector LO port. This
phase shifter is later used for calibrating the measurement set-up.
The tuning voltage port of the reference receiver VCO can be set to an arbitrary
value within the locking range. For the following measurement a tuning voltage of
Vtune2 = −2.19 V has been applied. This voltage is in the center of the locking range and,
thus, ensures that the VCO in the reference receiver is injection locked during the whole
measurement. The exact operation point is not of interest, since the variable phase shifter
is used for calibrating. However, when taking more than one receiver into operation, the
operation point of the reference receiver has to be same when measuring the different
receivers. The tuning voltage port of the measured receiver VCO is connected to a ramp
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Figure 5.23: Phase detector voltages in four operation points and ramp generator output voltage.
VCO operation point is Vd = 1.79 V and Vbase = 1.36 V.
generator. A 10-Hz ramp signal has been chosen for the experiment. The phase detector
output port is connected to an oscilloscope.
Similar to Section 4.2.5, the measurement set-up is used for finding an operation point
with a wide locking range. An optimal operation point has been found by setting the
VCO operation point to Vdd = 1.79 V and Vbase = 1.36 V and applying a reference oscillator power of −15 dBm. The power level has been adjusted by LNAs and attenuators
and is measured at the attenuator output (marked with a * in Figure 5.22).
The oscilloscope signal in Figure 5.23 has been taken in the optimal operation point
for four settings of the variable phase shifter. Based on this plot, the phase tuning range
can be estimated: The distance between two zeros or two maxima represents an 180◦
phase shift. Since the phase detector output voltage exhibits two zeros within the locking
range, the estimated phase tuning range is larger than 360◦ . The ramp signal is also
plotted in Figure 5.23. It can be seen that this ramp signal is noisy. This noisy signal is
also the reason for the noisy output voltage of the phase detector.
5.7.3 Measured Phase Shift
Since the phase detector has its highest sensitivity for output voltages around Vdetector ≈
0 V, the phase measurement has been performed in this detector operation point. The
5.7 PROTOTYPE OF RECEIVER WITH BEAMFORMING CAPABILITY
83
Figure 5.24: Receiver phase shift vs. VCO tuning voltage. VCO bias: Vdd = 1.79 V and
Vbase = 1.36 V.
ramp generator has been exchanged by a battery with potentiometer and the oscilloscope
has been replaced by a voltmeter. The changed devices are labeled in brackets (Figure
5.22).
Figure 5.24 shows the measured phase tuning behavior. A continuous phase tuning
range of > 360◦ can be observed. The comparisons with the derived tuning ranges
(Chapter 2.2) and the measured phase tuning range of the phase shifters (Chapter 4)
show a good agreement. In both cases, a tuning range of > 180◦ has been shown. This
range is doubled in the receiver by the frequency doubler.
The variation of the detector voltage while keeping the tuning voltage constant is in
the range of 1 mV. This detector voltage variation is equal to a phase variation of 0.1◦ or
less. However, this accuracy can only be achieved with highly constant tuning and bias
voltages. As shown in Section 4.2.5, the bias voltages must not vary more than 1 mV to
achieve a phase accuracy of ±2◦ .
5.7.4 Noise Figure Measurement
The measurement of the receiver noise figure is performed similarly to the measurements
of the module in Section 5.6. The ENR has been connected to the 35-GHz input port.
An isolator has been added between ENR and input port connectors. The 5-GHz output
port is connected with the Agilent N8975A NFA Series Noise Figure Analyzer [99]. The
noise figure analyzer is calibrated with 2.92-mm coaxial connectors. Since no MMPXthrough connecters were available, the K-to-MMPX adapters are not taken into account
when calibrating the measurement set-up. In the measurement set-up, the receiver is
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5 RECEIVER
Figure 5.25: Measured receiver noise figure without (dash-dotted and dashed lines) and with
(solid line) additional amplifier.
biased by batteries. The reference oscillator is used for injection locking the VCO. The
phase shifters are in the injection locked state.
Figure 5.25 shows the measured noise figure values vs. the input frequency. The
dash-dotted line represents the measured noise figure of the receiver. The value around
4 dB is higher as the value around 3 dB measured with the testing devise in Section 5.6.
As mentioned before, the K-to-MMPX adapters were not calibrated out. Therefore the
measured noise figure can be corrected by the losses of the adaptor used for connecting
the input port. The measured loss of this adapter are about 0.8 dB. The dashed line
in Figure 5.25 shows the corrected noise figure. The noise figure value of around 3.2 dB
is still higher as the noise figure of the module, due to additional losses caused by the
MMPX connectors on the receiver and the longer line between the connector and the
35-GHz LNA.
The reason for using the MMPX connectors is the simplified connectivity to an antenna
array. Figure 5.26 shows photographs of an antenna array developed for the receiver
array. The MMPX connectors can be seen on the photographs of the front and the back
side of the antenna array. The distance between two connectors is 5 mm. The front side
photograph shows the microstrip line on a Rogers RT/duroidr . The antenna is realized
on a Al2 O3 substrate, which has significant lower losses than the Rogers RT/duroidr .
The Vivaldi antennas on the back side are coupled with microstrip lines to the front
side. A metallic plate is bonded to the back side of the Rogers RT/duroidr substrate
to simplify the assembly of the LNAs. The additional LNAs are placed close to the
antennas. This LNA compensates the losses of the transmission lines and connectors
used for connecting the antennas with the receiver.
5.7 PROTOTYPE OF RECEIVER WITH BEAMFORMING CAPABILITY
(a)
85
(b)
Figure 5.26: Front (a) and back (b) sides of vivaldi antenna array. LNAs are placed next to
the antennas. Antenna module includes MMPX connectors for simplifying connections with
receiver array.
Figure 5.27: Measured receiver gain without (solid line) and with (dash-dotted line) additional
amplifier.
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5 RECEIVER
Part
Component
Amplifiers
- 35-GHz input LNA
Hittite HMC 263
- 15-GHz buffer LNAs (phase shifter)
Hittite HMC 516
- 5-GHz LNA @ receiver output
Hittite HMC 392
Frequency doubler
Hittite HMC 449
Mixer
Hittite HMC 329
Attenuation pads
Barry Industries
nn dB
AV0405CB-nn00JN-90
Connectors
HUBER + SUHNERr
- horizontal
92 MMPX-S50-0-1/111 NM
- vertical
82 MMPX-S50-0-1/111 NM
Substrate
Rogers RT/duroidr 6010
Table 5.6: Components used in Receiver.
In order to provide noise figure measurements including the additional LNA as well
as the lines and connectors between the LNA and the receiver, the noise figure has been
measured with an additional LNA. The measured noise figure is plotted with the solid
line (Figure 5.25). The measured noise figure of about 2.5 dB is not significantly higher
than the value of 2.2 dB predicted in Section 5.2.
The receiver with and without additional LNA has also been used for measuring the
receiver gain. The receiver is operated in the same operation point as for the noise
figure measurements. The solid and the dash-dotted curves in Figure 5.27 show the
measured gain of the receiver without and with additional amplifier, respectively. The
gain without additional amplifier is about 3 dB. Due to the in comparison to the test
device high losses, the gain is also lower as the value of 5 dB measured with the test
device. Also here, the gain is sufficient due to the usage of the additional LNA. The gain
with the additional LNA is around 21 dB.
Table 5.6 gives a list of the components used in the receiver. A summery table of the
receiver properties can be found in Table 5.7.
5.7 PROTOTYPE OF RECEIVER WITH BEAMFORMING CAPABILITY
Parameter
Value
VCO
- bias voltage Vdd
1.79 V
- base voltage Vbase
1.36 V
LNAs bias voltages
3V
Frequency doubler bias voltage
5V
Input frequency range
Output frequency range
Reference oscillator frequency f0
Gain
32 − 38 GHz
2 − 8 GHz
15 GHz
3 dB
Noise figure
- w/o additional LNA
3 dB
- w/ additional LNA
2.5 dB
Suppression
- input frequency fin (35 GHz)
> 55 dB
- reference oscillator f0 (15 GHz)
> 25 dB
- LO frequency 2 · f0 (30 GHz)
> 30 dB
- 1st harmonic 2 · fout (10 GHz)
> 30 dB
Phase tuning range
400◦
Table 5.7: Measured receiver parameters.
87
88
5 RECEIVER
5.8 Conclusion
The implementation of a receiver with a measured noise figure of only 3 dB has been
demonstrated. The receiver has been designed such that it is only 5 mm wide. This
allows building a receiver array with a receiver-to-receiver distance similar to the distance
between two antennas in an antenna array.
The mixer is driven by the frequency doubled signal of the phase shifter introduced
in Chapter 4. In comparison to the phase tuning range measured in Chapter 4, the
tuning range is doubled due to the employed frequency doubler: A phase tuning range
of 400◦ has been measured in the found operation point. A stability of the phase in the
range of ±0.5◦ has been measured when using batteries for biasing the VCO. It has been
shown that the used VCO is highly sensitive to changes of the bias voltages. Variations
of the bias voltages in the range of ±1 mV lead to phase variations in the range of a
few degrees. This high sensitivity makes an implementation of this phase type into a
commercial product impossible. Using a different oscillator with lower pushing might
solve this issue.
The receiver design has been made such that it can be transferred to a LTCC implementation. The used microstrip filter can be replaced by the demonstrated SIW filter.
The design of such an LTCC-based receiver has been finished. However, it has not been
manufactured due to organizational issues of the foundry. The submitted design is shown
in Appendix B to give an idea how the shown receiver could look like when implemented
on LTCC.
6 Conclusion
6.1 Discussion
The implementation of 35-GHz receivers for radiometry has been discussed in this dissertation. The receiver is based on planar structures and exhibits a noise figure of 3 dB.
The comparison with the targeted value of 4 dB shows that the receiver noise figure is
low enough to be used in a radiometer.
The receiver mixer is driven by the frequency doubled signal of an injection-locked
VCO. The VCO acts as a phase shifter and enables controlling the phase. Implementing
the VCO at half the frequency and frequency doubling has the advantage that the phasetuning range is doubled. Additionally, the VCO implementation at half the frequency
benefits from lower relative tolerances, better component availability and lower costs.
Two different phase shifters, one based on a low-Q chip VCO and one based on an
LNA fed back by a tunable resonator, have been tested. For the chip VCO phase shifter,
an operation point with a wide locking range and a wide phase tuning range has been
found. The measured phase tuning range is 200◦ . The big bias-voltage-to-frequency
dependence allows adjusting the operation point, but requires also highly stable bias
voltages. It has been shown that the bias voltage variation has to be below 1 mV to keep
the phase error in the range of a few degrees. Since unbuffered VCOs are not available
as standard chips, a costumer specific VCO is required. This might increase the price of
the receiver significatively.
The second phase shifter is based on an LNA fed back by a low-Q varactor tuned
resonator. The input and output ports have been implemented to enable injection locking. Input and output amplifier provide a stable load to the VCO tank and buffer the
signals. Pushing is not an issue in this phase shifter type; voltage variations in the range
if ±100 mV have no influence on the phase. The disadvantage of this phase shifter is the
much bigger size. Additionally the tolerances when manufacturing the resonator cause
a notable parameter spreading between the manufactured samples.
The phase shifter with the chip VCO is used in the implementation of two receivers.
One receiver is used for the phase measurements and the other one acts as a phase
reference. These two receivers act as proof of concept for a receiver array. Besides the
measured receiver-to-receiver phase and the low noise figure, also the small receiver width
in the range of the antenna-to-antenna distance of an antenna array is demonstrated.
The full receiver array connected to an antenna array allows for electronic beamforming.
Since the VCO properties of the unbuffered VCOs are highly dependent on the load,
also the VCO properties in the receivers differ. The main reason for differing loads are
manufacturing tolerances. For this reason, every receiver has to be calibrated separately.
Different VCO operation points might be required. Therefore, also different bias voltages
are needed. Considering a relative small array with eight elements, 3 · 8 = 24 highly
constant voltages are required for biasing and controlling the VCOs. In addition to these
89
90
6 CONCLUSION
voltages, 3 V and 5 V sources are needed for LNAs and frequency doublers. This number
shows that the complexity of a receiver array is growing significatively. The complexity –
and also the size – of a large array may exceed the complexity of a waveguide radiometer.
Therefore, the proposed advantages of the receiver are not fulfilled and the usage of the
receiver may become unattractive when increasing the array size.
Building a smaller receiver array decreases the complexity, but using as a radiometer
might not be possible due to the small aperture of the small antenna array. Connecting
a single receiver to a high-gain antenna might work as a radiometer, but phase controllability is not required in this case. Thus, a basic receiver would be sufficient for this
purpose.
6.2 Future Work
The demonstrated implementations of the receivers and phase shifters act as a proof of
concept. The consequent next steps are implementing the receiver array together with
an antenna array. Two different scenarios are possible: The implementation as a hybrid
device utilizing a PCB board on a metallic structure, or the implementation as an LTCC
module. The hybrid device requires intensive mechanical processing, but also the LTCC
implementation results in a complex – and thus expensive – structure. This is especially
due to the fact that the active components required lots of bias lines, heat sinks and
shielding. The measurement of the over-all system (receivers connected to an antenna
array) poses high demands on the measurement system.
The implementation in LTCC allows for a more advanced design. For example, the
shown SIW filter can be used. LTCC might also lead to a higher packaging density.
Since it has been mentioned that not only the receiver size, but also the biasing and
controlling increase the system size, it is questionable if LTCC leads to a notable size
reduction of the radiometer.
Extending the beamforming capability to two dimensions is also possible with the
introduced system. However, the number of injection-locked VCOs is increased significantly. The reference oscillator would be used for injection locking a line of VCOs. Every
VCO in such a system corresponds to a column. In the next level, every of these VCOs
is used for injection locking another line of VCOs, where every VCO corresponds to a
row. A n × m receiver array would require m · n + m VCOs.
A solution for the growing complexity could be integrating the phase shifter. Having
an MMIC VCO with input and output ports would have lower tolerances and smaller
distances between the amplifiers and VCO. Latter leads to smaller spreadings between
the phase shifter samples and also to a smaller frequency dependency. Also pushing
could be minimized in such a VCO design and internal biasing could reduce the number
of required bias lines.
A Photographs of Measurement Set-Ups
A.1 Introduction
A number of measurements have been performed in order to characterize the components
discussed in this dissertation . All measurements are described in the respective chapters
within the dissertation. The photographs in this Appendix should give an impression
about the measurements performed in the laboratory.
A.2 Wiltron Universal Test Fixture
The photograph in Figure A.1 shows a measurement set-up utilizing one of the Wiltron
universal test fixtures [100]. The measured substrate can be seen between the two
metallic parts in the center of the picture. The substrate in the shown measurement is
a test for specifying flip-chip attenuation pads on a Rogers RT/duroidr 6006 substrate.
Figure A.1: Photograph of Wiltron universal test fixture.
91
92
APPENDIX A PHOTOGRAPHS OF MEASUREMENT SET-UPS
Figure A.2: Photograph of measurement showing batteries used for biasing.
A.3 Phase Shifter
The photograph in Figure A.2 shows one of the chip-VCO based phase shifters connected
to batteries. The big red block with 8 round batteries is used for biasing the amplifiers.
These batteries provide sufficient capacity for biasing the amplifiers. Thus, a constant
voltage over the measurement can be guaranteed. Since the VCO has a much lower
power consumption, block batteries are used for the VCO tuning and bias voltages.
An overview photograph of one of the measurement set-ups used for measuring the
phase shifters phase is shown in Figure A.3. The chip-VCO based phase shifter can
be seen in the fixture in the center of the picture. Furthermore, the battery powered
DC-source, which has been used for several measurements can be seen as well. This
battery powered DC-source has been used for finding operation points. Noise and phase
measurements have been performed using batteries, as shown in Figure A.2. The oscilloscope, the ramp generator and the 15-GHz variable phase shifter are marked in Figure
A.3. The visible computer has been used for controlling the devices via HP-IB bus.
The two shown spectrum analyzers are used for checking if wether or not the VCOs are
injection locked.
APPENDIX A PHOTOGRAPHS OF MEASUREMENT SET-UPS
93
Figure A.3: Photograph of measurement set-up for measuring phase tuning behavior of phaseshifter.
94
APPENDIX A PHOTOGRAPHS OF MEASUREMENT SET-UPS
Figure A.4: Photograph of receiver measurement set-up.
A.4 Receiver
The photograph in Figure A.4 shows an overview picture of the measurement set-up used
for measuring the receiver phase-shift. The receiver, DC-unit, ramp generator, an the
oscilloscope are visible in the photograph. The shown 15-GHz generator has been used
instead of the reference oscillator in the first measurements. The two shown spectrum
analyzers have been used for observing if the VCOs are injection locked. Also here,
the DC unit is only used for finding the operation point and replaced by batteries for
measuring the receiver noise and phase.
Figure A.5(a) shows a more detailed photograph of the connected receiver. The photograph has been taken before closing the receiver. Therefore, the inside of the receiver
but not the top cap is visible. Figure A.5(b) shows a detailed photograph of the 35-GHz
variable phase shifter, the mixer, and the voltmeter.
Although both, the voltmeter and the oscilloscope are visible in the pictures, only one
device is used at the time. The voltmeter is used in conjunction with a battery for the
phase measurement, whereas the oscilloscope is used with the ramp generator for finding
the operation point.
APPENDIX A PHOTOGRAPHS OF MEASUREMENT SET-UPS
95
(a)
(b)
Figure A.5: Photograph of receiver in measurement fixture connected to measurement cables
(a) and photograph employed 35-GHz variable phase shifter with mixer and voltmeter (b).
Receiver shown without top cap.
96
B LTTC Receiver Design
B.1 Introduction
A design of an LTCC-based two-receiver system has been completed and sent to the
foundry. This two-receiver system is similar to the hybrid implemented system presented
in the dissertation, but includes the SIW 35-GHz input filter instead of the microstrip
filter. Unfortunately, it has not been manufactured due to organizational and financial
constraints.
Although the design has not been manufactured, it is shown in the following to give
an idea how such an LTCC-based receiver could look like.
B.2 Design
The LTCC-design is based on five different levels having two or four LTCC layers per
level. Every layer is 3.6 mil thick. Using this thin layers was a requirement of the foundry.
The sketch in Figure B.1 shows a cross section of the LTCC-module. Transmission lines
and components are placed to illustrate the most important parts of the design process.
In this figure, every level is indicated by a different color. The lowest level consists of
two layers. The bottom of layer 0 and top of layer 1 is metalized. The lines realized
between layers 1 and 2 are used for distributing the DC signals.
All microstrip and stiplines are realized in levels 1, 2 and 3. Each of these levels
consists of 4 layers, the bottoms of the lowest and the tops of the highest layers are
metalized. The microstrip and striplines are realized between the 2nd and 3rd layer of
each level. As indicated in the drawing, all lines are shielded by via fences. The fences are
implemented using a distances of 0.22 mm between the vias with a diameter of 0.12 mm.
This is required to avoid resonances within the module.
Active components are placed in cavities. The 35-GHz LNAs, the mixers and the
frequency doublers are placed in level 2 cavities, whereas the 15-GHz LNAs and the
VCO are placed in level 3 cavities. The cavity depth is such that the top of the MMICs
is on the same height as the top of layer 7 (for level 2 components) or layer 11 (for level
3 components). This equal height allows connecting the MMICs and the microstrip lines
with the shortest possible bond wire length. The bottom of the MMICs is connected to
the bottom of the LTTC module by 0.25 mm wide thermal vias. An external heat sink
has than to be added at the bottom of the module.
The 35-GHz SIW filter, discussed in Chapter 5.5, is implemented in layers 5 and 6.
The microstrip lines used for connecting the filter are based on a different system as
the other lines. This microstrip lines have only a 1 · 3.6 mil mil thick tape between the
ground and the line, whereas all other lines are realized with two layers. A transition
between the two systems is not required, since the microstrip lines of both systems are
97
98
APPENDIX B LTTC RECEIVER DESIGN
Figure B.1: Cross-section of LTCC module showing implemented levels (different colors). Placed
active components and vias are shown as well.
carried to two opposing edges of a cavity. The LNA in the cavity is than bonded to the
two different 50-Ω lines at the respective input and output ports of the LNA.
The cavities are kept as small as possible. However, the size has to be chosen such
APPENDIX B LTTC RECEIVER DESIGN
99
(a)
(b)
Figure B.2: Microstrip-to-stripline (a) and layer (b) transitions.
that bonding is possible. A microstrip-to-stipline transition is required at the cavity
walls. The transitions are marked in Figure B.1; Figure B.2(a) shows a 3D-plot of
such a transition. The lower two layers of the level are plotted as solid block (solid
lines), whereas the upper two layers are plotted as transparent block (dashed lines).
The transmission lines in both sections are plotted in gray. The wider line on the left
side of the plot represents the microstrip line, whereas the narrow line on the right
side represents the stripline. The vias including their catch pads can be seen in the
stripline section. Es mentioned before, this via fences are required to reduce radiation
of the transmission lines. Although radiation of microstrip lines is often considered as
100
APPENDIX B LTTC RECEIVER DESIGN
unproblematic, also the microstrip lines have been shielded in this design. The catch
pads of the vias can be seen in the microstrip section. Since the foundry design guidelines
require a distance of 0.3 mm between the cavity wall and the via edge, it was not possible
to put a metallization close to the cavity wall. Due to this limitation, two vias have
been set close to the stripline in order to improve the transmission characteristic of the
transmission. 3D field simulations show that the optimum of the gap between the vias
is 0.7 mm.
In order to allow transmission lines in different levels, layer transitions are required.
Figure B.2(b) shows a 3D-plot of such a transition. The stripline in level 2 can be
seen in the left part of the drawing, whereas the stripline in level 1 can be seen in
the metallization hole between the two levels. The two striplines are connected using
a via, which is carried through the hole in the metallization. The connecting via goes
through the upper two layers of level 1 and the lower two layers of level 2. Catch pads
required at the layer edges are not shown for the fence vias. In addition to the fence on
both sides of the line, a fence is also placed at the ending side of the transmission line.
This measure improves the transmission characteristic of the transitions significantly.
The dimensions of the transitions have also been optimized in Ansoft HFSSTM . The
transition is implemented in two different versions: straight and with a 90◦ bend.
The cross-section in Figure B.1 shows further that level 4 has no transmission lines.
The 4 layers in level 4 are required to increase the modules hight. This is necessary, since
the cavities can be closed on top of level 4 by using a metallic structure. This structure
provides than also shielding to all microstrip lines and MMICs.
Pads for connecting the DC-sources are placed on top of level 4. From here, the DC
signals are carried by long vias to level 0. Vias are then used again for connecting the
MMIC bias ports with the DC-lines. The reference oscillator input port, the 35-GHz
input port and the 5-GHz output port are realized as microstrip lines on top of layer7.
Cavities have been implemented to make this layer accessible. The LTCC-module is
designed such that in can be put into a motherboard which has the same height as layer
7. Thus, microstrip lines can be connected to the motherboard by using bond wires or
bond bands.
All layer transitions which were considered as metalized have not a full metallizations.
To increase the thermal stability, 0.5 mm wide metallic strips with a distance of 0.5 mm
are used as metallization. The strips are placed in both directions such that they built
a grid. Although such a grid provides an excellent ground for the targeted frequency
range, a permanent metallization has been put at crucial points. Such points are for
example below microstrip lines, at via transitions, etc.
Figure B.3 shows the top-view of all lines and components. Vias, ground planes,
and transitions are not plotted in order to make the design demonstrative. The design
is similar to the one realized on Rogers RT/duroidr , but the implementation of the
transmission lines is simplified by the possible 3D-integration in LTCC. This results in
the reduced length of only 50 mm. Similar to the receiver on Rogers RT/duroidr , a
receiver-to-receiver distance of 5 mm has been achieved.
APPENDIX B LTTC RECEIVER DESIGN
101
Figure B.3: Top-view of striplines and microstrip lines as well as placed active components in
LTCC design. Different colors demonstrate different transmission line and component levels.
102
APPENDIX B LTTC RECEIVER DESIGN
B.3 Complexity
Via fences and transitions, make the design complex. Measures, such as minimum distances between vias in different levels, distances to via walls or distances to the substrate
edges, which are required to comply with the foundry’s design guidelines increase the
number of elements further.
The demonstrated design has around 43.000 objects. Most of the objects are placed
manually during the design process.
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List of Publications
Journal Papers
P1
H. Grubinger, H. Barth, and R. Vahldieck, “An active low-noise receiver with
electronic beamforming capability at Ka-band frequencies,” IEEE Trans. Microwave Theory and Techn., vol. 56, no. 5, pp. 1013– 1023, May 2008.
Conference Papers
P2
H. Grubinger, G. von Büren, H. Barth, and R. Vahldieck, “Continuous tunable
phase shifter based on injection locked local oscillators at 30 GHz,” IEEE MTTS Int. Microwave Symp. Dig., San Francisco, USA, pp. 1821-1824, June 2006.
P3
H. Grubinger, H. Barth, and R. Vahldieck, “An active electronic Ka-band antenna beam-forming network based on injection-locked local oscillators,” IEEE
MTT-S Int. Microwave Symp. Dig., Honolulu, USA, pp. 1875–1878, October
2007.
P41
——, “A low-noise front-end with beam-steering capability at 35 GHz,” European Microwave Conference, Munich, Germany, pp. 1594-1697, October 2007.
——,“A low-noise front-end with beam-steering capability at 35 GHz,” European
Radar Conference, Munich, Germany pp. 315-318, October 2007.
P52
H. Grubinger, B. Hofer, H. Barth, and R. Vahldieck, “A voltage-controlled
oscillator with injection-locking input for phase-shifting applications at 30 GHz,”
European Microwave Conference, Amsterdam, The Netherlands, pp. 1731–1734,
October 2008.
——, “A voltage-controlled oscillator with injection-locking input for phaseshifting applications at 30 GHz,” European Wireless Technology Conference,
Amsterdam, The Netherlands, pp. 310–313, October 2008.
1 This
paper has been presented in a joint session of the 2007 European Microwave Conference and the
2007 European Radar Conference. Therefore, the paper is published in both conference proceedings.
2 This paper has been presented in a joint session of the 2008 European Microwave Conference and
the 2008 European Wireless Technology Conference. Therefore, the paper is published in both
conference proceedings.
111
112
LIST OF PUBLICATIONS
P6
H. Grubinger, H. Barth, and R. Vahldieck, “A low-loss, wideband combiner
for power amplification at Ka-Band frequencies,” IEEE MTT-S Int. Microwave
Symp. Dig., Atlanta, USA, pp. 1139-1142, June 2008.
P7
——,“An LTCC-based 35-GHz substrate-integrated-waveguide bandpass filter,”
IEEE MTT-S Int. Microwave Symp. Dig., Boston, USA, pp. 1605-1608, June
2009.
P8
D. Marti, A. R. Alt, H. Sun, H. Grubinger, H. Benedickter, and C. R. Bolognesi,
“Wideband distributed amplifiers in a hybrid microstrip-environment using 0.1
um (Al,Aa)N/GaN HEMTs grown on silicon,” European Microwave Integrated
Circuits Conference, Rome, Italy, pp. 93–96, October 2009.
P9
——, “Hybrid distributed amplifiers with deep-submicrometer AlGaN/GaN
HEMTs on silicon,” 33rd Workshop on Compound Semiconductor Devices and
Intergrated Circuits, Malaga, Spain, pp. 14-17, May 2009.
Technical Reports
P10 H. Grubinger, R. Vahldieck, and H. Barth, “Feasibility study: 35 GHz radiometer,” ETH Zurich, Laboratory for Electromagnetic Fields and Microwave Electronics, Gloriastrasse 35, 8092 Zurich, Switzerland, Tech. Rep., October 2004.
P11 H. Grubinger, G. Tudosie, R. Vahldieck, and H. Barth, “Zwischenbericht: Design, Simulation und Bau eines 35 GHz Radiometers mit elektronischer Antennenstrahlsteuerung in LTCC Technologie,” ETH Zurich, Laboratory for Electromagnetic Fields and Microwave Electronics, Gloriastrasse 35, 8092 Zurich,
Switzerland, Tech. Rep., January 2006.
P12 H. Grubinger, G. Tudosie, H. Barth, and R. Vahldieck, “Zwischenbericht:
Phased Array Antennen: Demonstration eines 35 GHz Radiometers in LTCC
Technologie,” ETH Zurich, Laboratory for Electromagnetic Fields and Microwave Electronics, Gloriastrasse 35, 8092 Zurich, Switzerland, Tech. Rep., July
2007.
P13 ——, “Abschlussbericht: Phased Array Antennen: Demonstration eines 35 GHz
Radiometers in LTCC Technologie,” ETH Zurich, Laboratory for Electromagnetic Fields and Microwave Electronics, Gloriastrasse 35, 8092 Zurich, Switzerland, Tech. Rep., January 2008.
Master’s Thesis
P14 H. Grubinger, “Entwurf und Modellierung von magnetischen Wänden mit Hilfe
periodischer Strukturenn in LTCC-Technologie,” Master’s thesis, Technische
Universität München, Germany, 2003.
Curriculum Vitae
Personal data
Name:
Citizenship:
Austria
Date of birth:
September 2, 1978
E-mail:
grubinger@ifh.ee.ethz.ch
Hannes Grubinger
Professional experience
03/04 – present: ETH Zürich, Zurich, Switzerland
Laboratory for Electromagnetic Fields and Microwave Electronics
Teaching and Research Assistent
06/99 – 02/04: NIKA(former E&L Wirtschaftstreuhand), Salzburg, Austria
System Administrator
Project Manager and Developer for Inhouse-Software Tools
10/00 – 05/03: Technische Univerität München, Munich, Germany
Institute for Data Processing: Teaching Assistant
Institute for Measurement Systems and Sensor Technology: Teaching
Assistent
06/97 – 09/98: Sony DADC Austria, Thalgau, Austria
Assembly Engineering: Internship
07/96 – 08/96: Salzburger Stadtwerke AG, Salzburg, Austria
Technicall Department: Internship
Education
03/04 – present: ETH Zürich, Zurich, Switzerland
Laboratory for Electromagnetic Fields and Microwave Electronics
Doctorate in Electrical Engineering
11/99 – 11/03: Technische Universität München, Munich, Germany
B.Sc. and Dipl.-Ing. in Electrical Engineering
09/93 – 06/99: Secondary College for Electronics (HTL) Salzburg, Salzburg,
Austria
High School and Engineering Graduation
113