2013 First International Conference on Artificial Intelligence, Modelling & Simulation
Conversion of UHF Composite Low Pass Filter Into Microstrip Line Form
Liew Hui Fang
Syed Idris Syed Hassan
Mohd Fareq Abd. Malek
School of Microelectronic
Engineering
Universiti Malaysia Perlis
Pauh Putra Campus
02600 Arau, Perlis
MALAYSIA
Email:alicefang88@yahoo.com
School of Electrical System
Engineering
Universiti Malaysia Perlis
Pauh Putra Campus
02600 Arau, Perlis
MALAYSIA
Email: syedidris@unimap.edu.my
School of Electrical System
Engineering
Universiti Malaysia Perlis
Pauh Putra Campus
02600 Arau, Perlis
MALAYSIA
Email: mfareq@unimap.edu.my
Yufridin Wahab
Norshafinash Saudin
School of Microelectronic Engineering
Universiti Malaysia Perlis
Pauh Putra Campus
02600 Arau, Perlis
MALAYSIA
Email: yufridin@unimap.edu.my
School of Electrical System Engineering
Universiti Malaysia Perlis
Pauh Putra Campus
02600 Arau, Perlis
MALAYSIA
Email:norshafinash@unimap.edu.my
good selectivity near the passband since they have no
attenuation poles [2,9]. So, both types of low-pass filters
always have the problem of converting the lumped circuit
prototype into microstrip when the order number, N,
increases, making the circuit larger or more complex.
Elliptic–function filters have attenuation poles near their
passband, making them attractive for highly-selective
applications [2,4]. The disadvantages of elliptic design
and implantation are very complicated, and have a ripple
at both in passband and stopband section as well. And the
passband elliptic filter consists of highly non-linear
response especially near with band-edge [2,8-9].
The composite low-pass filter is less complex and
having a sharp roll –off. It was designed by applying the
image parameter method [1-2]. The image parameter [6]
was initiated by defining the image impendence and
voltage function for arbitrary reciprocals of a two-port
network because these designed results are required for
the cutoff frequency and attenuation characteristics.
During the design of the composite, low-pass filter, two
of the important factors that must be taken into
consideration are the constant-k filter section and the mderived section.
Ashwani Kumar etal [3,4] designed a microstrip line
composite filter using the defected ground structure(DGS)
method .Its shunt connected series LC circuits are
transformed with either quarter-wave short circuiting
stubs or quarter-wave open circuiting stub[4,7]. The
performance of DGS composite filter was verified by
comparing lumped elements, microstrip line and DGS
measurement results. Overall, the result of DGS based
Abstract— This paper presents the design of a compact,
composite, low-pass filter circuit into microstrip line form
using a new, transforming method. The composite, low-pass
filter operating in the UHF range were designed and
implemented on an FR4 substrate. The circuits were
simulated and developed using Advanced Design Software
(ADS) for both lumped element and microstrip filters. A
correction factor was considered due to fringing inductance
and capacitance. The ADS simulation results showed that
the response of the microstrip line circuit of the composite,
low-pass filter with fringing correction factor was well
agreement with its lumped circuit. This showed that the new
transforming method enabled to the lumped element circuit
into a microstrip line to solve the complex design of
composite filters.
Keywords — microstrip line filter; constant-k filter; mderived; microwave communication; composite low pass filter
I.
INTRODUCTION
Microstrip filters always find an important place in
many RF microwave applications. They are most widely
preferred for selecting or confining the microwave signals
within specified spectral ranges. The challenges on the
microwave filters with requirements such as improved
performance, miniature size, lighter weight, and lower
cost are ever increasing with the emerging applications of
wireless communications.
When the order of the filter increases, the method of
calculating the dimensions becomes complicated, and it is
adequate to specify that the response occurs at minimum
stopband and passband attenuation. The most Butterworth
and chebyshev require a high-order design to ensure a
978-1-4799-3251-1/13 $31.00 © 2013 IEEE
DOI 10.1109/AIMS.2013.79
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low pass filter achieved good stability and more shaper
cut off response than that of microstrip line and the DGS.
It is also having large rejection bandwidth. The
performances of composite filter are improved by using
defected ground structure. But the disadvantages of DGS
is complex circuitry, high power consumption and image
frequency problems.
Stephane Pinel etal [5] state the compact planar and
vialess composite filter are designed by using the image
parameter method and semiconductor component
approaches which operate at C-V band. The lumped
element vialess composite filter are fabricated by using
liquid crystal polymer substrate, which consists of
characteristic low cost solution RF, high performance,
ultra compact, and millimeter wave application. The
overall folded layout of composite filter occupies an ultra
compact area and optimized by using full wave simulation
IE3D. The combination of stepped impedance filter and
folded stepped impedance resonator performed by lumped
elements schematic filter. And the measurement result
exhibit rejection of attenuation pole which is greater than
-40dB.The design was only present lumped element at
final layout optimization. Fine tune was performed for
the overall structure in order to miniaturize the circuit and
to avoid the impact of excessive stub length [5].
Mostly all works showed that image parameter
method using in designing of lumped elements of
composite filter have not been mentioned clearly the ways
of transforming the circuit into mictrosip line. So, a new
approach of transforming lumped circuit into microtsrip
line is presented correction factor due to fringing is
introducing so that accurate dimension can be determined
without changing the properties of the composite filter.
When a new simple and direct approach method is
applied, it can solve the combination complexity of 4
important sections, that is constant-k, matching section,
m-drive and bisected- π section, of transforming lumped
elements.
Matching
section
High-f
cutoff
m=0.6
Zo
1
2
constant
k
T
Matching
section
Sharp
cutoff
ZiT
ZiT
m=0.6
mderived
m<0.6
1
2
Zo
ZiT
Figure. 1. Block diagram of circuit components in the composite filter
[1, 6]
A. Constant-k T-section
The nominal characteristic impedance of constant–k
section is made a constant value for the assigned
frequency, which is given in [1, 2, 3].
The values of L and C for constant K can be calculated
by using the following formula.
L 2Z o / c
(1)
C 2 / Zoc
(2)
L/2
L/2
C
Figure 2. Low-pass, constant-k filter section in T-network [2]
An m-derived, low-pass, T-section is shown in Figure 3.
mL/2
mL/2
mC
1 m2
L
4m
II. T HEORY OF COMPOSITE LOW -PASS F ILTER
DESIGN
Figure 3. m-derived T-section [2]
The inductance and capacitance values can be calculated
using [1, 2].
C" mC
(3)
The design of composite filter involved the input and
output impedance fixed as 50 ohm, and the required
cutoff frequency response sets as 2.5GHz. The
development of composite low pass filter consideration
the condition is compulsory to combining the constant-K
in cascade and m-derived sharp roll off and matching
section at input and output. Figure1show that the
important section combination of network constituted in
composite filter circuit.
L' 1 m2
L
4m
(4)
mL
2
(5)
Series component
L" where L and C have the same values as the k-constant
section.
392
386
B. Matching Section
By combining in cascade, the constant–k section, the
m-derived of sharp-cutoff section, and the m-derived
matching section, we can produce a filter with the desired
attenuation and matching properties. The sharp-cutoff
section with m < 0.6 places an attenuation pole near the
cutoff frequency to provide a sharp attenuation reaponse,
and the constant-k section provides the high attenuation
further into the stopbands. The bisected π- section with
m=0.6 are palced at the ends of the filter to match the
norminal source and load impendance, Z o, to the internal
, of the constant-K section and the
image impendance,
m-derived section.The matching networks are using the m
= 0.6 bisected –π section, as shown in Figure 4 [1-2].
mL/2
mL/2
mC/2
Zo
Figure 6. Model for series inductor with fringing capacitors
Similarly the capacitance, C with fringing its inductance
is modeled as a T-network as shown in Fig. 7
mC/2
1 m L
Zo
1 m L
2
2
2m
2m
Figure 7. Model for shunt capacitor with fringing inductors
For inductance, L, the length of the microstrip with
characteristic impedance ZOL = 100 ohm can be calculated
using Equation (6):
ZiT
Figure 4. Bisected π- matching section [2]
dL III. MICROSTRIP LINE DESIGN TECHNIQUES
The microstrip inductor and capacitor always
produce fringing, which must be taken into account and
must be corrected. Four conditions have been studied i.e.,
1) the filter is converted directly without correction, 2) the
resonance LC circuit is achieved with a quarter wave stub
short to ground, and 3) the resonance LC circuit is
achieved with a quarter wave stub without ground.4) the
filter is converted directly with correction The typical
composite filter in lumped components is shown in Fig 5.
L d
sin 1 2
Z oL (6)
And its fringing capacitor can be calculated as:
C fL d 1
tan Z oL d (7)
For capacitor, C, the length of the microstrip with
characteristic impedance ZOC = 20 ohm can be calculated
using Equation (8):
dC d
sin 1 CZ oC 2
(8)
And the fringing inductance can be calculated as;
d
tan L
d
L fC Z oC
d c
(9)
where
f r
and
Figure 5. Schematic diagram of composite filters
d
To convert the filter into a microstrip line, first
the inductance L with its fringing capacitor is modeled as
a π-network, as shown in Figure.6
= wavelength
С =velocity of light -3.0e8
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387
(10)
= dielectric constants
dC
C4 n C4 -
= length of fringing capacitance
dL = length of fringing inductance
C fL = fringing capacitance
C fl 3 C fL 5 C fL 6
2
-
2
-
2
(22)
Thus, the circuit with the new values is shown in Figure
9. The lengths of the microstrips for the inductor and the
capacitor were calculated using Equations (6) and (8),
respectively, based on these new values.
L fC = fringing inductance
The width of the microstrip line for the capacitor
and inductor was calculated using the following formula
(approximation):
377
Zo (11)
wn
r 1.57 h
377
wn 1.57 h
Z r
(12)
where Wn refers to W100 , W50 , W20, and Z refers to ZoL,
Zo, and ZoC. where:
Lfc1 = fringing inductance due to capacitor C1
Lfc2 = fringing inductance due to capacitor C2
CfL1 = fringing capacitance due to inductor L1
CfL2 = fringing capacitance due to inductor L2
CfL3 = fringing capacitance due to inductor L3
Figure 8. Composite, low-pass filters after correction due to fringing
The complete microstrip line circuit design of the
composite filter using ADS without considers grounding
on stub is shown in Figure 10.
By considering fringing, the new value of L1, L2, L3 , L4 ,
L5 , L6 , C1, C2 C3,C4 are:
L fC1
L1n L1 -
L2n L2 -
2
-
L fC 2
2
(13)
L fC 2 L fC 3
-
2
2
L fC 3 L fC 4
(14)
2
2
L fC1 L fC 2
(15)
2
2
L fC 2 L fC 3 L fC 4
(16)
2
2
L fC 3 L fC 4
(17)
L3n L3 L4 n L4 L5n L5 -
L6 n L6 C1n C1 -
-
-
-
-
Figure 9. Composite low pass filter in microstrip line filter without
grounding at circle part
2
-
2
2
C fL1 C fL 4
C2 n C2 C3n C3 -
The complete microstrip line circuit design of the
composite filter using ADS by considering grounding
stub is shown in Figure 11 as all simulation result analysis
by using ADS include lumped elements and microstrip
line composite filter.
(18)
-
2
2
C fL1 C fL 2 C fL 5
(19)
2
2
2
C fL 2 C fl 3 C fL 5 C fL 6
(20)
-
2
-
IV. ADS RESULT AND SIMULATIONS
To verify whether this approach is satisfactory or not,
we simulated all four options of composite, low-pass
filters. One is without correction factor with grounding
and another one is without grounding with the cut-off
frequency was set at 2.5 GHz on a substrate that had a
-
2
-
2
-
2
(21)
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388
dieletric constant of 4.5 and a thickness of 1.5 mm; the
second was the microstrip with considering the fringing
correction factor for grounded stub and without ground.
All the results are given in Figure 11a to Figure 11e
below.
m1
freq=1.073GHz
dB(S_50(1,1))=-12.676
m2
m2
freq=500.0MHz
dB(S_50(2,1))=-0.133
m3
0
m3
freq=1.840GHz
dB(S_50(2,1))=-2.930
dB(S_50(1,1))
dB(S_50(2,1))
m1
-20
m4
freq=6.003GHz
dB(S_50(2,1))=-40.552
m4
-40
-60
0
1
2
3
4
5
6
7
8
freq, GHz
(c). Microstripline filter without correction but grounded stub
m1
freq= 1.056GHz
dB(S_50(1,1))=-20.017
m3
m2
dB(S_50(1,1))
dB(S_50(2,1))
0
m2
freq= 500.0MHz
dB(S_50(2,1))=-0.091
m3
freq= 2.219GHz
dB(S_50(2,1))=-3.071
-10
m1
-20
m4
m4
freq= 6.009GHz
dB(S_50(2,1))=-29.056
-30
-40
-50
0
1
2
3
4
5
6
7
8
freq, GHz
(d). Microstripline filter with correction but without grounded stub
Figure 10. Composite low pass filter in microstrip line filter with
grounding at circle part
m2
m1
0
m2
freq=500.0MHz
dB(S(2,1))=-6.260E-5
m3
freq=2.214GHz
dB(S_50(2,1))=-3.021
m1
-20
m4
freq=6.002GHz
dB(S_50(2,1))=-28.730
m4
-30
-40
-50
0
1
2
3
4
5
6
7
8
freq, GHz
m3
freq=2.472GHz
dB(S(2,1))=-3.123
-50
dB(S(1,1))
dB(S(2,1))
0
m2
freq=500.0MHz
dB(S_50(2,1))=-0.091
-10
dB(S_50(1,1))
dB(S_50(2,1))
m1
freq=1.405GHz
dB(S(1,1))=-31.712
m3
m4
m1
freq=1.040GHz
dB(S_50(1,1))=-20.282
m3
m2
(e). Microstripline filter with correction and with grounded stub
-100
m4
freq=6.000GHz
dB(S(2,1))=-39.799
-150
Figure 11 Simulation results of ADS
-200
-250
0
1
2
3
4
5
6
7
V. ANALYSIS AND DISCUSSION
8
freq, GHz
Overall Comparison between all the results are given
in Table 1. The table show that the simulation results of
circuit have good matching where the return loss is below
-20dB and the 2fc attenuation frequency is seemed good
where the amplitude is fall below to -40dB. This means
the attenuation are good enough to suppress the unwanted
frequency signal.
(a) The comparison of the lumped-element and microstrip
line circuits without fringing method (without grounding)
showed that the microstrip line circuit without fringing
method (without grounding) for S11 return loss and -3 dB
cut-off point were farther away from the design frequency
and that, for S21, the insertion loss and the 2fc attenuation
point were close to the lumped-element values.
(b) The comparison between the lumped-element and the
microstrip line circuit without fringing method (with
grounding) showed that the microstrip line circuit without
fringing method (with grounding) had a return loss of for
S11 and a -3 dB cut-off point that were farther away from
(a).Lumped circuit filter
m2
freq= 500.0MHz
dB(S_50(2,1))=-0.133
m1
freq= 1.021GHz
dB(S_50(1,1))=-13.464
m2
m3
0
m3
freq= 1.845GHz
dB(S_50(2,1))=-3.009
dB(S_50(1,1))
dB(S_50(2,1))
m1
-20
m4
freq= 6.008GHz
dB(S_50(2,1))=-41.073
m4
-40
-60
0
1
2
3
4
5
6
7
8
freq, GHz
(b). Microstriple filter without correction and without grounded stub
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389
the design frequency and that, for S21, the insertion loss
and the 2fc attenuation point were close to the lumpedelement values.
(c)The comparison between lumped element and
microstrip line circuit with fringing method (without
grounding), the result show that microstrip line circuit
with fringing method (with grounding), for S11 point are
close to lumped element compare to design frequency but
getting more better than compare to microstrip line
without fringing method, and for insertion loss, -3dB
cutoff and 2fc attenuation point are near with lumped
element value.
(d)The comparison between lumped element and
microstrip line circuit with fringing method (with
grounding), the result show that microstrip line circuit
without fringing method (with grounding), for S11 point
are close to lumped element compare to design frequency
but getting more better than compare to microstrip line
without fringing method,, and for insertion loss, -3dB
cutoff and 2fc attenuation point are near with lumped
element value.
line, the parameter S11 return loss frequency was close to
that of the lumped-element approach with the microstrip
line without fringing method. Further work will be done to
fabricate the filters using micro-electro mechanical system
(MEMS) technology. The new approach is applicable for
solving complex circuits for such composite filters, but it
also can be applied for other types of low-pass filters, such
as Butterworth, Chebyshev, and elliptical, low-pass filters.
ACKNOWLEDGMENT
The authors acknowledge University Malaysia
Perlis and the Malaysian Ministry of Higher Education
for providing the Fundamental Research Grant Scheme
(FRGS Grant No: 9011-00011), which made it possible to
conduct and publish this research.
REFERENCES
[1]
[2]
TABLE I.
SIMULATION RESULT OF COMPOSITE FILTER IN ADS
Param
Lumped
element
circuit
Microstrip
line
without
fringing
(with
grounding)
-12.67dB
Microstrip
line with
fringing
(without
grounding)
Microstrip
line with
fringing
(with
grounding)
-31.71dB
Microstrip
line
without
fringing
(without
grounding)
-13.46dB
S11-return
loss-20dB
S21insertion
loss0.5GHz
Cut off
Freq
-3dB
S21-2fc
attenuatio
n-6GHz
-20.02dB
-20.28dB
0 dB
-0.13dB
-0.13dB
-0.09dB
-0.09dB
2.47GHz
1.84GHz
1.84GHz
2.22GHz
2.21GHz
-39.80dB
-41.16dB
-40.55dB
-29.06dB
-28.73dB
[3]
[4]
[5]
[6]
VI. CONCLUSIONS
[7]
Overall, the simulation results showed that the
composite filter with fringing taking into consideration
without grounding give values closer to the designed
prototype than the filter without taking fringing into
consideration. The design of the -3 dB cut-off frequency of
the lumped-element values fell at 2.472 GHz. The
simulation results of the lumped-element and microstrip
filters were in good agreement with each other. However,
the microstrip filter with the new approach with
fringing taking into consideration without grounding with a
longer microtsrip line cut-off point was very close to 2.5
GHz, and the parameter S21 insertion loss was close to the
value of the lumped-element approach. But, for the
microstrip filter with the new approach with
fringing taking into consideration with a longer microstrip
[8]
[9]
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