DC-DC High Power Converter
Gustavo Lambert, UDESC, Brazil, lambert.g.l@ieee.org
Yales R. de Novaes, UDESC, Brazil, novaes@ieee.org
Marcelo L. Heldwein, UFSC, Brazil, heldwein@inep.ufsc.br
Abstract
This paper presents a multilevel multipulse medium frequency-isolated dc-dc converter. It is
based on the Modular Multilevel Converter (MMC) associated to an isolated Multipulse Rectifier
(MR). Its application is targeted to non-conventional distribution and sub-transmission systems,
while operating at tens of kVs and a few MWs. The main characteristics of the converter structure operation such as harmonic cancellation and transformer voltage pulses are presented.
Simulation and experimental results of a downsized prototype are presented to demonstrate
the converter operation.
1.
Introduction
Over a hundred years ago, in the beginning of the XIX century, names like Thomas Alva Edison
and George Westinghouse Jr. clashed into an episode later known as the “War of currents”.
During this time Edison was in favor of the direct current (dc) while Westinghouse bet into the
possibilities of the alternating current (ac). This episode had it withdraw on October 17th 1893,
when Westinghouse’s company won the contract for the Niagara Falls distribution system [1].
Since then the world’s power system has grown mostly around the ac configurations. According
to the available technology it was easier and more efficient than dc to change voltage levels, it
could transmit energy over longer distances and also, with the development of the ac motor, it
found a number of applications. However, the world has also changed the way it uses electrical
energy. It is known that around thirty percent of all the electrical energy generated is in some
way processed by electronic means before it is finally used. Moreover it is said that this percentage will be substantially grown in the next ten to fifteen years [2]. Transmission systems
in High Voltage Direct Current (HVDC) using Line-Commuted Converter (LCC) or even Voltage
Source Converter (VSC) technologies are well established but high power dc-dc solutions for
high or medium voltage still have few known alternatives [3, 4].
The MMC was introduced in 2002 by Marquardt [5]. In the last few years the MMC has attracted
a lot of attention from researchers for application in medium/high power energy conversion
systems around the world. This happened especially because of the MMC advantages like
modularity, scalability, the possibility of handling medium or high voltage levels processed by
low-voltage semiconductors, possible implementation of redundancy and others [6]. However
the MMC has drawbacks as the need to balance the submodule capacitors [7], the MMC control
is not straight forward [8] and it uses a high number of capacitors, power semiconductors and
communication lines.
The proposed structure for dc-dc conversion is based on the connection between the ac links of
two converters. This method can be galvanically isolated or not. An example of a non-isolated
dc-dc bidirectional conversion using two MMCs is presented by Luth [9]. Another bidirectional
alternative using two MMCs connected through a medium frequency transformer is presented
by Kenzelmann [10].
(a)
(b)
Fig. 1: (a) Submarine dc transmission and distribution system for oil and gas plataform and (b) proposed
dc-dc converter structure.
Bidirectional converters have interesting capabilities but these are not always required like in
non-conventional applications presented in Fig. 1 (a). This figure presents a subsea transmission and distribution system applied to oil and gas extraction, where an umbilical cable delivers
dc voltages and currents to the loads. The dc alternative for this system is very attractive since
such application is supposed to operate in tens of MWs and the cable length can reach distances up to hundreds of miles while not suffering from the well known reduction in the ac active
current-carrying on cable capability due to high capacitance underwater. The rectifier stage at
the oile processing platform could be performed by an unidirectional rectifier [11]. Once reaching the loads, a dc multiterminal distribution system is more adequate since all machines are
fed by Variable Speed Drives. Also, there is no need for a bidirectional dc-dc converter. Then,
in such a system, the use of unidirectional converter can lead to the reduction of costs and
complexity. Moreover in this application or in the ship concept [12], the converters are required
to handle tens to hundreds of kilovolts.
2.
Converter structure
The proposed converter structure is meant to withstand the characteristics presented for the
non-conventional application but also an effort was made to avoid adding semiconductors and
keep the transmision electrically isolated from distribution. Then the structure is composed of
three stages as presented in Fig. 1 (b). The Stage I has an inverter connected by its ac link
into Stage II, which is meant to control the output voltage by controlling the input current on
Stage II. The MMC topology was chosen to be the inverter for being able to handle high voltage
with lower voltage semiconductors and because of its modularity. The proposed structure is
composed of series connected Half-Bridge Submodules (HBS). By choosing so, a natural output voltage reduction is achieved. Other submodule topologies will lead to different operation
possibilities and these will not be discussed in this article. The stages II and III comprise a
Multipulse Rectifier (MR), which is well covered in the literature when fed by voltage sources
[13, 14]. In this proposal, the MMC feeds the MR as a sinusoidal current controlled source.
Once the operation differs from conventional, the differences will be explained later. The number of pulses and the association type (series or parallel) of the six pulse rectifiers are defined
according to the output voltage and desired current to be handled at the output. In this work
the specifications were as follows: input voltage of Vd =55 kV, output voltage of Vo =15 kV and
power of Po =20 MW. These specifications led to the associations presented in Fig. 1 (b).
In addition the converter was meant to have three-phases because, when balanced, it is able
(a)
(b)
Fig. 2: (a) Three-phase MMC and (b) downscale diagram of the proposed dc-dc structure.
to provide (or consume) constant power circulation at the dc port. This feature leads to the
reduction of the need for bulky passive filters in both input and output. In addition, the frequency
is chosen according to transformer losses, volume and weight criteria.
Another issue for converters meant to operate as interface between transmission and distribution systems is the fault handling. In the case of MMCs operating with HBS there is a problem
related to the submodules’s anti-parallel diodes that create a path during a fault between the
dc poles. The derivative of this current is limited only by the arm inductors [15, 16]. Circuit
breakers for dc are still in development, the most of the systems uses a vacuum interrupter and
some sort of method to create artificial current transition through zero [17]. Although the fault
clearing on the dc side is not completely solved by the structure but the ac link enables the use
of well established ac interruption techniques to isolate the load from the transmission system
in the case of fault at the load side. Faults can be selectively handled in the case of multiple
outputs, where each output rectifier would have its own protection system exemplarily at the ac
side. Other option is the use of different submodules [18].
2.1.
MMC principles
The three-phase MMC is formed by the connection of three MMC legs to a common source, dc
or with a different frequency than its output ac link. Such MMC leg is formed by the connection
of two MMC arms. Each of these arms have N identical submodules and a single inductor. The
submodules are two terminal devices composed of switches and a local dc-storage capacitor
[19]. The MMC does not need any external power source for its submodules. This set is able
to operate as a controlled voltage source by controlling the insertion time of the submodules.
Fig. 2 (a) shows a three-phase MMC connection with Half-Bridge Submodules (HBS).
3.
Converter control strategy
For the dc-dc operation the converter control strategy is performed by two main loops presented
in Fig. 3, which are based on [20]. The first one is the output voltage control Fig. 3 (a), which
uses the output voltage compensated error as the peak reference for the MMC phase current
control. As the system has a three-phase three-wire connection, only two of the MMC’s phase
(a)
(b)
Fig. 3: Converter control strategy: (a) Output voltage control and (b) internal MMC variables control.
currents are controlled, the reference for the third one is obtained by Kirchoff’s current law. The
control diagram for the output voltage is presented in Fig. 3 (a).
The second loop is necessary to control the MMC internal variables and its control diagram
is presented in Fig. 3 (b). It ensures the power equilibrium among the input source (dc link)
and the output (ac link) by the means of controlling the leg current, idj . The leg current is defined as the instantaneous average between the upper arm current and the lower arm current
and defined by: idj = (ip,j + in,j ) /2. Considering the power to flow from the dc input to the
ac link, the leg current is supposed to have an ac component at the fundamental frequency
which drains energy from the capacitors and delivers to the ac link and a dc component which
recharges the submodule capacitor. But there are also other currents that are known as circulating currents [21], resulting from submodules voltage ripples, submodule switching transitions
and arm voltage unbalances. For the sake of control simplicity the reference of the leg current
has two components. The first component is dc and realizes the power balance from dc to
ac links by means of monitoring the total voltage of the submodules for each leg. The second
component is a fundamental ac link frequency, which is synchronized with each MMC phase.
This component has as a reference the difference between the total voltage of the capacitors
from upper arm and the total capacitors voltage from the lower arm.
In addition, the method proposed in [22] is used to equalize the energy distribution among the
arm capacitors. When an arm needs to supply energy, the submodule with the higher voltage
is selected and if the arm needs to absorb energy the submodule with the lower voltage is
selected. This method is also called by other authors as selection algorithm.
4.
Converter Operation and Simulation Results
The proposed converter operation has some particularities when compared to a conventional
MR converter. In a conventional MR fed by balanced voltages the current harmonics generated
in the transformer secondary by the six pulse rectifiers are canceled in the primary side because
the secondaries are phase-shifted by the transformer [13]. When an inverter keeps a sinusoidal
current waveform at the primary windings it should be expected to have only sinusoidal currents
in the secondaries windings. But because of the phase-shifting, the current harmonics which
are not present in the primary appear in the secondary since they are generated by the six
pulse rectifiers. Also, as a reflect of the diode commutation from the six pulse rectifiers, six
voltages steps appear in each secondary and they are reflected to the primary line voltage
shifted.
For example, in a 12-pulse rectifier the 5th and 7th current components are canceled by the 30◦
phase-shift of the secondary windings. Once canceled, even while maintaining a sinusoidal
current at the primary, it is mathematically possible to have harmonic currents at the secondary
windings. This peculiarity leads to the harmonics circulation in higher frequencies, so the output
passive filters become smaller. The rectifier diodes commutate with a limited current slope,
defined by the rectifer output voltage and the transformer leakage inductance.
For the experimental tests a downsized dc-dc converter was built keeping a similar voltage
ratio, equivalent frequency at the transformer and main characteristics of the structure. The
schematic of the prototype is represented in Fig. 2 (b) and the main parameters are: input voltage 800 V, output voltage 150 V, input power 4.3
√ kW, submodule switching frequency 10 kHz,
transformer voltage ratios 6.5:1 (Y-Y) and 6.5: 3 (Y-∆), ac link frequency 400 Hz, the sample
frequency 20 kHz and the output capacitance of each six pulse rectifier is 11 µF. The modulation chosen was the POD-PWM (Phase Opposition Disposition) which generates 2N+1 voltage
levels in the phase voltage, N is the number of submodules per arm. The simulation results are
presented in Fig.4. Especial attention is given to the 12-pulses, which appear in the transformer
voltage as the MR is fed by a sinusoidal current.
170
Amplitude (V)
Amplitude (A)
10
5
0
−5
−10
160
150
140
130
1.444
1.446
1.448
Time (s)
1.45
1.444
Amplitude (V)
(a)
1.446
1.448
Time (s)
1.45
(b)
1 2
500
11
3
4
0
12
10
9
5
6 78
−500
1.443
1.444
1.445
1.446
1.447
Time (s)
1.448
1.449
1.45
(c)
Fig. 4: Simulated closed-loop converter waveforms: (a) Transformer primary-side currents; (b) output dc
voltage; and, (c) transformer primary-side line voltages.
5.
Experimental Results
The preliminary experimental results were obtained with all control loops active. The operating
point is: input voltage of 500 V, output voltage reference 74.5 V, a load of 5 Ω. The MMC current
for all its windings is presented in Fig. 5 as well its Fast Fourier Transform (FFT) spectrum. By
the FFT and THD its noticed that the primary currents are mainly composed by fundamental
frequency. For the secondary and terciary windings there are harmonics generated by each
rectifier. Secondary and terciary main harmonics are present as multiples of 5th and 7th .
The MMC phase voltages are presented in Fig. 6(a). Beyond the switching noise can be noticed
the twelve pulses of the phase-shifted rectifiers. The output voltage is presented in Fig. 6 (b)
Amplitude (A)
Amplitude (A)
0.4
5
0
−5
0
2
4
6
Time (s)
0.3
5.69 A
0.2
0.1
0
8
0
−3
x 10
2000
4000
6000
Frequency (Hz)
8000
(b)
5
20
Amplitude (A)
Amplitude (A)
(a)
10
0
−10
−20
0
2
4
6
Time (s)
4
THD 20 %
THD 21 %
THD 25 %
2
1
0
8
18.02 A
3
0
−3
x 10
(c)
2000
4000
6000
Frequency (Hz)
8000
(d)
5
20
Amplitude (A)
Amplitude (A)
THD 5 %
THD 6 %
THD 5 %
10
0
−10
−20
0
2
4
6
Time (s)
4
2
1
0
8
17.48 A
3
THD 21 %
THD 20 %
THD 20 %
0
−3
x 10
(e)
2000
4000
6000
Frequency (Hz)
8000
(f)
Fig. 5: Experimental setup transformer current waveforms: (a) Primary-side (high voltage) windings wyeconnected line currents and (b) their FFT, (c) wye-connected low voltage side windings line currents and
(d) their FFT, (e) delta-connected low voltage side windings line currents and (f) their FFT.
and a picture of the prototype is presented in Fig. 6 (c).
6.
Conclusion
A new converter structure for unidirectional dc-dc conversion in high power systems has been
presented. In this arrangement special attention is given to the MR, which is current-fed. It
provides new features such as six pulse controlled di/dt during diode switching and limited
transformer dv/dt. An advantage of this arrangement is the possibility of avoiding the use of
semiconductors connected in series if multiple low voltage side rectifiers are used. In addtion,
the medium frequency transformer allows height and raw material reduction. The main drawback of this configuration is the increased transformer construction complexity of the MR when
a high number of pulses are required to avoid the use of semiconductor in series.
Amplitude (V)
200
100
0
−100
−200
−4
−3
−2
−1
0
Time (s)
1
2
3
4
−3
x 10
Amplitude (V)
(a)
80
70
60
50
−4
−2
0
Time (s)
(b)
2
4
−3
x 10
(c)
Fig. 6: Experimental transformer voltage waveforms: (a) primary windings phase voltage and (b) secondary windings phase voltage and (c) A prototype photo.
7.
References
[1] J.I. Vuckovic. Nikola tesla: the man time forgot. Potentials, IEEE, 9(3):53–54, Oct 1990.
[2] G.F. Reed. Dc technologies: Solutions to electric power system advancements [guest
editorial]. Power and Energy Magazine, IEEE, 10(6):10–17, 2012.
[3] R.W.A.A. De Doncker, D.M. Divan, and M.H. Kheraluwala. A three-phase soft-switched
high-power-density dc/dc converter for high-power applications. Industry Applications,
IEEE Transactions on, 27(1):63–73, Jan 1991.
[4] K. Friedrich. Modern hvdc plus application of vsc in modular multilevel converter topology. In Industrial Electronics (ISIE), 2010 IEEE International Symposium on, pages 3807–
3810, 2010.
[5] R. Marquardt, A. Lesnicar, and J. Hildinger. Modulares stromrichterkonzept fr netzkupplungsanwendungen bei hohen spannungen. ETG-Conference, 2002.
[6] S. Rohner, S. Bernet, M. Hiller, and R. Sommer. Modulation, losses, and semiconductor
requirements of modular multilevel converters. Industrial Electronics, IEEE Transactions
on, 57(8):2633 –2642, aug. 2010.
[7] E. Solas, G. Abad, J.A. Barrena, S. Aurtenetxea, A. Carcar, and L. Zajac. Modular multilevel converter with different submodule concepts-part i: Capacitor voltage balancing
method. Industrial Electronics, IEEE Transactions on, 60(10):4525–4535, 2013.
[8] M. Hagiwara and H. Akagi. Control and experiment of pulsewidth-modulated modular
multilevel converters. Power Electronics, IEEE Transactions on, 24(7):1737 –1746, july
2009.
[9] T. Lüth, M.M.C. Merlin, T.C. Green, C.D. Barker, F. Hassan, R.W. Critchley, R.W. Crookes,
and K. Dyke. Performance of a dc/ac/dc vsc system to interconnect hvdc systems. In AC
and DC Power Transmission (ACDC 2012), 10th IET International Conference on, pages
1–6, 2012.
[10] S. Kenzelmann, A. Rufer, D. Dujic, F. Canales, and Y.R. de Novaes. Isolated dc/dc structure based on modular multilevel converter. (in press), 2014.
[11] G.J.M. de Sousa and M.L. Heldwein. Three-phase unidirectional modular multilevel converter. In Power Electronics and Applications (EPE), 2013 15th European Conference on,
pages 1–10, Sept 2013.
[12] G.F. Reed, B.M. Grainger, A.R. Sparacino, and Zhi-Hong Mao. Ship to grid: Mediumvoltage dc concepts in theory and practice. Power and Energy Magazine, IEEE, 10(6):70–
79, 2012.
[13] D.A. Paice and IEEE Industry Applications Society. Power electronic converter harmonics:
multipulse methods for clean power. Electrical engineering/power and energy engineering.
IEEE Press, 1996.
[14] B. Wu. High-Power Converters and AC Drives. Wiley, 2006.
[15] Xiaofang Chen, Chengyong Zhao, and Chungang Cao. Research on the fault characteristics of hvdc based on modular multilevel converter. In Electrical Power and Energy
Conference (EPEC), 2011 IEEE, pages 91–96, 2011.
[16] P. Bordignon, M. Marchesoni, G. Parodi, and L. Vaccaro. Modular multilevel converter in
hvdc systems under fault conditions. In Power Electronics and Applications (EPE), 2013
15th European Conference on, pages 1–10, Sept 2013.
[17] D. Alferov, A. Budovsky, D. Evsin, V. Ivanov, V. Sidorov, and V. Yagnov. Dc vacuum circuitbreaker. In Discharges and Electrical Insulation in Vacuum, 2008. ISDEIV 2008. 23rd
International Symposium on, volume 1, pages 173–176, Sept 2008.
[18] D. Schmitt, Y. Wang, T. Weyh, and R. Marquardt. Dc-side fault current management in
extended multiterminal-hvdc-grids. In Systems, Signals and Devices (SSD), 2012 9th
International Multi-Conference on, pages 1–5, 2012.
[19] R. Marquardt and A. Lesnicar. A new modular voltagc source inverter topology. EPE 03,
2003. France.
[20] L. Mondardo Cunico, Y.R. de Novaes, and S. Vidal Garcia Oliveira. Modeling and voltage
method for modular multilevel converter. In Energy Conversion Congress and Exposition
(ECCE), 2013 IEEE, pages 109–116, Sept 2013.
[21] S. Rohner, S. Bernet, M. Hiller, and R. Sommer. Analysis and simulation of a 6 kv, 6
mva modular multilevel converter. In Industrial Electronics, 2009. IECON ’09. 35th Annual
Conference of IEEE, pages 225 –230, 2009.
[22] M. Glinka and R. Marquardt. A new ac/ac-multilevel converter family applied to a singlephase converter. In Power Electronics and Drive Systems, 2003. PEDS 2003. The Fifth
International Conference on, volume 1, pages 16 – 23 Vol.1, 2003.