Complementary MOS (CMOS) Inverter
■
Concept: transistor switches connect output either to VDD or to ground
VDD
INPUT
HIGH
VDD
OUTPUT
LOW
INPUT
LOW
OUTPUT
HIGH
CL
(a)
■
VDD
VIN
CL
(b)
+
CL
VOUT
−
(c)
Practical realization: connect input to gate of p-channel device.
VIN = VDD --> VSG2 = VDD - VIN = 0 < - VTp --> cutoff
VIN = 0 --> VSG2 = VDD - VIN = VDD >> - VTp --> on (triode region)
■
Graphical analysis: need to find family of load lines since input is connected to
gate of M2
EE 105 Spring 1997
Lecture 11
p-Channel MOSFET Characteristics
■
p-channel MOS load device:
VSGp = VDD - VIN
as VIN increases, the source-gate voltage VSGp decreases.
VDD
VSGp +
_
+
+
VSDp = vSUP
_
- IDp= iSUP
VIN
_
note that the bulk connection is tied to the source (VDD), which results in a constant
threshold voltage.
EE 105 Spring 1997
Lecture 11
Switchable Current-Source Pull-Up
* The drain characteristics are - IDp = - IDp (VSG, VSD), which can be expressed as
the “switchable” pull-up’s current-voltage characteristic,
iSUP = iSUP(VIN, vSUP)
since iSUP = -IDp and VSG = VDD - VIN and vSUP = VSD.
- IDp =
iSUP
1
2
3
4
5
VSD = vSUP
- VTp
EE 105 Spring 1997
Lecture 11
CMOS Transfer Characteristic
■
plotting the p-channel pull-up on the n-channel “driver’s” drain characteristics
allows us to find the input-output voltage pairs that satisfy the constraint that
IDn = - IDp
VOUT
VDD 1
2
3
4
5
VDD
VIN
−IDp = IDn
IDn = −IDp
VIN
3
3
4
1
5
VDD
n-channel
(a)
2
4
2
VOUT
1
VDD
5
VOUT
p-channel
(b)
EE 105 Spring 1997
Lecture 11
Simplified Voltage Transfer Curve
■
For CMOS inverters, the voltage transfer curve of the inverter is ideal enough
that we can approximate it with a construction that is suitable for quick hand
calculation
VOUT
VOH = VMAX
Slope Av
VM
VOL = VMIN
VIL
■
VM
VIH
VIN
We first observe that:
V OH ≈ V MAX = V DD and V OL ≈ V MIN = 0 V
The edges of the transition region are then found as the intersections of the
tangent to the voltage transfer curve at VIN = VM (a line of slope Av)
■
In order to construct the VTC for a CMOS inverter (and to find estimates of the
noise margins), we need to first:
(i) find the voltage VM
(ii) find the small-signal voltage gain Av at VIN = VM
EE 105 Spring 1997
Lecture 11
Step 1. Finding VM
■
Goal: find VM = input voltage for the output = VM
both transistors are saturated at VIN = VM since
VDSn = VM - 0 > VM - VTn
VSDp = VDD - VM = (VDD - VM) +VTp
■
Equate drain currents, omitting the channel length modulation terms
(1 + λn VDSn) and (1 + λp VSDp) since they tend to cancel out (if λn = λp,
they exactly cancel out)
2
W
I Dn = µ n C ox  ------ ( V M – V Tn )
 2L n
2
W
– I Dp = µ p C ox  ------ ( V DD – V + V Tp )
 2L p
M
■
Letting kn = µn Cox (W/L)n and kp = µp Cox (W/L)p --
2
2
1
1
--- k n ( V M – V Tn ) = --- k p ( V DD – V M + V Tp )
2
2
EE 105 Spring 1997
Lecture 11
Finding VM (cont.)
Result:
kp
V Tn + ----- ( V DD + V Tp )
kn
V M = -----------------------------------------------------------kp
1 + ----kn
We can set VM = VDD / 2 and achieve a symmetrical transfer curve
Example: suppose VTn = - VTp = 1 V and VDD = 5 V
kp
1 + 4 ----kn
V M = ----------------------- = 2.5 V --> kp = kn
kp
1 + ----kn
which makes sense since the transistors must have identical characteristics for the
transfer curve to be symmetrical.
The mobility of holes in p-channels is about half that of electrons in
n-channels, µp = µn / 2, which implies that we must adjust the width-length
ratios to compensate:
kn = kp --> (W/L)p = 2(W/L)n
EE 105 Spring 1997
Lecture 11
Step 2. Finding Av
s2
+
gmpvsg2
vsg2
_
g1 = g2
d1=d2
+
+
vgs1
vin
_
rop
vout
gmnvgs1
ron
_
s1
We note that vsg2 = - vin and can simplify the small-signal circuit
+
vin
+
−
v
−
+
gmnv
ron
rop
gmpv
vout
−
EE 105 Spring 1997
Lecture 11
Approximate Transfer Curve
■
The small-signal gain (which is the slope of the transfer curve when the input is
equal to the mid-point voltage) is:
v out ⁄ v in = – ( g mn + g mp ) ( r on r op ) = A v
CMOS inverters have a channel length that is as short as possible (to minimize the
area ... and maximum the density) ... the output resistances are relatively small and a
typical value is vout / vin = - 5 to - 10.
* The input-low and input-high voltages are:
VOUT
VOH = VDD
Av
VDD − VM
Av
VOL = 0 V
VIL
VM
VIH
VIN
V IL = V M – ( V DD ⁄ ( 2 A v ) )
V IH = V M + ( V DD ⁄ ( 2 A v ) )
EE 105 Spring 1997
Lecture 11
Noise Margins
■
For kN = kP, the mid-point voltage is VM = 2.5 V. For a slope Av = - 5, the inputlow voltage and input-high voltages are:
VIL = 2.5 V - (1/5) (2.5 V) = 2 V
VIH = 2.5 V + (1/5) (2.5 V) =3 V
The low and high noise margins are therefore:
NML = VIL - VOL = 2 - 0 = 2 V
NMH = VOH - VIH =5 - 3 = 2 V
The transition region (or “gray area”) is the interval
VIL < VIN < VIH
■
or 2 V < VIN < 3 V
Finding the actual transfer function requires solving the drain current equations
when the p-channel and n-channel are in the appropriate operating regions ... and
finding the transition voltages for the regions.
SPICE is good at this job!
EE 105 Spring 1997
Lecture 11
CMOS Inverter: Propagation Delay
■
The propagation delays tPHL and tPLH are obviously of major importance for
digital circuit design ...
Example:
clock frequency = 250 MHz --> clock period = 4 ns
complex systems (e.g., microprocessor) have around 20-50 propagation delays
per clock period, so we need to have
tPLH and tPHL < 100 ps = 10-10 s
■
Hand calculation of propagation delays: use approximation that input changes
instantaneously
VIN
VOH
tCYCLE
VOL
t
VOUT
tPHL
tPLH
VOH
VOH
50%
tCYCLE
VOL
t
EE 105 Spring 1997
Lecture 11
Estimating the Load Capacitance
■
The load capacitance CL consists of
CG, the input capacitances of the inverters 2 and 3, and
CP, the parasitic capacitance to the substrate from the drain regions of inverter 1
and the interconnections between the output of inverter 1 and the inputs of
inverters 2 and 3.
VDD
W
L p2
VDD
VDD
2
W
L n2
W
L p1
VIN
+
CL
VOUT
VIN
1
VDD
W
L n1
W
L p3
−
3
W
L n3
(a)
■
(b)
For hand calculation, we do a worst case estimate of CG by adding the maximum
gate capacitances for inverters 2 and 3
C G = C ox [ ( W ⋅ L ) p2 + ( W ⋅ L ) + ( W ⋅ L ) p3 + ( W ⋅ L ) ]
n2
n3
EE 105 Spring 1997
Lecture 11
Parasitic Capacitance from Drain Depletion Regions
■
The drain n and p regions have depletion regions whose stored charge changes
during the transient.
,,
,
,,
,,
Take the worst case and use the zero-bias depletion capacitance (the maximum
value) as a linear charge-storage element during the transient.
,,,,,,,,
,
,
,,,
, , ,,,,,,,,,,,,,,
,,,,, ,,,,,,,,,,,,
,,,,, , , , , , ,
,,,,, , , , , ,
,
,
,,,,,,,,,,,,
,,,,,,,,,,,,,,,,,,
active area (thin
oxide area)
gate contact
gate
interconnect
polysilicon gate
contact
n+ polysilicon gate
A
metal
interconnect
source contacts
W
bulk
contact
source
interconnect
drain
interconnect
L
drain
contacts
edge of
active area
(b)
L
W
Ldiff
EE 105 Spring 1997
Lecture 11
Calculation of Parasitic Depletion Capacitance
■
“Bottom” of depletion regions of the load inverters’ drain diffusions contribute a
depletion capacitance
CBOTT = CJn(WnLdiffn) + CJp(WpLdiffp)
with CJn and CJp being the zero-bias junction capacitances (fF/µm2) for the nchannel MOSFET drain-bulk junction and the p-channel MOSFET drain-bulk
junction, respectively.
■
“Sidewall” of depletion regions of the load inverters’ drain diffusions make an
additional contribution:
CSW = (Wn + 2Ldiffn)CJSWn + (Wp + 2Ldiffp)CJSWp
with CJSWn and CJSWp being the zero-bias sidewall capacitances (fF/µm) for the
n-channel MOSFET drain-bulk junction and the p-channel MOSFET drain-bulk
junction, respectively.
■
■
The total depletion capacitance CDB = CBOTT + CSW
Typical numbers: CJN and CJP are about 0.2 fF/µm2 and
CJSWn and CJSWp are about 0.5 fF/µm.
EE 105 Spring 1997
Lecture 11
Parasitic Capacitance from Interconnections
,,
■
“Wires” consist of metal lines connecting the output of the inverter to the input
of the next stage. In cross section,
,,
,,,,,
,,,,,
,,,,,
,,,,,
,,,,,
,,,,,
,,,,,
,,,,,
,
,
,
metal interconnect
(width Wm, length Lm)
polysilicon
gate
p+
0.6 µm deposited oxide
0.5 µm thermal oxide
p
(grounded)
gate oxide
■
The p+ layer (i.e., heavily doped with acceptors) under the thick thermal oxide
(500 nm = 0.5 µm) and deposited oxide (600 nm = 0.6 µm) depletes only
slightly when positive voltages appear on the metal line, so the capacitance is
approximately the oxide capacitance:
C WIRE = C thickox ( W m ⋅ L m )
where the oxide thickness = 500 nm + 600 nm = 1.1 µm.
* For large digital systems, the parasitic interconnect capacitance can dominate the
load capacitance -CL = CG + CP = CG + (CDB + CWIRE)
EE 105 Spring 1997
Lecture 11
EE 105 Spring 1997
Lecture 11