a
FEATURES
Recovers Signal from +100 dB Noise
2 MHz Channel Bandwidth
45 V/ms Slew Rate
–120 dB Crosstalk @ 1 kHz
Pin Programmable Closed Loop Gains of 61 and 62
0.05% Closed Loop Gain Accuracy and Match
100 mV Channel Offset Voltage (AD630BD)
350 kHz Full Power Bandwidth
Chips Available
Balanced Modulator/Demodulator
AD630
FUNCTIONAL BLOCK DIAGRAM
CM OFF
ADJ
CM OFF
ADJ
DIFF OFF
ADJ
DIFF OFF
ADJ
6
5
4
3
2.5kΩ
RINA
1
CHA+
2
AD630
AMP A
CHA– 20
A
12
COMP
11
+VS
13
VOUT
14
RB
15
RF
16
RA
2.5kΩ
RINB 17
AMP B
B
10kΩ
CHB+ 18
10kΩ
–V
CHB– 19
5kΩ
PRODUCT DESCRIPTION
The AD630 is a high precision balanced modulator which combines a flexible commutating architecture with the accuracy and
temperature stability afforded by laser wafer trimmed thin-film
resistors. Its signal processing applications include balanced
modulation and demodulation, synchronous detection, phase
detection, quadrature detection, phase sensitive detection,
lock-in amplification and square wave multiplication. A network
of on-board applications resistors provides precision closed loop
gains of ± 1 and ± 2 with 0.05% accuracy (AD630B). These resistors may also be used to accurately configure multiplexer
gains of +1, +2, +3 or +4. Alternatively, external feedback may
be employed allowing the designer to implement his own high
gain or complex switched feedback topologies.
The AD630 may be thought of as a precision op amp with two
independent differential input stages and a precision comparator
which is used to select the active front end. The rapid response
time of this comparator coupled with the high slew rate and fast
settling of the linear amplifiers minimize switching distortion. In
addition, the AD630 has extremely low crosstalk between channels of –100 dB @ 10 kHz.
The AD630 is intended for use in precision signal processing
and instrumentation applications requiring wide dynamic range.
When used as a synchronous demodulator in a lock-in amplifier
configuration, it can recover a small signal from 100 dB of interfering noise (see lock-in amplifier application). Although optimized
for operation up to 1 kHz, the circuit is useful at frequencies up
to several hundred kilohertz.
Other features of the AD630 include pin programmable frequency
compensation, optional input bias current compensation resistors, common-mode and differential-offset voltage adjustment,
and a channel status output which indicates which of the two
differential inputs is active. This device is now available to Standard Military Drawing (DESC) numbers 5962-8980701RA and
5962-89807012A.
COMP
7
SEL B 9
CHANNEL
STATUS
B/A
SEL A 10
8
–VS
PRODUCT HIGHLIGHTS
1. The configuration of the AD630 makes it ideal for signal
processing applications such as: balanced modulation and
demodulation, lock-in amplification, phase detection, and
square wave multiplication.
2. The application flexibility of the AD630 makes it the best
choice for many applications requiring precisely fixed gain,
switched gain, multiplexing, integrating-switching functions,
and high-speed precision amplification.
3. The 100 dB dynamic range of the AD630 exceeds that of any
hybrid or IC balanced modulator/demodulator and is comparable to that of costly signal processing instruments.
4. The op-amp format of the AD630 ensures easy implementation of high gain or complex switched feedback functions.
The application resistors facilitate the implementation of
most common applications with no additional parts.
5. The AD630 can be used as a two channel multiplexer with
gains of +1, +2, +3 or +4. The channel separation of
100 dB @ 10 kHz approaches the limit which is achievable
with an empty IC package.
6. The AD630 has pin-strappable frequency compensation (no
external capacitor required) for stable operation at unity gain
without sacrificing dynamic performance at higher gains.
7. Laser trimming of comparator and amplifying channel offsets
eliminates the need for external nulling in most cases.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
AD630–SPECIFICATIONS (@ + 258C and 6V = 615 V unless otherwise noted)
S
Model
Min
GAIN
Open Loop Gain
± 1, ± 2 Closed Loop Gain Error
Closed Loop Gain Match
Closed Loop Gain Drift
CHANNEL INPUTS
VIN Operational Limit1
Input Offset Voltage
Input Offset Voltage
TMIN to TMAX
Input Bias Current
Input Offset Current
Channel Separation @ 10 kHz
COMPARATOR
VIN Operational Limit1
Switching Window
Switching Window
TMIN to TMAX2
Input Bias Current
Response Time (–5 mV to +5 mV Step)
Channel Status
ISINK @ VOL = –VS + 0.4 V3
Pull-Up Voltage
AD630J/A
Typ
Max
110
0.1
0.1
2
90
100
200
AD630K/B
Typ
Max
120
800
300
50
100
10
100
62.0
300
100
200
610
160
300
50
4
62.0
300
100
200
90
90
65
110
110
4
± 16.5
5
0
–25
(–VS + 33 V)
90
90
65
0
–25
µV
nA
nA
dB
Volts
mV
mV
nA
ns
mA
Volts
MHz
V/µs
µs
110
110
dB
dB
Volts
mA
± 16.5
5
Volts
mA
25
+70
+85
Volts
µV
2
45
3
4
25
+70
+85
62.5
300
± 10
610
25
1000
300
50
1.6
(–VS + 33 V)
Units
dB
%
%
ppm °C
(–VS + 3 V) to (+VS – 1.3 V)
61.5
2
45
3
± 16.5
5
110
0.1
0.1
2
100
10
100
1.6
105
110
Max
(–VS +4 V) to (+VS – 1 V)
500
(–VS + 3 V) to (+VS – 1.5 V)
61.5
2
45
3
OUTPUT VOLTAGE, @ RL = 2 kΩ
TMIN to TMAX2
Output Short Circuit Current
90
(–VS +4 V) to (+VS – 1 V)
100
(–VS + 33 V)
85
90
65
AD630S
Typ
2
1.6
OPERATING CHARACTERISTICS
Common-Mode Rejection
Power Supply Rejection
Supply Voltage Range
Supply Current
Min
0.05
0.05
(–VS + 3 V) to (+VS – 1.5 V)
61.5
DYNAMIC PERFORMANCE
Unity Gain Bandwidth
Slew Rate4
Settling Time to 0.1% (20 V Step)
TEMPERATURE RANGES
Rated Performance–N Package
Rated Performance–D Package
100
(–VS + 4 V) to (+VS – 1 V)
500
100
10
100
Min
N/A
–55
+125
°C
°C
NOTES
1
If one terminal of each differential channel or comparator input is kept within these limits the other terminal may be taken to the positive supply.
2
These parameters are guaranteed but not tested for J and K grades. For A, B and S grades they are tested.
3
ISINK @ VOL = (–VS + 1) volt is typically 4 mA.
4
Pin 12 Open. Slew rate with Pins 12 and 13 shorted is typically 35 V/ µs.
Specifications subject to change without notice.
Specifications shown in boldface are tested on all production units at final electrical test. Results from those tests are used to calculate outgoing quality levels. All
min and max specifications are guaranteed, although only those shown in boldface are tested on all production units.
ABSOLUTE MAXIMUM RATINGS
ORDERING GUIDE
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .± 18 V
Internal Power Dissipation . . . . . . . . . . . . . . . . . . . . . . 600 mW
Output Short Circuit to Ground . . . . . . . . . . . . . . . . Indefinite
Storage Temperature, Ceramic Package . . . . –65°C to +150°C
Storage Temperature, Plastic Package . . . . . . –55°C to +125°C
Lead Temperature Range (Soldering, 10 sec ) . . . . . . . +300°C
Max Junction Temperature . . . . . . . . . . . . . . . . . . . . . +150°C
THERMAL CHARACTERISTICS
20-Pin Plastic DIP (N)
20-Pin Ceramic DIP (D)
20-Pin Leadless Chip Carrier (E)
θJC
θJA
24°C/W
35°C/W
35°C/W
61°C/W
120°C/W
120°C/W
–2–
Model
Temperature
Range
Package
Description
Package
Option
AD630JN
AD630KN
AD630AD
AD630BD
AD630SD
AD630SD/883B
5962-8980701RA
AD630SE/883B
5962-89807012A
AD630J Chip
AD630S Chip
0°C to +70°C
0°C to +70°C
–25°C to +85°C
–25°C to +85°C
–55°C to +125°C
–55°C to +125°C
–55°C to +125°C
–55°C to +125°C
–55°C to +125°C
0°C to +70°C
–55°C to +125°C
Plastic DIP
Plastic DIP
Side Brazed DIP
Side Brazed DIP
Side Brazed DIP
Side Brazed DIP
Side Brazed DIP
LCC
LCC
Chip
Chip
N-20
N-20
D-20
D-20
D-20
D-20
D-20
E-20A
E-20A
REV. A
AD630
CHIP METALIZATION AND PINOUT
PIN CONFIGURATIONS
Dimensions shown in inches and (mm).
Contact factory for latest dimensions
20-Pin DIP (D-20 and N-20)
RINA 1
20 CH A–
CH A+ 2
19 CH B–
DIFF OFF ADJ 3
18 CH B+
DIFF OFF ADJ 4
17 RIN B
AD630
CM OFF ADJ 5
16 RA
TOP VIEW
CM OFF ADJ 6 (Not to Scale) 15 RF
14 RB
CHANNEL STATUS B/A 7
–VS 8
13 VOUT
SEL B 9
12 COMP
SEL A 10
11 +VS
The AD630 is available in laser trimmed, passivated chip form.
The figure shows the AD630 metalization pattern, bonding pads
and dimensions. AD630 chips are available; consult factory for
details.
CH A–
2
1 20 19
CH B–
RIN A
3
CHIP AVAILABILITY
CH A+
DIFF
OFF ADJ
20-Contact LCC (E-20A)
18 CH B+
DIFF OFF ADJ 4
17 RIN B
CM OFF ADJ 5
AD630
CM OFF ADJ 6
TOP VIEW
(Not to Scale)
CHANNEL STATUS B/A 7
16 RA
15 RF
14 R
B
–VS 8
15
5kΩ
Vi
5kΩ
5kΩ
VO
5
2kΩ
100pF
CL = 100pF
f = 1kHz
10
5kΩ
5kΩ
Vi
5
VO
RL
CAP IN
1k
10k
100k
FREQUENCY – Hz
1M
1
100
1k
10k 100k
RESISTIVE LOAD – Ω
VO
5
f = 1kHz
CL = 100pF
5
10
15
SUPPLY VOLTAGE – ±V
Figure 3. Output Voltage Swing
vs. Supply Voltage
120
60
120
100pF
10
0
1M
5kΩ
Vi
2kΩ
Figure 2. Output Voltage vs.
Resistive Load
Figure 1. Output Voltage vs.
Frequency
0
80
20
40
0
COMPENSATED
–20
–40
20
0
dt
60
100
OPEN LOOP GAIN – dB
40
– V/µs
100
1
10
100
1k
10k
FREQUENCY – Hz
100k
Figure 4. Common-Mode
Rejection vs. Frequency
REV. A
–60
–5 –4 –3 –2 –1 0 1 2 3
INPUT VOLTAGE – V
Figure 5.
5
45
80
60
40
90
COMPENSATED
135
20
0
4
dVO
vs. Input Voltage
dt
–3–
UNCOMPENSATED
10
100
1k 10k 100k
FREQUENCY – Hz
1M
180
10M
Figure 6. Gain and Phase vs.
Frequency
OPEN LOOP PHASE – Degrees
UNCOMPENSATED
DVO
COMMON MODE REJECTION – dB
10
100pF
OUTPUT VOLTAGE – ±V
OUTPUT VOLTAGE – ±V
OUTPUT VOLTAGE – ±V
15
10
VOUT
+VS
18
15
RL= 2kΩ
CL = 100pF
COMP
Typical Performance Characteristics
SEL A
SEL B
9 10 11 12 13
AD630 –Typical Performance Characteristics
15
16
5kΩ
20
CH
A
VO
13
Vi
TOP
TRACE
12
CH
B
2 CH A
12
1kΩ
10kΩ
14
Vi
10kΩ
14 10kΩ 15 20
19
18
10kΩ
10kΩ
2
MIDDLE
TRACE
(A)
30pF
9
VO
BOTTOM
TRACE
13
Vi
TOP
TRACE
14
15
10kΩ
20
2 CH A
12
VO
BOTTOM
TRACE
13
10kΩ
10kΩ
10kΩ
10kΩ
(B)
MIDDLE
TRACE
HP5082-2811
TEKTRONIX
7A13
10
Figure 7. Channel-to-Channel SwitchSettling Characteristic
Figure 9. Large Signal Inverting
Step Response
Figure 8. Small Signal Noninverting
Step Response
TWO WAYS TO LOOK AT THE AD630
The functional block diagram of the AD630 also shows the pin
connections of the internal functions. An alternative architectural diagram is shown in Figure 10. In this diagram, the
individual A and B channel preamps, the switch, and the integrator output amplifier are combined in a single op amp. This
amplifier has two differential input channels, only one of which
is active at a time.
Vi
2
RA 5kΩ
VO
10
RB
10kΩ
2.5kΩ
Figure 11. AD630 Symmetric Gain (± 2)
RF
10kΩ
A
When channel B is selected, the resistors RA and RF are connected for inverting feedback as shown in the inverting gain
configuration diagram in Figure 12. The amplifier has sufficient
loop gain to minimize the loading effect of RB at the virtual
ground produced by the feedback connection. When the sign of
the comparator input is reversed, input B will be deselected and
A will be selected. The new equivalent circuit will be the noninverting gain configuration shown below. In this case RA will appear
across the op-amp input terminals, but since the amplifier drives
this difference voltage to zero the closed loop gain is unaffected.
13
19
B
12
2.5kΩ
17
7
SEL B
13
B
9
1
18
18
RF
10kΩ
14
14
20
A
19
RB
10kΩ
11
16
2
15
20
+VS
15
RA
16 5kΩ
9
B/A
SEL A 10
8
–VS
Figure 10. Architectural Block Diagram
HOW THE AD630 WORKS
The two closed loop gain magnitudes will be equal when RF/RA
= 1 + RF/RB, which will result from making RA equal to RFRB/
(RF + RB) the parallel equivalent resistance of RF and RB.
The basic mode of operation of the AD630 may be more easy to
recognize as two fixed gain stages which may be inserted into the
signal path under the control of a sensitive voltage comparator.
When the circuit is switched between inverting and noninverting
gain, it provides the basic modulation/demodulation function.
The AD630 is unique in that it includes Laser-Wafer-Trimmed
thin-film feedback resistors on the monolithic chip. The configuration shown below yields a gain of ± 2 and can be easily changed to
±1 by shifting RB from its ground connection to the output.
The 5k and the two 10k resistors on the AD630 chip can be
used to make a gain of two as shown here. By paralleling the
10k resistors to make RF equal 5k and omitting RB the circuit
can be programmed for a gain of ± 1 (as shown in Figure 18a).
These and other configurations using the on chip resistors
present the inverting inputs with a 2.5k source impedance. The
more complete AD630 diagrams show 2.5k resistors available at
the noninverting inputs which can be conveniently used to minimize errors resulting from input bias currents.
The comparator selects one of the two input stages to complete
an operational feedback connection around the AD630. The deselected input is off and has negligible effect on the operation.
–4–
REV. A
AD630
faster the output signal will move. This feature helps insure
rapid, symmetric settling when switching between inverting and
noninverting closed loop configurations.
RF 10kΩ
RA
5kΩ
Vi
RB
10kΩ
RF
V
RA i
VO = –
The output section of the AD630 includes a current mirror-load
(Q24 and Q25), an integrator-voltage gain stage (Q32), and
complementary output buffer (Q44 and Q74). The outputs of
both transconductance stages are connected in parallel to the
current mirror. Since the deselected input stage produces no
output current and presents a high impedance at its outputs,
there is no conflict. The current mirror translates the differential
output current from the active input transconductance amplifier
into single ended form for the output integrator. The complementary output driver then buffers the integrator output produce a low impedance output.
Figure 12. Inverting Gain Configuration
Vi
RA
5kΩ
VO = (1+
RF
RB
) Vi
RF
10kΩ
RB
10kΩ
OTHER GAIN CONFIGURATIONS
Many applications require switched gains other than the ± 1 and
± 2 which the self-contained applications resistors provide. The
AD630 can be readily programmed with three external resistors
over a wide range of positive and negative gain by selecting and
RB and RF to give the noninverting gain 1 + RF/RB and subsequent
RA to give the desired inverting gain. Note that when the inverting
magnitude equals the noninverting magnitude, the value of RA is
found to be RB RF/(RB + RF). That is, RA should equal the parallel
combination of RB and RF to match positive and negative gain.
Figure 13. Noninverting Gain Configuration
CIRCUIT DESCRIPTION
The simplified schematic of the AD630 is shown in Figure 15.
It has been subdivided into three major sections, the comparator,
the two input stages and the output integrator. The comparator
consists of a front end made up of Q52 and Q53, a flip-flop load
formed by Q3 and Q4, and two current steering switching cells
Q28, Q29 and Q30, Q31. This structure is designed so that a
differential input voltage greater than 1.5 mV in magnitude
2
20
+VS
The feedback synthesis of the AD630 may also include reactive
impedance. The gain magnitudes will match at all frequencies if
the A impedance is made to equal the parallel combination of
the B and F impedances. Essentially the same considerations
apply to the AD630 as to conventional op-amp feedback circuits.
Virtually any function which can be realized with simple noninverting “L network” feedback can be used with the AD630. A
common arrangement is shown in Figure 15. The low frequency
gain of this circuit is 10. The response will have a pole (–3 dB)
at a frequency f . 1/(2 π 100 kΩC) and a zero (3 dB from the
high frequency asymptote) at about 10 times this frequency.
The 2k resistor in series with each capacitor mitigates the loading effect on circuitry driving this circuit, eliminates stability
problems, and has a minor effect on the pole-zero locations.
CH B–
CH A+ CH B+
CH A–
18
19
11
Q33
Q35
Q34
Q36
i55
i73
Q44
SEL A
10
Q52
Q53
Q62
Q65
Q67
Q70
13
9
C121
SEL B
Q30
12
COMP
Q31
Q28
C122
Q29
Q24
Q3
–VS
VO
Q74
Q4
i22
Q32
Q25
i23
8
3
4
5
6
DIFF
OFF ADJ
DIFF
OFF ADJ
CM
OFF ADJ
CM
OFF ADJ
As a result of the reactive feedback, the high frequency components
of the switched input signal will be transmitted at unity gain
C
Figure 14. AD630 Simplified Schematic
C
2kΩ
10kΩ
applied to the comparator inputs will completely select one the
switching cells. The sign of this input voltage determine which
of the two switching cells is selected.
100kΩ
Vi
2
20
The collectors of each switching cell connect to an input transconductance stage. The selected cell conveys bias currents i22
and i23 to the input stage it controls causing it to become active.
The deselected cell blocks the bias to its input stage which, as a
consequence, remains off.
A
13
VO
19
11.11kΩ
18
B
12
7
9
10
8
The structure of the transconductance stages is such that they
present a high impedance at their input terminals and draw no
bias current when deselected. The deselected input does not
interfere with the operation of the selected input insuring maximum channel separation.
–V S
Figure 15. AD630 with External Feedback
while the low frequency components will be amplified. This arrangement is useful in demodulators and lock-in amplifiers. It
increases the circuit dynamic range when the modulation or
interference is substantially larger than the desired signal amplitude. The output signal will contain the desired signal multiplied by the low frequency gain (which may be several hundred
for large feedback ratios) with the switching signal and interference superimposed at unity gain.
Another feature of the input structure is that it enhances the
slew rate of the circuit. The current output of the active stage
follows a quasi-hyperbolic-sine relationship to the differential input voltage. This means that the greater the input voltage, the
harder this stage will drive the output integrator, and hence, the
REV. A
2kΩ
–5–
AD630
This is because the open collector channel status output inverts
the output sense of the internal comparator.
SWITCHED INPUT IMPEDANCE
The noninverting mode of operation is a high input impedance
configuration while the inverting mode is a low input impedance
configuration. This means that the input impedance of the circuit undergoes an abrupt change as the gain is switched under
control of the comparator. If gain is switched when the input
signal is not zero, as it is in many practical cases, a transient will
be delivered to the circuitry driving the AD630. In most applications, this will require the AD630 circuit to be driven by a low
impedance source which remains “stiff “ at high frequencies.
Generally this will be a wideband buffer amplifier.
+5V
100kΩ
1MΩ
100kΩ
9
7
10
8
–15V
100Ω
Figure 16. Comparator Hysteresis
FREQUENCY COMPENSATION
The AD630 combines the convenience of internal frequency
compensation with the flexibility of external compensation by
means of an optional self-contained compensation capacitor.
The channel status output may be interfaced with TTL inputs
as shown in Figure 17. This circuit provides appropriate level
shifting from the open-collector AD630 channel status output to
TTL inputs.
In gain of ± 2 applications the noise gain which must be addressed
for stability purposes is actually 4. In this circumstance, the
phase margin of the loop will be on the order of 60° without the
optional compensation. This condition provides the maximum
bandwidth and slew-rate for closed-loop gains of |2| and above.
+5V
+15V
6.8kΩ
When the AD630 is used as a multiplexer, or in other configurations where one or both inputs are connected for unity gain
feedback, the phase margin will be reduced to less than 20°.
This may be acceptable in applications where fast slewing is a
first priority, but the transient response will not be optimum.
For these applications, the self-contained compensation capacitor may be added by connecting Pin 12 to Pin 13. This connection reduces the closed loop bandwidth somewhat, and improves
the phase margin.
100kΩ
AD630
22kΩ
IN 914's
7
TTL INPUT
2N2222
8
–15V
Figure 17. Channel Status—TTL Interface
APPLICATIONS: BALANCED MODULATOR
Perhaps the most commonly used configuration of the AD630 is
the balanced modulator. The application resistors provide precise symmetric gains of ± 1 and ± 2. The ± 1 arrangement is
shown in Figure 18a and the ± 2 arrangement is shown in Figure 18b. These cases differ only in the connection of the 10k
feedback resistor (Pin 14) and the compensation capacitor (Pin
12). Note the use of the 2.5 kΩ bias current compensation resistors in these examples. These resistors perform the identical
function in the ± 1 gain case. Figure 19 demonstrates the performance of the AD630 when used to modulate a 100 kHz square
wave carrier with a 10 kHz sinusoid. The result is the double
sideband suppressed carrier waveform.
For intermediate conditions, such as gain of ± 1 where loop attenuation is 2, use of the compensation should be determined
by whether bandwidth or settling response must be optimized.
The optional compensation should also be used when the AD630
is driving capacitive loads or whenever conservative frequency
compensation is desired.
OFFSET VOLTAGE NULLING
The offset voltages of both input stages and the comparator
have been pretrimmed so that external trimming will only be required in the most demanding applications. The offset adjustment of the two input channels is accomplished by means of a
differential and common-mode scheme. This facilitates fine adjustment of system errors in switched gain applications. With
system input tied to 0 V, and a switching or carrier waveform
applied to the comparator, a low level square wave will appear at
the output. The differential offset adjustment pot can be used
to null the amplitude of this square wave (Pins 3 and 4). The
common-mode offset adjustment can be used to zero the residual dc output voltage (Pins 5 and 6). These functions should
be implemented using 10k trim pots with wipers connected
directly to Pin 8 as shown in Figures 18a and 18b.
These balanced modulator topologies accept two inputs, a signal
(or modulation) input applied to the amplifying channels, and a
reference (or carrier) input applied to the comparator.
10kΩ
6
MODULATION
INPUT
10kΩ
CM
ADJ
DIFF
ADJ
4
5
3
2.5kΩ
1
AMP A
12
A
2
11
20
2.5kΩ
CHANNEL STATUS OUTPUT
17
The channel status output, Pin 7, is an open collector output
referenced to –VS which can be used to indicate which of the
two input channels is active. The output will be active (pulled
low) when Channel A is selected. This output can also be used
to supply positive feedback around the comparator. This produces hysteresis which serves to increase noise immunity. Figure 16 shows an example of how hysteresis may be implemented. Note that the feedback signal is applied to the inverting
(–) terminal of the comparator to achieve positive feedback.
18
B
AMP B
CARRIER
INPUT
10kΩ
14
–V
19
+VS
13
10kΩ
AD630
COMP
MODULATED
OUTPUT
SIGNAL
15
16
5kΩ
7
9
10
8
–VS
Figure 18a. AD630 Configured as a Gain-of-One Balanced
Modulator
–6–
REV. A
AD630
10kΩ
6
MODULATION
INPUT
10kΩ
CM
ADJ
4
5
LVDT SIGNAL CONDITIONER
DIFF
ADJ
Many transducers function by modulating an ac carrier. A Linear Variable Differential Transformer (LVDT) is a transducer of
this type. The amplitude of the output signal corresponds to
core displacement. Figure 20 shows an accurate synchronous
demodulation system which can be used to produce a dc voltage
which corresponds to the LVDT core position. The inherent
precision and temperature stability of the AD630 reduce demodulator drift to a second order effect.
3
2.5kΩ
1
AMP A
12
A
2
11
20
2.5kΩ
17
B
AMP B
19
CARRIER
INPUT
10kΩ
14
–V
18
+VS
13
10kΩ
AD630
COMP
MODULATED
OUTPUT
SIGNAL
15
16
E1000
AD544
SCHAEVITZ
FOLLOWER
LVDT
16 B 5kΩ
5kΩ
7
9
A
10
8
AD630
±2 DEMODULATOR
15
2.5kHZ
2V pk-pk
SINUSOIDAL
EXCITATION
–VS
Figure 18b. AD630 Configured as a Gain-of-Two Balanced
Modulator
1
2.5kΩ
14
10kΩ
20
10kΩ
A
C 13 100kΩ
19
B
17
12
D
1µF
2.5kΩ
9
PHASE
SHIFTER
10
Figure 20. LVDT Signal Conditioner
AC BRIDGE
Bridge circuits which use dc excitation are often plagued by errors caused by thermocouple effects, 1/f noise, dc drifts in the
electronics, and line noise pick-up. One way to get around these
problems is to excite the bridge with an ac waveform, amplify
the bridge output with an ac amplifier, and synchronously demodulate the resulting signal. The ac phase and amplitude information from the bridge is recovered as a dc signal at the
output of the synchronous demodulator. The low frequency system noise, dc drifts, and demodulator noise all get mixed to the
carrier frequency and can be removed by means of a low-pass
filter. Dynamic response of the bridge must be traded off against
the amount of attenuation required to adequately suppress these
residual carrier components in the selection of the filter.
Figure 21 is an example of an ac bridge system with the AD630
used as a synchronous demodulator. The oscilloscope photograph shows the results of a 0.05% bridge imbalance caused by
the 1 Meg resistor in parallel with one leg of the bridge. The top
trace represents the bridge excitation, the upper-middle trace is
the amplified bridge output, the lower-middle trace is the output of the synchronous demodulator and the bottom trace is the
filtered dc system output.
This system can easily resolve a 0.5 ppm change in bridge impedance. Such a change will produce a 3.2 mV change in the
low-pass filtered dc output, well above the RTO drifts and noise.
Figure 19. Gain-of-Two Balanced Modulator Sample
Waveforms
BALANCED DEMODULATOR
The balanced modulator topology described above will also act
as a balanced demodulator if a double sideband suppressed carrier waveform is applied to the signal input and the carrier signal
is applied to the reference input. The output under these circumstances will be the baseband modulation signal. Higher
order carrier components will also be present which can be
removed with a low-pass filter. Other names for this function are
synchronous demodulation and phase-sensitive detection.
PRECISION PHASE COMPARATOR
The balanced modulator topologies of Figures 18a and 18b can
also be used as precision phase comparators. In this case, an ac
waveform of a particular frequency is applied to the signal input
and a waveform of the same frequency is applied to the reference input. The dc level of the output (obtained by low-pass filtering) will be proportional to the signal amplitude and phase
difference between the input signals. If the signal amplitude is
held constant, then the output can be used as a direct indication
of the phase. When these input signals are 90° out of phase, they
are said to be in quadrature and the AD630 dc output will be zero.
1kHz
BRIDGE
EXCITATION
A
1kΩ
AD630
±2 DEMODULATOR
1kΩ
AD524
GAIN 1000
1kΩ
16
B
1kΩ
5kΩ
15
2.5
kΩ
20
1
1MΩ
PRECISION RECTIFIER-ABSOLUTE VALUE
17
If the input signal is used as its own reference in the balanced
modulator topologies, the AD630 will act as a precision rectifier. The high frequency performance will be superior to that
which can be achieved with diode feedback and op amps. There
are no diode drops which the op amp must “leap over” with the
commutating amplifier.
REV. A
2.5
kΩ
10kΩ
A
B
FILTER
5kΩ 5kΩ 5kΩ
13
2
12
C
10kΩ
14
PHASE
SHIFTER
9
10
Figure 21. AC Bridge System
–7–
2µF
2µF
D
2µF
to Figure 18b and is shown in the upper trace of Figure 24. It is
attenuated 100,000 times normalized to the output, B, of the
summing amplifier. A noise signal which might represent, for example, background and detector noise in the chopped radiation
case, is added to the modulated signal by the summing amplifier. This signal is simply band limited clipped white noise. Figure 24 shows the sum of attenuated signal plus noise in the
center trace. This combined signal is demodulated synchronously using phase information derived from the modulator, and
the result is low-pass filtered using a 2-pole simple filter which
also provides a gain of 100 to the output. This recovered signal
is the lower trace of Figure 24.
Figure 22. AC Bridge Waveforms
The combined modulated signal and interfering noise used for
this illustration is similar to the signals often requiring a lock-in
amplifier for detection. The precision input performance of the
AD630 provides more than 100 dB of signal range and its dynamic response permits it to be used with carrier frequencies
more than two orders of magnitude higher than in this example.
A more sophisticated low-pass output filter will aid in rejecting
wider bandwidth interference.
LOCK-IN AMPLIFIER APPLICATIONS
Lock-in amplification is a technique which is used to separate a
small, narrow band signal from interfering noise. The lock-in
amplifiers acts as a detector and narrow band filter combined.
Very small signals can be detected in the presence of large
amounts of uncorrelated noise when the frequency and phase of
the desired signal are known.
The lock-in amplifier is basically a synchronous demodulator
followed by a low-pass filter. An important measure of performance in a lock-in amplifier is the dynamic range of its demodulator. The schematic diagram of a demonstration circuit which
exhibits the dynamic range of an AD630 as it might be used in a
lock-in amplifier is shown in Figure 23. Figure 24 is an oscilloscope photo showing the recovery of a signal modulated at
400 Hz from a noise signal approximately 100,000 times larger;
a dynamic range of 100 dB.
CLIPPED
BAND-LIMITED
WHITE NOISE
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
20-Pin Ceramic DIP (D-20)
C
B 16
AD542
5kΩ
AD630
15
20
19
100R
10kΩ
1 2.5kΩ
17 2.5kΩ
100dB
ATTENUATION
C784a–2–7/84
AD630
A
AD542
13
R
B
100R
20-Pin Plastic DIP (N-20)
14 10kΩ
C
OUTPUT
A
10
0.1Hz
9
MODULATED
400Hz
CARRIER
CARRIER
PHASE
REFERENCE
LOW PASS
FILTER
The test signal is produced by modulating a 400 Hz carrier with
a 0.1 Hz sine wave. The signals produced, for example, by
chopped radiation (IR, optical, etc.) detectors may have similar
low frequency components. A sinusoidal modulation is used for
clarity of illustration. This signal is produced by a circuit similar
PRINTED IN U.S.A.
Figure 23. Lock-In Amplifier
LCC (E-20A)
0.075
(1.91)
REF
0.100 (2.54)
0.064 (1.63)
0.358 (9.09)
0.342 (8.69)
SQ
0.358
(9.09)
MAX
SQ
0.095 (2.41)
0.075 (1.90)
0.011 (0.28)
0.007 (0.18)
R TYP
0.088 (2.24)
0.054 (1.37)
–8–
0.015 (0.38)
MIN
AA AA
20
0.028 (0.71)
0.022 (0.56)
1
0.075
(1.91)
REF
Figure 24. Lock-In Amplifier Waveforms
0.200 (5.08)
BSC
0.100
(2.54)
BSC
0.055 (1.40)
0.045 (1.14)
BOTTOM
VIEW
13
9
0.150
(3.81)
BSC
0.050
(1.27)
BSC
45°
TYP
REV. A