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Digital Object Identifier (DOI): 10.1109/PESC.2008.4592408
Proceedings of the IEEE Power Electronics Specialists Conference (PESC 2008), Rhodes, Greece, 1519 June, 2008.
A ZCS Full-Bridge PWM Converter with Self-Adaptable Soft-Switching Snubber Energy
Qian Sun
Huai Wang
River Tin-Ho Li
Henry Shu-Hung Chung
Saad Tapuchi
Nianci Huang
Adrian Ioinovici
Suggested Citation
Q. Sun, H. Wang, R. T. H. Li, H. S. H. Chung, S. Tapuchi, N. Huang, and A. Ioinovici, "A ZCS full-bridge
PWM converter with self-adaptable soft-switching snubber energy," in Proc. IEEE Power Electron.
Spec. Conf., 2008, pp. 3001-3007.
A ZCS Full-Bridge PWM Converter with SelfAdaptable Soft-Switching Snubber Energy
Qian Sun1, Huai Wang 2, Student Member, IEEE, River T.H. Li2,
Henry S.H. Chung2, Senior Member, IEEE, Saad Tapuchi3,
Nianci Huang1, and Adrian Ioinovici4, Fellow, IEEE
1
Sichuan University, Chengdu, China
City University of Hong Kong, Hong Kong, China
3
SamiShamoon College of Engineering, Beer Sheva, Israel
4
Holon Institute of Technology, Israel
2
Abstract - A new soft-switching full-bridge current-driven
DC-DC converter is proposed for high voltage and high
power applications: its power transistors turn-on/off with
ZCS, what is suitable for IGBT switches. An active snubber
formed by two active switches and one capacitor is inserted
in the primary side of the converter. The
snubber's accumulated energy is proportional to the input
current, what allows ZCS at a very wide line and load range,
by minimizing the required energy: for each value of the
input current and load, the snubber's capacitor is
charged only at the minimum voltage necessary for
achieving ZCS. Thus, the level of the snubber energy
necessary for soft-switching is self-adaptable. The duration
of the resonant stages required for achieving the ZCS
conditions are very short and independent of the values of
the input current and load. No resonant current peaks
appear. The experimental results obtained on a 3 kW
prototype, built for the input voltage of 250V and output
voltage of 15kV, confirmed the expected performances.
I. MOTIVATION
In order to operate the full-bridge (FB) DC-DC
converters at a high switching frequency, with high
efficiency, soft-switching was proposed. A rich
literature of zero-voltage-switched (ZVS) and zerovoltage-zero-current-switched (ZVZCS) converters is
available, of particular interest being those solutions
which realize soft-switching even at light load, by using
snubbers whose energy is load-adaptable, i.e. more
resonant energy is available only when needed at light
load [1]. In ZVS converters, one uses MOSFETs as
primary-side switches. One of the disadvantages of the
ZVS operation is the primary current circulation during
the freewheeling stage, which leads to unnecessary
conduction losses. This problem is solved in ZVZCS
converters, where, by bringing the primary current to zero
during the freewheeling stage, the lagging-leg switches
turn-on/off with ZCS. Consequently, the lagging-leg
switches can be implemented with IGBT transistors,
which can withstand a higher voltage. However, as the
leading leg's switches turn-on/off with ZVS, they have to
be MOSFETs, in order to use their parasitic parallel
_______________________________________________
The work described in this paper was fully supported by a grant from
the Research Grants Council of the Hong Kong Special Administrative
Region, China (Project No.: CityU 112406).
978-1-4244-1668-4/08/$25.00 ©2008 IEEE
capacitance. This limits the use of the ZVZCS converters.
In high-input voltage applications, such as medical X-ray
imaging equipment, traveling wave tubes, RF generation,
it is preferable to use only IGBTs, or other minoritycarrier devices for all the primary switches ,as they can
withstand the high line voltage. The best soft-switching
way for these transistors is ZCS.
Very little research has been devoted to FB-ZCS
converters: in [2] and [3], ZCS is obtained in currentdriven converters, by using a passive snubber [2], or an
active one [3]. Boost-type input FB converters have many
applications, for example in high-power single-phase
PFC designs as that described in [4] for an electric
vehicle inductive charging, etc. ZCS was obtained due to
a resonance process in a snubber formed by a resonant
inductance Lr, which includes the leakage transformer
inductance, and a resonant capacitor Cr, which includes
the reflected winding capacitance. ZCS is lost at high
input current. In order to extent the ZCS range, the
characteristic impedance of the resonant tank has to be
decreased, either by decreasing Lr, or by increasing Cr.
The first solution leads to an increasing current stress in
the primary switches, the second one leads to an increase
of the duration of Cr discharge interval, and consequently
to a loss of duty-cycle.
Voltage-driven FB ZCS converters are proposed in [59], using snubbers inserted in the primary side [5, 7, 8] or
in the secondary side [6, 9]. In [5] and [8], two auxiliary
resonant active snubbers are used. In [5], a resonant cell,
formed by a transistor, an inductor, a capacitor and a
diode, is placed in parallel with each lower primary-side
switch. For turning- off the lower switch with ZCS, the
snubber’s transistor is turned-on with ZCS, and the
primary current is deviated to the auxiliary path. However,
the auxiliary switch is then turned- off with hardswitching. A similar cell is placed in parallel with each
upper switch in [8], creating ZCS for that switch. The
lower switches turn with ZCS without the help of the
resonant cells. ZCS is lost at high input current. The
rectifier diodes are submitted to double the output voltage.
In [7], a single snubber formed by two uni-directional
transistors, a resonant inductor and a resonant capacitor is
used. The primary current is brought to zero following a
resonant process in the snubber, the resonant current must
reach a value higher than the primary current. The
resonant capacitor voltage also reaches a value larger
3001
than the input voltage. In these circuits, the interaction
problem between the leakage inductance and the rectifier
parasitic capacitances is not solved.
The secondary-side solutions make use of a single
resonant snubber, which contains a uni-directional
transistor, an inductor, a capacitor and a diode. Soft
commutation for the rectifier diodes is assured, such that
parasitic oscillations are avoided. Compared with the
solution presented in [6], in [9] the reverse recovery
problem of the anti-parallel diode of the auxiliary switch
is solved. As the resonant inductor current has to
overpass the output current in order to bring to zero the
primary current, the secondary resonant current peak
implies a high peak in the primary current, which leads to
extra conduction losses in the primary-side switches. The
voltage stress across the rectifier diodes reaches three
times the output voltage at light load condition (when the
resonant capacitor voltage reaches two times the output
voltage). ZCS is lost at low line voltage and heavy load
current, as then the resonant inductor current could not
reach the output current in a reasonable time duration, i.e.
without a serious duty-cycle loss.
In all available solutions, the energy used for getting
soft-switching is independent of the line and load values.
A trade-off between the ZCS range and duty-cycle loss
has to be accomplished, meaning a loss of the ZCS at
certain line voltage or load variation end. Current and/or
voltage resonant peaks lead to additional conduction
losses.
This paper proposes a new snubber that is inserted in
the primary side of a current-driven FB converter. ZCS of
the power switches is obtained at a very wide line and
load range, starting from a low value of the input current,
by always using the minimum necessary energy. The
snubber's transistors are turned on/off with ZVS. Section
II presents the circuit and its switching operation. Section
III is devoted to a DC analysis. The ZCS conditions are
discussed, conducting to the design of the circuit. The
self-adjustability of the ZCS-purpose energy accumulated
in the snubber is explained. The control circuit is
described in Section IV, followed by experimental results.
II. PROPOSED ZCS FB CONVERTER WITH ADAPTIVE
SOFT-SWITCHING ENERGY
The current-driven FB converter is shown in Fig. 1.
The snubber is formed by two uni-directional transistors
Sa1, Sa2, and one capacitor Cr. The switching diagram is
given in Fig. 2, and the switching topologies in Fig. 3.
ip(t) is the primary current and vCr is the voltage across Cr.
1st Topology [t0, t1] [Fig. 3(b)] – Before t0, the circuit
topology is shown in Fig. 3(a). The input energy is
transferred to the load via diode D2. At t0, a new cycle
begins by turning-off Sa1 with ZVS. The energy transfer
continues (this Mode does not lead to a loss of dutycycle), and Cr is charged
Let’s define
i p (t )
I in
(1a)
vCr (t )
I in
(t t0 )
Cr
(1b)
VCr '
vCr (t1 )
I in
(t1 t0 )
Cr
CrVCr'
I in
t 01
(1c)
The duration of this topology is determined as
explained in Section III in order to obtain the
minimum capacitor voltage vCr (t1 ) necessary to achieve
ZCS for each value of the line and load currents. ZCS is
obtained even at high input current, without increasing
unnecessary the capacitor's accumulated energy at lower
input current, i.e., the snubber's accumulated energy is
self adjustable depending on the value of the input (and
implicit load) current.
2nd Topology [t1, t2] Fig. 3(c) - At t1, S3 turns on with
ZCS, the primary current starts reducing.
i p (t )
Llk ,
Z
Cr
Zp
vCr (t )
Iin cos Zt 1
,n
Llk Cr
n Vo VCr '
sin Zt
Zp
(2a)
N p / Ns .
I in Z p sinZ t t1 VCr' nVo cosZ t t1 nVo (2b)
The topology ends when the primary current drops to
zero, giving the duration of this stage
1
t12
vCr (t 2 )
Z
V
'
Cr
tan 1
I in Z p
(2c)
VCr' nVo
2
2
nVo I in Z p nVo
(2d)
According to (2a), at a large Iin, one needs a larger VCr'
to bring to zero the primary current. At a low Iin, a
smaller VCr' is needed. Consequently, the duration of the
first and second intervals is only slightly dependent on
the value of Iin, no loss of duty-cycle arises from the large
ZCS range.
3rd Topology [t2, t3] Fig. 3(d) - At t2, ip reached zero, S4
is switched off with ZCS .As a result, the secondary
current reaches zero, and the rectifier diode turns-off
naturally (ZCS).The load voltage is assured by the output
capacitor (freewheeling stage)
i p (t )
th
0
(3a)
vCr (t )˙vCr (t2 )
(3b)
vCr (t3 )˙vCr (t2 )
(3c)
4 Topology [t3, t4] Fig. 3(e) - The PWM will dictate
the instant t3 when S2 is turned on with ZCS. The Cr
3002
energy is transferred to the load. ip goes negative and
increases in absolute value , the presence of Llk in the ip
path assures the turn-on of the rectifier diode D1 with
ZCS. The current through S1, iS 1 (t ) I in i p (t ) ,
decreases
v (t ) n V o
s inZ
Cr 3
Z p
i p (t )
t
t3
(4a)
(4b)
vCr (t ) nVo V nVo cosZ t t3 '
Cr
The stage ends when the primary current reaches the
input current, giving the duration of this topology
Ts / 2
Ts / 2
DTs
S1
S2
DTs
S3
S4
S a1
Sa 2
vCr
VCr'
'
1
t34
Z
(4c)
I in Z p
sin 1
vCr (t 2 ) nVo
(4d)
vCr (t4 ) nVo VCr' nVo cosZ t4 t3 VCr
ip
I in
I in
io
nI in
th
5 Topology [t4, t5] Fig. 3(f) - At t4, ip reaches -Iin, the
current through S1 dropped to zero, so S1 turns-off with
ZCS. The energy is transferred from the line to the load.
Cr is discharged to the load, thus recuperating the energy
i p (t )
vCr (t )
vCr (t 4 ) t45
(5a)
I in
(5b)
I in
t t4 Cr
(5c)
Cr vCr (t 4 )
I in
6th Topology [t5, t6] Fig. 3(g) - At t5, Cr is completely
discharged, Sa2 turns on with ZVS, and the circuit
operates in this transfer-energy mode until a new halfcycle begins by turning off Sa2
i p (t )
vS1
nVo
iS 1
I in
vS 3
nVo VCr'
nVo
iS 3
I in
vSa1
VCr
iSa1
I in
t 0 t1 t2
(6b)
vCr (t ) 0
It can be noticed that the primary current in all the
stages never overpasses the input current, keeping thus
the conduction losses in the primary-side switches at their
nominal value.
S1
S2
NP : Ns
S a1
S1
NP : Ns
S a1
S3
Co
Llk
Vin
Cr
D1
R
S3
S4
S4
Sa2
D2
Cr
(a) Before t0
Sa 2
Fig. 1 Circuit diagram
3003
D1
Vo
ip
Llk
I in
Lin
S2
t9 t10 t11 t12
Fig.2 Timing diagram for the converter
(6a)
I in
t3 t4 t5 t6 t7 t8
D2
S1
S2
NP : Ns
ip
Llk
I in
*
Vo
*
NP : Ns
Cr
S3
Sa2
Sa2
(b) [t0, t1].
(f) [t4, t5].
S2
NP : Ns
S2
*
*
Vo
D2
Cr
D2
S4
S3
+ -
Sa2
+ -
S4
*
Vo
D1
*
ip
Llk
I in
Cr
S3
NP : Ns
S a1
*
S a1
D1
*
ip
Llk
I in
D2
+ -
S4
S3
S1
S1
*
Vo
Cr
+ -
S4
*
S a1
D2
D1
*
ip
Llk
I in
*
S a1
S2
S1
D1
Sa2
(g) [t5, t6].
Fig.3 Modes of operations
(c) [t1, t2].
S1
S2
I in
NP : Ns
*
ip
Llk
*
S a1
*
III. DC ANALYSIS OF THE PROPOSED CONVERTER
D1
In order to insure soft-switching of the primary
switches, two conditions have to be fulfilled: in Stage 2,
ip has to drop to 0, and in Stage 4, ip given by (4a) has to
reach - Iin. Taking into account (2d), which guarantees the
fulfillment of the Ist condition, and (3c), from (4a) it
results
D2
Cr
S3
+ -
S4
3.1 ZCS Conditions
Vo
Sa2
V cr'
(d) [t2, t3].
S1
S2
NP : Ns
ip
Llk
I in
*
S a1
Cr
S3
S4
+ -
*
2
Z p 2 nV o I in Z p nV o
1
1
C r ( vCr ( t 3˅ nV o ) 2 t Llk I in2
2
2
Vo
D2
2
in
(7)
It can be noted that to say that ip in (4a) has to reach
- Iin is equivalent to saying that
D1
*
I
‫ق‬8‫ك‬
i.e. there is enough energy in Cr for decreasing the
primary current from 0 to - Iin . It is possible to see from
(7) that VCr' depends on the value of Iin, i.e. the resonant
energy used to get ZCS is self-adaptable.
Sa2
3.2 DC Conversion Ratio
(e) [t3, t4].
A phase-shift control is used. The energy transfer is
controlled by the delay between S1 and S3 in the first halfcycle, and S2 and S4 in the second half-cycle. By
assimilating the shift angle with a duty-cycle D, and
similar with the boost converter where the on-topology is
characterized by the charging of the input inductor, one
can define
3004
DTs
t12 t 23 t34
0.05
(9)
In order to get the formula of the DC conversion ratio,
a current-second balance written for a half- cycle is
obtained
6
³
0.035
tk
tk 1
0.005
0
0
15
Iin (A) 20
(Vo=15kV, Llk=10uH, n=1/40, f=20kHZ)
0
200
IS 4
Cr I in Z p
IS 5
C r nV o
160
(1 2 D )T s
I in C rVCr' C r nVo
2
140
180
Hard-switching FB
boost converter
M
120
100
­°
ª
Cr nVo 2 I in Z p VCr' º ½°
»¾
nI in ®1 2 « D I inTs
«
»¼ ¿°
¬
¯°
80
­
ª
§
Cr ¨ 2nVo 2 I in Z p °°
«
nI in ®1 2 « D ©
«
°
«¬
¯°
40
Proposed
converter
60
2
2 · º½
Z p 2nVo I in Z p ¸ » °
¹ »°
¾
I inTs
»°
»¼ ¿°
I
10
Fig.4. Duty cycle loss as a function of Iin
which replaced in (10), together with (9), give
Io
5
C r I in Z p nVo VCr' IS3
IS 6
Cr=33pF
Cr=22pF
0.01
CrVCr'
IS1
Cr=100pF
Cr=68pF
0.015
into
IS 2
0.025
0.02
(10)
Ip(t ) dt
account that t23˙DTs t12 t34 ,
t56 Ts / 2 DTs t01 t45 , and using 1(a, c) - 5(a, c) and
(7), one gets
Taking
Cr=220pF
0.03
Dloss
k 0
Cr=470pF
0.04
Ts
Io
2
n¦ IS k
ISk
0.045
in
0
0.1
0.2 D 0.3
0.4
0.5
Fig.5 DC voltage conversion ratio as a function of duty-cycle
(Vo=15kV, Llk=10uH, Cr =100pF, Iin=12A, n=1/40, f=20kHZ)
M
Vo
Vin
1
­
ª
§
Cr ¨ 2nVo 2 I in Z p °°
«
n ®1 2 « D ©
«
°
°¯
¬«
2
2 · º½
Z p 2nVo I in Z p ¸ » °
¹ »°
¾
I inTs
»°
¼» °¿
I
in
(11)
975
Comparing with the conversion ratio formula for a
boost hard-switching FB: Vo = 1/ [n (1-2D)], it results
that the ZCS-purposed intervals lead to a loss of
regulation capability (i.e. DTs cannot be reduced to zero,
if so required for regulation purpose, but its lower limit is
t12+t34 ). This loss of regulation is expressed by Dloss
§
Cr ¨ 2nVo 2 I in Z p Dloss
©
I
2
in
2
Vcrmax (V)
775
675
575
·
475
¹
375
Z p 2nVo I in Z p ¸
I inTs
Llk=5uH
Llk=10uH
Llk=15uH
Llk=20uH
875
and it is represented in Fig.4. It can be noticed that Dloss
increases with the value of Cr, however it has a very
small value.
Eq. 11 is represented in Fig. 5 as a function of D, for
Llk=10uH, Cr=100pF, Iin=12A, n=1/40, f=20kHZ.
According to Eq. (5c), ZVS of turning-on of the
auxiliary switch is lost at very low input current.
0
400
600
Fig.6. vCr,max as a function of Cr for different Llk
( Vo=15kV, Iin=12A, n=1/40, f=20kHZ)
maximum value of the voltage across Cr, vCr,max , one gets
Cr t
3.3 Circuit Elements Design
Cr is designed such that to get ZCS in the 4th topology
at the largest value of the input current, I in , max , i.e.,
according to (8), and taking into account that vCr(t3) is the
200
Cr (pF)
Llk I in,max 2
(vCr ,max nVo )2
(12)
As the voltage stress on the auxiliary switch, VSa , max
is given by vCr,max, Cr is designed such that to limit this
stress to 1.35 times of nVo. In Fig. 6, vCr,max is represented
as a function of Cr and Llk..
3005
Fig. 7 Block diagram of the control system
The second design condition comes for limiting the
resonant period Tr to less than 15% of Ts
Tr
2S
The experimental waveforms are shown in Fig.
9‫؜‬it can be seen that the primary transistors S1, S3
turn-on/off at zero current, and the snubber’s
transistors turn-on/off with ZVS. Experimentally,
the efficiency was measured at 89.5 %
Llk C r
(13)
Accordingly, Llk=10uH, Cr=100pF, giving vCr',
calculated for the maximum input current, equal to
486.7 V.
Main switches. The primary switches have to
withstand a maximum stress voltage given by vCr(t2)
plus nVo, the maximum current through these
switches is Iin. For Vo=15kV, Iin,max =12A, it results
Vsmax=870V, Ismax=12A. Transistors of type IGBT
IRG4PH50UD have been chosen (VCES=1200V,
Ic=24A)
Auxiliary switches. These switches turn-on/off
with ZVS, and have to withstand the voltage vCr(t2)
calculated for the upper end of Iin range, the
maximum current through them being the input
current. Vsa,max=495V, Isa,max=12A. Transistors of
type MOSFET SPW20N60C3 have been chosen
(VDS=650V, ID=20.7A).
D1
D2
C
D3
IV. SELF-ADAPTIVE CONTROL CIRCUIT AND
EXPERIMENTAL RESULTS
Figure 7 represents the block diagram of the
control circuit. A sensor measures Iin (which is
variable due to changes in the supply or load). An
analogue device calculates VCr ' according to (7) by
using the measured value of Iin. A sensor measures
vCr. When the measured voltage of the resonant
capacitor reaches the calculated value, S3 is turned
on, thus practically determining the instant t1 as a
function of the input current / load. Consequently,
the energy accumulated in the snubber is the
minimum one necessary to get ZCS for the actual
input/load current. The maximum vCr(t1) for getting
ZCS is reached at the upper limit of the range of
the input current. This value is used when selecting
the switches according to the voltage stress they
have to withstand. At any other input/load current,
the actual stress is smaller.
A prototype was built for Vin = 250V, Vo =
15kV, output power Po = 3kW, f = 20kHz, Cr
=100pF, Llk = 10PH. To ensure safe operation of
the rectifier, ten modules are employed to make up
the secondary-side rectification circuitry, as
illustrated in Fig.8.
D4
Fig.8. Schematic of the rectifier stage
Gate signal of Sa1
Gate signal of Sa2
vcr
Gate signal of Sa1 and Sa2: 20V/div. vcr: 500V/div.
Timebase: 20Ps/div
3006
achieving soft-switching, is self-adaptable: it is
larger when needed (at high input current). The
design allows using the minimum necessary energy
to get ZCS, such that the actual voltage stress on
the switches is kept at the minimum. Unlike the
available solutions, no resonant current peaks
appear, never the current through the switches
overpasses the input current. The durations of the
resonant stages required to achieve the ZCS
conditions are short. The soft-switching conditions
are achieved for a very wide line and load range,
starting from a low value of the input current. The
experimental results confirmed the good features of
the proposed solution, an efficiency of 89.5 %
being measured.
vS1
iS1
REFERENCES
vS1: 500V/div. iS1: 20A/div. Timebase: 10Ps/div
[1] Y. Jang, M.M. Jovanovic, and Y.M. Chang, “A new ZVS PWM full-bridge converter,” IEEE Trans. Power Electron., vol.
18, pp. 1122-1129, Sept. 2003.
[2] C. Iannello, S. Luo and I. Batarseh,“Small-signal and
transient analysis of a full-bridge, zero-current-switched PWM
converter using an average model,” IEEE Trans. Power
Electron., vol.18, pp. 793-801, May 2003.
[3] A. Leung, H. Chung, and K. Chan,“A ZCS Isolated FullBridge Boost Converter with Multiple Inputs,” in Proc. IEEE
Power Electron. Spec. Conf., 2007, pp. 2542-2548.
[4] R.Watson and F.C.Lee, “A soft-switched, full-bridge boost
converter employing an active- clamp circuit,” in Proc. IEEE
Power Electron. Spec. Conf., 1996, pp.1948-1954.
[5] M. Marx and D.Schroder, "A novel zero-curent-transition
full-bridge dc-dc converter," in Proc .IEEE Power Electron.
Spec. Conf., vol. 1, 1996, pp. 664-669.
[6] D.M. Xu, X.H. Wu, J.M. Zhang, and Z. Qian, "High power
high frequency half-wave-mode ZCT-PWM full bridge DC/DC
converter," in Proc. IEEE Appl. Power Electron. Conf., APEC
2000, pp. 99-103.
[7] X.H. Wu, D.M. Xu, J.H. Kong, C. Yang, and Z. Qian, "High
power high frequency zero current transition full bridge DC/DC
converter," in Proc. IEEE Appl. Power Electron. Conf., vol.2,
APEC 1998, pp.823-828.
[8] D.Y. Lee, M.K. Lee, D.S. Hyun, and I. Choy, “New zerocurrent-transition PWM DC/DC converters without current
stress,” IEEE Trans. Power Electron., vol. 18, pp. 95-104, Jan.
2003.
[9] J. Zhang, X. Xie, X. Wu, G.Wu, and Z. Qian, "A novel
zero-current-transition full bridge DC/DC converter," IEEE
Trans. Power Electron., vol. 21, pp.354-360, Mar. 2006.
vS3
iS3
vS1: 300V/div. iS1: 20A/div. Timebase: 20Ps/div
Fig. 9 Experimental results
CONCLUSIONS
A new soft-switching high-input voltage highpower full-bridge DC-DC converter was proposed.
Its primary switches turn-on/off with ZCS, the
snubber's switches turn-on/off with ZVS. The
rectifier diodes turn-on/off with ZCS. The energy
accumulated in the capacitor's snubber, used for
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