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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 9, SEPTEMBER 2008
Digital Control of a Low-Frequency Square-Wave
Electronic Ballast With Resonant Ignition
F. Javier Díaz, Francisco J. Azcondo, Senior Member, IEEE, Rosario Casanueva, Member, IEEE,
Christian Brañas, Member, IEEE, and Regan Zane, Senior Member, IEEE
Abstract—This paper proposes a two-stage low-frequency
square-wave (LFSW) electronic ballast with digital control. The
first stage of the ballast is a power factor correction (PFC) stage,
and the second is a full-bridge (FB) converter used for both lamp
ignition and LFSW drive. As a novelty for LFSW ballasts, ignition
is achieved without an additional igniter circuit by operating the
FB during start-up as a high-frequency resonant inverter. After
ignition, the converter operates as an LFSW inverter to avoid
exciting acoustic resonances by controlling the FB as a buck converter and regulating alternately positive or negative current to the
lamp. Lamp power is regulated by adjusting the average current
supplied by the PFC stage. Another contribution of this paper is
to utilize digital control as a simple solution to achieve multimode
control, including resonant lamp ignition, LFSW transitions, and
lamp current and power regulation.
Index Terms—Acoustic resonance, digital control, dimming
control, low-frequency square-wave (LFSW) converter, metal
halide (MH) lamps, resonant ignition.
I. I NTRODUCTION
T
HE PRIMARY motivation for using a low-frequency
square-wave (LFSW) drive in electronic ballasts is to
avoid the excitation of acoustic resonances in metal halide
(MH) lamps. The MH lamp has become very popular as a
practical light source for general and specific applications due
to its high efficacy, compact design, and superior color rendering properties. LFSW electronic ballasts are an alternative
to resonant converters and, theoretically, provide a definitive
solution to prevent acoustic resonance, provided that the lamp
power has no ac component. The frequencies at which acoustic
resonances appear depend on the size of the arc tube, gas
pressure, and its composition, and they may vary with the lamp
aging. The elements that compose the gas enclosed in the vessel
determine the lamp chromatic rendering; in this way, a more
complete spectrum of the light source requires a more complex
composition of the lamp gas that presents more resonant modes.
Manuscript received February 8, 2008; revised June 17, 2008. First published
July 9, 2008; last published August 29, 2008 (projected). This work was
supported in part by the Spanish Government under project CICYT TEC200801753: “Digital power processing for the control of gaseous discharges” and in
part by the National Science Foundation under project “CAREER: Modeling,
Control, and Design of Energy-Efficient Lighting Systems.”
F. J. Díaz, F. J. Azcondo, R. Casanueva, and C. Brañas are with the
Department of Electronics Technology, Systems and Automation Engineering,
University of Cantabria, 39005 Santander, Spain (e-mail: diazrf@unican.es;
azcondof@unican.es; casanuer@unican.es; branasc@unican.es).
R. Zane is with the Department of Electrical and Computer Engineering,
University of Colorado, Boulder, CO 80309-0425 USA (e-mail: regan.zane@
colorado.edu).
Digital Object Identifier 10.1109/TIE.2008.927959
Standard solutions for LFSW high-intensity discharge (HID)
lamp drivers over 100 W require three power conversion stages
plus a lamp igniter circuit, as shown in Fig. 1(a). The three
converter stages include the following: 1) a power factor correction (PFC) stage [1], [2]; 2) a current-mode-controlled dc–dc
converter; and 3) a full-bridge (FB) inverter. The additional
lamp igniter circuit is required to achieve a sufficiently high
voltage for lamp ignition [3], [4]. A two-stage solution with
an external igniter circuit is presented in [5] where the lamp
current control provides the required system stability.
The proposed two-stage solution with integrated lamp ignition is shown in Fig. 1(b). All three functions from Fig. 1(a)
of stages 2 and 3 and the igniter circuit are integrated in a
single FB stage. Lamp ignition is achieved without an additional igniter circuit by operating the FB during start-up as a
high-frequency resonant inverter. After ignition, the converter
operates as an LFSW inverter by controlling the FB as a
buck converter and supplying alternately positive or negative
current to the lamp. Lamp power is regulated by adjusting the
average current supplied by the PFC stage [6]. A simple digital
microcontroller is used to achieve multimode control, including
resonant lamp ignition, LFSW transitions, and lamp current and
power regulation. Sampled signals from a single current sense
resistor Rg are used in LFSW mode as inputs to two key control
loops, as shown in Fig. 1(b). A fast proportional inner current
control loop regulates the buck converter inductor current to
provide stability to the lamp, and a slow power regulation loop
maintains the steady-state lamp power.
The operating modes of the converter and the controller are
described in Section II, followed by design constraints and
guidelines for the LC filter in the FB converter in Section III.
Converter models and controller design are provided in
Section IV. Experimental results are presented in Section V,
demonstrating complete ballast system operation driving a
150-W MH lamp.
II. B ALLAST AND C ONTROLLER O PERATING M ODES
The inverter switching frequency is denominated fisw , and
the square output voltage is symmetric, 50% positive and
negative, in the different modes. The converter operation is
summarized in the flowchart of the dsPIC30F2010 program in
Fig. 2. The microcontroller program starts with the ignition
sequence until the lamp turns on, and then, the converter is
operated in resonant inverter mode during the warm-up time.
Finally, it changes to LFSW mode with double loop control.
The FB has two different operation modes.
0278-0046/$25.00 © 2008 IEEE
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JAVIER DÍAZ et al.: DIGITAL CONTROL OF A LOW-FREQUENCY SQUARE-WAVE ELECTRONIC BALLAST
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Fig. 1. Square-wave electronic ballast. (a) Design with three stages. (b) Proposed design where stage 1 is a power factor corrector stage and stage 2 is an FB that
works as a resonant or an LFSW inverter according to the state of the lamp.
A. FB Converter Operation as Resonant Inverter
The bridge converter, the inductor L, and the capacitor C
form a parallel resonant inverter before the lamp ignition.
During the ignition sequence, resonant inverter operation is
obtained, and, in this mode, fisw is higher than the resonant frequency fo of the LC filter [see Fig. 1(b)] when the lamp is off,
and it gradually approaches fo , where the voltage gain is high
enough to produce the discharge. The microcontroller generates
the transistor drive signals shown in Fig. 3. The lamp ignition
occurs above the unloaded resonant frequency of the LC circuit
√
(1)
fo = 1/2π LC.
One benefit of sweeping the frequency for ignition is that the
lamp voltage rate of rise is properly controlled, and as a result,
the required overvoltage to achieve the ignition is reduced
[7]–[12].
Part of the lamp warm-up is also carried out using the highfrequency inverter mode and is achieved during resonant converter operation at constant lamp current. During ignition and
warm-up, the duty cycle control is disabled, and the converter
operates as a traditional FB resonant inverter.
B. FB Converter Operation as LFSW Ballast
After lamp ignition and warm-up, the control circuit enters in
LFSW mode, where the bridge converter is driven alternately as
a positive and negative buck converter (see Figs. 4–6). The buck
converter switching frequency fsw = 1/T fo is constant.
Control signals of the FB transistors for the buck operation
modes are defined in Fig. 4.
In the LFSW mode, the duty cycle d = ton /T is regulated to
control the inductor current i and stabilize the lamp current in
the short term, and to regulate ig in the long term for power
adjustment and dimming control if required.
The inductor L and the capacitor C define the converter lowpass filter that limits the ac component of the lamp power below
the level that can excite acoustic resonance. Current mode
control is achieved by regulating the inductor current i, which is
the PFC output current ig during dT , sensed by Rg (see Figs. 5
and 6). The inductor current i circulates, as shown in Fig. 5,
with the continuous line during dT time and the discontinuous
line during (1 − d)T time. In addition, Fig. 6 shows the positive
buck (positive lamp current) and negative buck (negative lamp
current) operation modes.
Since the buck converter input current ig is pulsating, as
shown in Fig. 6, during on time (dT ), the inductor current is
sampled at Rg (i∗ ) and is multiplied by d to obtain the sampled
average input current i∗g . The inner high-speed current loop
provides the ballast action since it stabilizes the lamp current,
which is assumed to be equal to the average inductor current i,
resulting in high output impedance for the inverter. The outer
power loop provides the current reference iref to the inductor
current loop to achieve the designed lamp power. The power
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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 9, SEPTEMBER 2008
Fig. 2. Control program flowchart.
Fig. 4. Control signals of the FB transistors. (a) For the positive lamp current
(positive buck). (b) For the negative lamp current (negative buck).
Fig. 3. FB transistor drive signals during start-up sequence and warm-up of
the lamp.
sample data is i∗g , provided that the PFC output voltage Vg is
considered constant (see Fig. 5). The power mode control does
not necessitate high-speed performance [13]–[15].
The generation of open-loop transitions from positive to
negative current through the lamp and vice versa is proposed
to achieve the maximum transition speed. In this way, there is
no reignition effect even at very low fisw . Since the duty cycle
does not change, the transition response depends on the quality
factor of the filter LC loaded with the lamp. The control circuit
detects the end of the transition to retake the current and power
mode control.
III. LC F ILTER D ESIGN
The LFSW converter operation generates an ac component
whose amplitude in steady state at the fundamental (switching)
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JAVIER DÍAZ et al.: DIGITAL CONTROL OF A LOW-FREQUENCY SQUARE-WAVE ELECTRONIC BALLAST
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Therefore, the resulting amplitude of the ac component of the
lamp power at the switching frequency is
P̂ac,fsw =
2DVg V̂ac,fsw G (jfsw )
.
Rlamp
(5)
To prevent the excitation of acoustic resonant modes, the
limitation
P̂ac,fsw < kPlamp
(6)
is imposed.
This limitation is based on the acoustic resonance studies
in [16] and [17], where a typical value for the parameter k to
prevent the excitation of acoustic resonance under any condition is k < 5%. The requirement for the low-pass filter is then
derived from
V̂ 2
G(jfsw (DVg )2 + ac,fsw 2
2DVg V̂ac,fsw G(jfsw <k
Rlamp
Rlamp
(DVg )
∼
=k
Rlamp
2
Fig. 5. Input (ig ) and inductor (i) current direction, in continuous line during
dT time and in discontinuous line during (1 − d)T time. (a) Positive buck.
(b) Negative buck.
(7)
and, using (2), leads to
G(jfsw ) <
kπD
.
4 sin(Dπ)
(8)
As an illustrative example, the LC filter design for a converter that drives a 150-W HID lamp is presented.
Fig. 6. Steady-state current waveforms. Above: Inductor current. Below: PFC
front-end output current. (a) Positive buck. (b) Negative buck.
frequency is
V̂ac,fsw =
2Vg
sin(Dπ).
π
(2)
The low-pass filter gain at the switching frequency
G(jfsw ) reduces the ac component of the lamp voltage,
which generates ac power amplitude supplied to the lamp
at twice the switching frequency. By using the fundamental
approximation
plamp =
DVg + V̂ac,fsw sin(ωsw t) G(jfsw )
2
(3)
Rlamp
where the magnitude of the buck filter response at the switching
frequency is given by
G(jfsw ) = 1−
fsw
fo
1
2 2
+
1
Q2
fsw
fo
2
.
(4)
Step 1) Collect input data. PFC output voltage, i.e., LFSW
converter input voltage and lamp equivalent resistance data are collected. Vg = 400 V, which is the
common value for a boost-based PFC, and the equivalent lamp resistance ranges from Rlamp,min = 50 Ω
(new lamp) to Rlamp,max = 150 Ω at the end of its
lifetime.
Step 2) Estimate the duty cycle. The required duty cycle to
supply the rated lamp power is calculated. In this
case, the duty cycle varies from D = 0.22 (new
lamp) to D = 0.38 (old lamp).
Step 3) Determine filter gain. For the specified k < 5%, the
filter gain is calculated resulting in G(jωsw ) <
−37.4 dB for D = 0.22 and G(jωsw ) <
−35.9 dB for D = 0.38. A practical solution for
(8) considering the lamp lifetime is G(jωsw ) <
−40 dB leading to ωsw = 10 ωo .
Step 4) Select the filter quality factor. Inductor and capacitor values are chosen to obtain the characteristic
impedance
Z=
L
C
(9)
which limits the resonant current during the ignition
sequence, and a quality factor Q that results in a
fast and nonoscillatory transient from positive to
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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 9, SEPTEMBER 2008
Fig. 7. Gain (in decibels) of the double-pole low-pass filter around fo for different Rlamp ’s (50, 75, 100, and 150 Ω).
Fig. 8. Block diagram of the proposed digital power control.
negative buck converter and vice versa. The practical
limit of the quality factor is
Qmax =
Rmax
.
Lωo
(10)
Qmax = 1.2 is selected in order to limit the lowfrequency transient oscillations.
Step 5) Select the switching frequency. A practical upper
limit for the switching frequency is ωsw = 2π ·
200 krad/s to avoid excessive switching losses,
which results in ωo = 2π · 20 krad/s.
Step 6) Calculate filter components. By using (10),
L = 995 µH, and with (9), C = 63.6 nF.
A simulation of this case is shown in Fig. 7.
IV. C ONVERTER M ODELING AND C ONTROLLER D ESIGN
As aforementioned, the implementation of a suitable regulation of the lamp should solve the ballast performance in the
short and long terms. The proposed solution is to use a fast
current mode control that stabilizes the lamp arc and a slow
power mode control that regulates the nominal lamp power and
implements dimming control [18], [19], as shown in Fig. 8.
Since i = ig during dT (on time), the inductor current can
be captured at Rg [see Fig. 1(b)] for the current loop. The
power control variable is the average input current ig , which
is calculated by multiplying the dc bus current sampled during
the on time i∗ by the duty cycle d, giving i∗g .
By considering the buck converter and assuming that the
ripple of the PFC output voltage Vg is negligible, the averaged
model equation is described by
dVg − νTs
= iTs
sL
(11)
where the average inductor current iTs is the variable to
control to achieve the current mode control. The resulting
averaged small-signal model is given by
ˆ g − ν̂
dV
= î.
sL
(12)
Equation (12) analyzes the effects of the perturbations of the
duty cycle and output voltage on the inductor current i. The
desired small-signal control transfer function from duty cycle
to inductor current is derived introducing (13) in (12) and is
given by
ν̂ = î
Gid (s) =
zlamp
1 + sCzlamp
(13)
1 + sCzlamp
Vg
î
=
dˆ zlamp 1 + s z L + s2 LC
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lamp
(14)
JAVIER DÍAZ et al.: DIGITAL CONTROL OF A LOW-FREQUENCY SQUARE-WAVE ELECTRONIC BALLAST
Fig. 9.
3185
Experimental small-signal response with the frequency of 150-W HID lamp. The gain is in continuous line, and the phase is in dashed line.
Fig. 10. Estimated Bode plot of Gid (s). Above: gain in decibels. Below:
phase.
where zlamp is the lamp incremental impedance [20]–[22].
The experimental small-signal response of 150-W HID lamp
is shown in Fig. 9. The gain and phase of the transfer function
given in (14) is shown in Fig. 10. This model, where a resistive
load is considered, is valid in a rather restricted range of frequencies. The continuous averaged model results in a transfer
function that fits the discrete model in a frequency range up
to fsw /30 with no significant error in gain and phase. On the
other hand, due to the lamp behavior, the frequency range where
zlamp behaves as positive impedance is minimum [22]. For the
case presented, it is considered that Gid in (14) is valid from
approximately 3 to 10 kHz, i.e., between the dashed lines in
Fig. 10, which is also the target frequency area to locate the
crossover frequency of the current control loop gain.
For the power mode control, it is considered that
Plamp = ηlfsw · Vg · ig ig = d i
(15)
(16)
Fig. 11. Top: measured input current νsen = ig Rg . Middle: input current ig .
Bottom: transistor drive signal νgs .
where ηlfsw is the efficiency of the LFSW converter, and the
reference for the average inductor current iref (see Fig. 2) is the
variable under control for power mode operation.
In Fig. 11, the transistor drive signal νgs , the input current ig ,
and its sample Rg ig = νsen are shown. Provided that the buck
converter is operating in continuous conduction mode with
small ∆i, one sample of i during dT is valid to obtain the
average inductor current with a sufficiently small error.
Controlled transition from positive to negative lamp current
and vice versa may lead to a slow transient that might produce
reignition, flickering effect, and low-frequency harmonics [23].
As an example, in [24], a pulsewidth modulation (PWM) controller modulates the output current with a square wave whose
rising time is several times that of the buck converter switching
period. In this paper, to achieve the fastest transient, an openloop transition is proposed.
As L and C form a low-pass filter loaded with Rlamp ,
the step response depends on the quality factor. If the quality
factor Q is between 0.4 and 1.2 (see Fig. 7), the desired output
voltage step response, around the critical damping, is obtained.
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Four operation modes are distinguished according to the type
of the drive signals generated for the switches.
A. Soft Start-Up Ignition Sequence Mode
The frequency sweep from 100 to 23 kHz is performed for
the FB resonant inverter. When the lamp is off, the FB switches
Vg at a high frequency, the lamp voltage ν is sinusoidal, and
the switching frequency fisw is reduced to approach fo until
ν reaches the ignition voltage. In the case of hot restrike or
simply when the ignition fails, the frequency sweep is restarted
before excessive overvoltage is reached, and the sweep is
repeated until the lamp ignition is achieved. The sweep is until
23 kHz to avoid an output voltage higher than 1600 V during
lamp start-up.
Fig. 12.
B. Warm-Up Mode
When the lamp ignition is detected, the circuit provides a
fixed switching frequency slightly above the unloaded resonant
frequency during a specified period. The warm-up time is 240 s
for 150-W high pressure sodium (HPS) lamps and 30 s for
150-W MH lamps. The lamp ignition generates a signal that
keeps fisw near and above fo during the lamp warm-up. If the
ignition fails or a significant decrease in the input current is
detected, the ignition sequence is repeated.
C. LFSW: Positive and Negative Buck Modes
After the lamp warm-up, fisw is shifted to a low frequency,
and the lamp voltage becomes a square wave. Current and
power mode controls are then activated to regulate the average
inductor current and the average lamp power [25]–[27]. The
control circuit establishes an alternate operation of the FB
inverter as a positive and negative buck converter with a frequency of 200 Hz. A dead time is included to assure safe operation. Positive and negative buck modes are achieved, generating
the corresponding drive signal at fsw = 200 kHz, as shown in
Fig. 4. In positive and negative buck modes, the inductor current
control and the power mode control are activated.
An equivalent for the digital inductor current control loop
gain Td in the continuous time domain is
Td (s) = Gc (esT1 )H(s)Gid (s)e−std
(17)
where Gc (z) is the discrete-time compensator with z = esT1 ,
T1 = 2/fsw is the sampling period, and e−std represents the
total delay of the controller from the input current sampling
instant to the time of corresponding duty cycle actuation. The
sample constant is H(s) = 0.25. The sample frequency is set to
half of the buck converter switching frequency, with sampling
instants just prior to the end of the duty cycle. The gains of the
input current A/D converter and DPWM modulator are assumed
to be unity.
Based on Gid (s) given in (14), and as shown in Fig. 9,
the inductor current loop requires a simple proportional
compensator to stabilize the electrical discharge, achieving the
required bandwidth. By assuming a total controller delay of
(a) Controlled system and (b) equivalent reduced controlled system.
one sample td = T and a desired loop gain crossover frequency
of fc = 5 kHz, a suitable discrete-time compensator is given by
Gc (s) = kc = 0.5. Discretization requires the verification that
no limit cycling condition [28]–[30] is produced or at least that
one bit oscillation around the operation point causes minimum
disturbances to the system, in this case lamp flickering. The
resolution of the sampled current i∗ is that one bit increment
means ∆I = 10 mA, whereas the 10-b DPWM resolution
is that one bit increment (∆D) produces ∆I = 5 mA, i.e.,
∆Plamp = 433 mW over Plamp = 150 W. Therefore, the duty
cycle resolution is higher than the analog-to-digital conversion.
Integral action to achieve zero error is performed by the
outer loop.
On the other hand, the power loop requires a simple integrator to achieve zero error lamp power control. T2 = 250 T1 is
the sampling time selected for this loop
Gp (s) = 0.2
s + 105
.
s
(18)
The controlled system is shown in Fig. 12(a) that is reduced to the equivalent system in Fig. 12(b), where Gip (s)
in Fig. 13(b) is the transfer function of the inductor current loop
Gip (s) =
i
iref
=
Td (s)/H(s)
1 + Td (s)
(19)
and Td is given in (17).
A stability analysis of the current loop using the SISO design
tool of MATLAB is shown in Fig. 13(a), whereas Fig. 13(b)
shows the stability analysis of the power loop.
Zero-order hold (ZOH) and bilinear transformation (BLT)
were used to obtain the equivalent of Gp (s) in the z domain,
for the z-transform of the Gp (s) to obtain GpZOH (z) and
GpBLT (z), respectively,
z
z−1
z − 0.332
GpBLT (z) = 0.3
.
z−1
GpZOH (z) = 0.2
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(20)
(21)
JAVIER DÍAZ et al.: DIGITAL CONTROL OF A LOW-FREQUENCY SQUARE-WAVE ELECTRONIC BALLAST
Fig. 13.
3187
Bode diagram and root locus of (a) the inductor current loop with the controller Gc (s) and (b) power + current loop with controllers Gc (s) and Gp (s).
In this case, GpZOH (z) is used as it is easier to implement by
software than GpBLT (z) with minimal error.
The corresponding digital current control algorithm is
given by
d[n] = 0.5 (i∗ [n] − iref ) .
(22)
Moreover, the digital power control algorithm is given by
iref [n] = iref [n − 1] + 0.2ep [n]
(23)
ep [n] = i∗g [n] − igref
(24)
where ep [n] is
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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 9, SEPTEMBER 2008
Fig. 14. LFSW electronic ballast controlled by the microcontroller dsPIC30F2010.
where i∗ is the inductor current sample, iref is the inductor
current reference for current loop, i∗g is the input current sample,
and igref is the input current reference for power loop.
igref =
Pgref
Plamp
=
Vg
ηlfsw · Vg
(25)
where Pgref and Plamp are the input and lamp target power,
respectively, and ηlfsw is the efficiency of the converter that is
considered constant in control law.
V. E XPERIMENTAL R ESULTS
The power stage is implemented according to the schematic
shown in Fig. 14. Switches S1 and S3 are implemented using a new generation of insulated-gate bipolar transistors,
HGTP12N60A4D, which have a good switching frequency
performance with anti-parallel hyperfast diodes D1 and D3. In
order to achieve high efficiency at a switching frequency of
200 kHz, HEXFET Power MOSFETs, IRFP340, are adequate
to implement S2 and S4 with antiparallel diodes D2 and D4.
Gate drive signals are generated with two IR2110 drivers, and
a dead time of 1 µs is selected for the inverter to prevent cross
conduction between the upper and lower transistors.
The inductor (1.3 mH) of the LC filter uses an E42 core
size, material N27, and the capacitor is a 1600-V metallized
polypropylene film, 47 nF [31]. The number of turns of
the inductor L is calculated to prevent core saturation during
the ignition sequence, where the flux density in the core is the
maximum. On the other hand, the copper section is calculated
according to the inductor current in steady state.
Two types of lamps have been used to verify the ballast performance: 150-W MH lamp (SYLVANIA) HSI-TD Metalarc
and 150-W HPS lamp (PHILIPS), whose nominal electrical
parameters are Vlamp = 100 Vrms and Ilamp = 1.5 Arms .
The digital control circuit is implemented in a dsPIC30F2010
whose clock frequency is 120 MHz. The dsPIC has peripheral
Fig. 15. Digitally controlled ignition sequence and detail of the breakdown
instant. Top: lamp voltage Vlamp . Bottom: lamp current Ilamp .
capabilities, including six PWM outputs and ADC with 10-b
resolution, 154-ns sample and hold time, and 2-µs conversion
time. The value of the sense resistor is Rg = 0.5 Ω.
Experimental results are given for a 150-W MH lamp and
include the resonant lamp ignition sequence where a close to
1-kV ignition voltage is shown in Fig. 15 and a square wave
operation is shown in Fig. 16. As comparative data, the typical
ignition voltage is 4 kV for 150-W MH lamp and 2.7 kV for
150-W HPS lamps using peak ignition voltage. Overcurrent
causes power spikes that make the instantaneous plamp differ
from the ideal constant value. The ac component of the lamp
power is under the specified 5% [16], [17], and no acoustic
resonances have been observed. Fig. 17 shows the lamp waveforms under dimming control when Plamp is reduced to 90 W.
Transitions of the current through the lamp are shown in Fig. 18.
Despite the low-frequency operation, in contrast to the case
when the lamp is supplied through other low-frequency ballasts,
fast transitions do not produce lamp reignition as is shown in the
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JAVIER DÍAZ et al.: DIGITAL CONTROL OF A LOW-FREQUENCY SQUARE-WAVE ELECTRONIC BALLAST
Fig. 16. Experimental waveforms in steady-state operation. Top: lamp
voltage (Ch1) Vlamp . Middle: lamp current (Ch2) Ilamp . Bottom: lamp
power (M1) Plamp .
Fig. 17. Experimental waveforms in steady state under dimming control
operation. Top: lamp voltage (Ch1) Vlamp . Middle: lamp current (Ch2) Ilamp .
Bottom: lamp power (M1) Plamp .
Fig. 18. Transitions from (left) positive to negative and (right) negative to
positive current through the lamp which depend on the quality factor of the
filter. Top: lamp voltage (Ch1) Vlamp . Middle: lamp current (Ch2) Ilamp .
Bottom: lamp power (M1) Plamp .
measured ilamp versus νlamp characteristic given in Fig. 19. The
transition from resonant to LFSW mode is stable, as shown in
Fig. 20. For the switching frequency utilized during the warmup time in the resonant converter mode, no acoustic resonances
have been observed in the tested MH and HPS lamps. The
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Fig. 19. Fast transition that does not produce lamp reignition. Horizontal axis:
lamp voltage (Ch1) Vlamp . Vertical axis: lamp current (Ch2) Ilamp .
Fig. 20. Transition from resonant to LFSW mode. Top: lamp voltage (Ch1:
100 V/div) and lamp current (Ch2: 2 A/div) Ilamp , time/div: 40 ms. Bottom:
zoom of lamp voltage (Ch1: 100 V/div) and lamp current (Ch2: 2 A/div) Ilamp ,
time/div: 400 µs.
Fig. 21. Experimental waveforms in steady-state operation. Top: sensed ig .
Middle: tsh is a sample and hold time, and tc is a converter time. Bottom:
transistor-driven signal.
situation when the microcontroller takes the sample of input
current is shown in Fig. 21. The sample and hold time required
by the ADC to obtain correct data is only 154 ns. The possibility
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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 9, SEPTEMBER 2008
Fig. 22. Experimental waveforms in steady-state operation under power loop
of an old lamp. Top: lamp voltage (Ch1) Vlamp . Middle: lamp current (Ch2)
Ilamp . Bottom: lamp power (M1) Plamp .
Fig. 23. (a) Arc of the HPS lamp, in this case a straight line. (b) Arc of the
MH lamp.
of controlling the time where the sampling is produced enables
noisy zones to be avoided, i.e., during the transistor switching.
In order to verify the power loop control under load changes, as
shown in Fig. 22, the lamp has been replaced by an old lamp.
The arcs in HPS and MH lamps are shown in Fig. 23(a) and (b)
respectively, where no arc distortion appears. The efficiency of
the LFSW section measurement is around 92% and corresponds
to a prototype that verifies the proposed new control technique
and is not optimized for maximum efficiency.
VI. C ONCLUSION
A new digitally controlled LFSW ballast using a single FB
circuit for both low-frequency drive and resonant lamp ignition
has been presented. One of the benefits of this electronic ballast
is the reduction of components due to use of digital control
and fewer power stages than similar ballasts. In this way, the
proposed converter is a universal solution for different lamps
of the same wattage (HPS and MH lamps). The digital control
defines three operation modes: as a resonant inverter to achieve
the lamp ignition and initiate the warm-up, and as a positive and
negative buck converter during the LFSW operation mode. The
buck converter operation is both inductor current controlled to
stabilize the lamp operation in the short term (ballast action)
and input power controlled to provide constant lamp power
during the lamp lifetime (long term). The design of the LC filter
assures a quality factor that limits the resonant current during
the ignition and generates the fastest transitions of the LFSW
around the critically damped behavior. Experimental results
confirm the system stability with different lamp types and aging
and that fast low-frequency transitions of the lamp current do
not produce lamp reignition effects. No light arc distortion has
been observed, and the lamp power is accurately established in
nominal conditions and under dimming control operation.
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F. Javier Díaz was born in Torrelavega, Spain,
in 1973. He received the M.S. degree in electrical and control engineering from the University of
Cantabria, Santander, Spain, in 1998, where he is
currently working toward the Ph.D. degree in electronics engineering.
Since 1998, he has been an Assistant Professor
with the Department of Electronics Technology,
Systems and Automation Engineering, University
of Cantabria. His research interests include design, modeling, and control of switch-mode power
converters for discharge lamps and topologies for power factor correction
applications.
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Francisco J. Azcondo (S’90–M’92–SM’00) was
born in Santander, Spain, in 1965. He received the
M.S. degree in electrical engineering from the Universidad Politécnica de Madrid, Madrid, Spain, in
1989, and the Ph.D. degree from the University of
Cantabria, Santander, in 1993.
From 1990 to 1995, he worked on the design of
highly stable quartz crystal oscillators. Since 1995,
he has been an Associate Professor with the Department of Electronics Technology, Systems and Automation Engineering, University of Cantabria. His
research interests include switch-mode power converters and their control for
discharge lamps, electrical discharge machining, and power factor correction
applications.
Rosario Casanueva (M’04) received the M.S. and
Ph.D. degrees in physics from the University of
Cantabria, Santander, Spain, in 1991 and 2004,
respectively.
From 1991 to 1993, she worked on the design of
highly stable quartz crystal oscillators. Since 1993,
she has been an Assistant Professor with the Department of Electronics Technology, Systems and Automation Engineering, University of Cantabria. Her
research interests include digital and analog microelectronic circuit design, switch-mode power converters, resonant converters, and their control for electrical discharge machining
and discharge lamp applications.
Christian Brañas (M’03) received the M.S. degree
in electronics engineering from the Instituto Superior
Politécnico José A. Echeverría (ISPJAE), Havana,
Cuba, in 1992, and the Ph.D. degree in electronics engineering from the University of Cantabria,
Santander, Spain, in 2001.
From 1992 to 1995, he was an Instructor Professor
with the ISPJAE. Since 1995, he has been with the
Department of Electronics Technology, Systems and
Automation Engineering, University of Cantabria,
where he is currently an Assistant Professor. His research interests include the design, modeling, and control of resonant inverters
and switch-mode power supplies.
Regan Zane (M’99–SM’07) received the B.S., M.S.,
and Ph.D. degrees in electrical engineering from the
University of Colorado, Boulder, in 1996, 1998, and
1999, respectively.
From 1999 to 2001, he was a Senior Research
Engineer with the GE Global Research Center,
Niskayuna, NY. At GE, he developed custom integrated circuit controllers for power management in
electronic ballasts and lighting systems. He joined
the University of Colorado as a Faculty Member
in 2001 and is currently an Associate Professor of
electrical engineering with the Department of Electrical and Computer Engineering. At the university, he has ongoing research programs in energy-efficient
lighting systems, adaptive and robust power management systems, and lowpower energy harvesting for wireless sensors.
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