Improving the Regulation of
Multi-Output Flyback Converters
C. Mullett
F. Cathell
ON Semiconductor
732 Montclair Drive
Santa Paula, CA 93060
chuck.mullett@onsemi.com
ON Semiconductor
7402 W. Detroit St., Suite 140
Chandler, AZ 85226
frank.cathell@onsemi.com
Abstract---Flyback switched-mode converters are among the most
popular in the low-power range, due to their simplicity and low
cost. They are especially popular in multi-output applications,
due to the low parts count---one diode and capacitor per output.
They would be even more popular if the output regulation were
better. This paper reviews the common techniques to tighten the
output regulation, and summarizes the measurements of
performance of each, to aid the designer in making accurate
choices from the available approaches. [1] Finally, a four-output
design example is built and tested.
drops across the diodes. Taking into account the diodes, we
have
Vout1 = ( n ) (Vout 2 + Vd 2 ) − Vd 1
m
(1)
I. INTRODUCTION
Flyback converters store energy in the transformer during
the conduction time of the primary switch, then deliver that
stored energy to the secondary circuits after the primary
switch is turned off. Because the energy storage occurs in the
transformer, there is no need for energy-storage inductors in
the secondary circuits as in the usual buck-derived topologies
such as forward converters, half- and full-bridge converters,
etc. This results in very simple secondary circuits---one diode
and filter capacitor for each output---making the flyback a
very popular choice for multi-output power supplies. This
paper provides insight into various ways to improve the
regulation of these multiple outputs and includes the results of
tests on the basic schemes.
II. THE BASIC CIRCUIT
Figure 1 shows a typical two-output off-line flyback, in
which output 2 is chosen is the master, leaving output 1 quasiregulated. Output 2 is well-regulated, but output 1 is affected
by the load on output 2, and also has poor regulation at light
load. The mechanism for this behavior is the coupling
between the secondary windings of the transformer. The
voltage across the secondary of output 2 is determined by the
desired output voltage, Vout2, since the PWM controller on the
primary side will adjust the pulse width to deliver the required
output, and the voltage on the m-turn winding that feeds
output 2 will be clamped at Vout + Vd, where Vd is the forward
voltage drop of the rectifier diode feeding output 2. the turns
ratio, n/m, of the other secondary winding will determine the
output voltage of the other output, Vout1. Taking into account
the voltage drop of the diodes, the output voltage on output 1
will be approximately (n/m)•Vout2, neglecting the voltage
978-1-422-2812-0/09/$25.00 ©2009 IEEE
Figure 1. Dual-output off-line flyback converter.
The equations for the output voltages as a function of the
input voltage, Vin, as shown on the figure, assume continuous
conduction of the transformer, where D is the duty ratio of the
primary switch and D’ is the complement, 1-D.
Two facts can be inferred from Equation 1. First, Vout1
depends on both Vd1 and Vd2, and second, the relative
contributions of the two diode drops are affected by the turns
ratio, n/m. However, even if n = m, the diodes would not
compensate each other due to mismatch of the diodes and the
fact that the two loads, R1 and R2 are usually unequal. Also,
the two secondary windings are not perfectly coupled, and this
leads to a load-dependent inaccuracy of the ratio of voltages
between the two windings. The usual remedies for these
shortcomings are to tightly couple the two windings by
interleaving or bifilar winding construction, and sometimes to
alter the turns ration on the windings to compensate for the
diodes.
III. STACKED WINDINGS
Perhaps the most popular step to improve the second
output’s regulation is to stack the windings as shown in Figure
2. In this case, only the upper winding is unique to the upper
output. Because the lower winding is feeding the regulated
output, and the current through the lower diode is not
1923
conducting any of the upper output’s current, the voltage
across the lower winding will remain well-regulated, leaving
only the upper winding as quasi-regulated. A consequence of
this, however, is that when the load is changed on the lower
output, the voltage on the upper output will also be affected.
[2, 3, 4]
and the desired compromise between one output’s “tightness”
compared to that of the other.
V. COMBINED FEEDBACK
Another technique to improve the regulation of the
subordinate output, but this time with some degradation to the
main output, is to combine the feedback, as shown in Figure 4.
Vout 1
n
Vin
Load (R1) Vout 1 =
m+n)D
Vin
D'
1
Vout 2
m
PWM
Controller
Load (R2) Vout 2 =
mD
Vin
D'
Optocoupler
R6
R3
R4
TL431
2.5 V ref. amplifier
Figure 2. Stacked windings.
R5
Figure 4. Combined feedback.
In this configuration, Vout1 is larger than Vout2, since the two
windings are stacked in “series aiding” fashion. Note,
however, that it is also possible to stack them in “series
subtracting” fashion by reversing the polarity of the second (n)
winding.
IV. STACKED OUTPUTS
It is, of course, possible to combine any number of outputs
into the one feedback path, and the feedback can be weighted
to favor one or more outputs over the others by choosing the
feedback resistors appropriately. [7] Figure 5 illustrates the
process of combining two outputs into the single feedback
system, with the ability to set the importance of each output
via weighting the feedback network.
Another method is to stack the outputs, rather than stack the
windings. [5,6] Although very similar in appearance, the
performance is decidedly different. As shown in Figure 3, the
bottom of the upper winding is connected to the terminal of
the lower output, which is precisely regulated by the feedback
loop. Now upper output is virtually unaffected by changes in
load on the lower output, and the regulation of the upper
output is improved due to the smaller number of turns that are
unregulated. This configuration is shown in Figure 3.
Figure 5. Combined feedback with weighting of the two outputs.
Designing the feedback network is simple, as illustrated in
Figure 6 and the equations that follow.
Vout 2
Vout 1
Optocoupler
Figure 3. Stacked outputs.
In all three of the above cases, individual load regulation and
cross-regulation are different, and not one is a clear winner.
The choice will depend on the ranges of loads on the outputs
978-1-422-2812-0/09/$25.00 ©2009 IEEE
1924
TL431
2.5 V ref.
amplifier
i2 = W2 • i0
R2
i0
R0
R1
i1 = W1 • i0
Vref
Figure 6. Details of the weighted feedback network.
Remembering that W1 and W2 must sum to one (W1 + W2 =
1) and choosing an arbitrary value for i0, the two resistors can
be determined as follows:
Vout1 − Vref = i1 R1
R1 =
Vout1 − Vref
R2 =
(2)
=
i1
capacitor between the unloaded winding and another that is
loaded can reduce or eliminate the ringing on the unloaded
output. For this to work, it is absolutely necessary that the two
terminals of the capacitor are connected between two nodes
that exhibit the same waveform. This requires that the two
windings have the same number of turns, as shown in Figure
8. In this example, the number of turns on the upper winding
(n) must equal the number of turns on the lower winding (m),
hence, m = n.
Vout 2 − Vref
=
i2
Vout1 − Vref
(3)
W1i0
Vout 2 − Vref
(4)
W2i0
With cap: Clean pulse; improved
regulation at low-current load
Vout 1
Vout 1 =
n
As an example, consider a dual-output circuit where Vout1 = 5
V and Vout2 = 12 V, and the current through R0 is chosen to be
1 mA. In this example, outputs 1 and 2 are weighted at 0.7 and
0.3, respectively. Then:
R0 =
R1 =
R2 =
Vref
i0
=
2.5
= 2.5 kΩ
1 mA
Vout1 − Vref
W1i0
Vout 2 − Vref
W2i0
=
12 − 2.5
= 31.7 kΩ
0.3 ⋅1 mA
(7)
R0
Vout 2 = Vout 1 =
nD
Vin
D'
R4
TL431
2.5 V ref. amplifier
Vout 1
i0
TL431
2.5 V ref.
amplifier
Vout 2
m=n
PWM
Controller
R3
R1
Vout n
i1 = W1 • i0
R5
Figure 8. A capacitor couples two windings to reduce ringing on the unloaded
winding.
Optocoupler
R2
1
Load (R2)
There is no theoretical limit to the number of outputs that
can be included in the feedback scheme. The general method
for handling a total of n outputs is shown in Figure 7 and in
the equations that follow:
i2 = W2 • i0
Vin
Optocoupler
(6)
Vout 2
D'
Low-current load (R1 = large)
Vin
(5)
5 − 2.5
= 3.57 kΩ
0.7 ⋅1 mA
=
nD
It is possible to apply the capacitor coupling technique
where the two windings have unequal turns by tapping the
winding that contains the greater number of turns, such that
the waveform at the tap is identical to the waveform of the
other winding, as shown in Figure 9. In this case, the 12 V
winding would be tapped at 5/7 of its total turns, since it is
actually a 7 V winding connected to the 5 V dc output.
in = Wn • i0
Rn
Vref
Figure 7. Feedback circuit for a number (n) of outputs.
remembering again that W1 + W2 + … +Wn = 1, each resistor
is determined as shown in equation 8 below.
Rn =
Vout n − Vref
Wn ⋅ i0
(8)
Figure 9. Using the capacitor coupling to a tapped winding.
VI. CLAMPING THE RINGING
When one of the outputs becomes unloaded the waveform
on the secondary exhibits a “ringing” on top of the pulse,
causing the output to rise as a result of the diode “peak
detecting” the signal, as shown in Figure 8. Installing a small
978-1-422-2812-0/09/$25.00 ©2009 IEEE
VII. ANOTHER MULTI-OUTPUT TECHNIQUE
The flyback multiple-output technique is not limited to
flyback converters. In fact, the technique can be applied to
1925
buck-derived converters by adding one or more windings to
the output inductor as shown in Figure 10. [8]
Figure 10. Additional output formed by a flyback winding on a buck
regulator’s output inductor.
The circuit takes some of the energy stored in the output
inductor and delivers it to a second output during the
“flyback” portion of the switching period of the buck
regulator.
During this interval the voltage across the
inductor’s main winding is simply the output voltage, Vout1,
plus a diode drop, with the voltage positive at the dotted end
of the winding. The same voltage, scaled by the turns ratio,
appears on the added winding, and the added output, Vout2,
will be the winding voltage minus the forward voltage of the
added rectifier diode. Operation of this circuit is best when
the inductor is in continuous conduction, because the two
windings are well-coupled when the main diode is conducting.
It is important to realize that continuous conduction requires
that the added load current not drive the current to zero,
meaning that the ampere-turns of the added winding be less
that the ampere-turns of the main winding.
Figure 11. Circuit for measuring the regulation of an added output.
E f f e c t s of St a c k i ng a nd A ddi ng t he C a pa c i t or
( A l l a r e nor ma l i z e d t o 5 . 0 0 V a t 1 A l oa d)
7
6. 5
6
Conf i g. 1
Conf i g. 2
Conf i g. 2 wi t h Cap.
5. 5
VIII. EXPERIMENTAL RESULTS
5
The circuit shown in Figure 11 is used to evaluate the effect
of adding a second winding (output C) by either stacking the
winding (configuration 1) or stacking the output
(configuration 2), and also the effect of adding the coupling
capacitor to configuration 2. The results are shown in the
graph of Figure 12. Note that the zero-load value of output C
is improved in configuration 2, even without the capacitor, due
to placing the bottom of the winding of output C on the output
terminal of output b, which has virtually no overshoot and
ringing. Finally, adding the coupling capacitor further reduces
the no-load output. Note that the capacitor could not be added
to configuration 1, because it would result in a high-frequency
short circuit across the winding of output C. It also requires
that the two windings have equal turns (m).
These techniques are not new. Stacking windings and
outputs has been done throughout the history of transformerbased power supplies. It has been the purpose of this paper to
remind designers of the effectiveness, especially in multioutput flyback converters as they become even more popular
in modern electronic products. An example in a modern
consumer product is given in Reference 6, “50 W Four-Output
Internal Power Supply for Set Top Box.” The complete
documentation package is available at no charge on the ON
Semiconductor Web site, www.onsemi.com.
978-1-422-2812-0/09/$25.00 ©2009 IEEE
4. 5
0
0. 2
0. 4
0. 6
0. 8
1
1. 2
Load on Out put C ( A )
Figure 12. Regulation of the added output (C) with output B held at 1 A.
REFERENCES
[1]
[2]
[3]
[4]
[5]
[6]
[7]
[8]
1926
C. Basso, Switch-Mode Power Supplies. New York: McGraw-Hill,
2008, p. 714.
“Wide Input Range DC to DC Converter, Design Note DN06007/D, ON
Semiconductor, www.onsemi.com .
“8 W, 3-Output Off-Line Switcher,” Design Note DN06003/D, ON
Semiconductor, www.onsemi.com .
“10 W, Dual Output Power Supply,” Design Note DN06020/D, ON
Semiconductor, www.onsemi.com .
F. Cathell, “High Efficiency Eight Output, 60 W Set Top Box Power
Supply Design,” Application Note AND8252/D, ON Semiconductor,
www.onsemi.com.
“50 W Four-Output Internal Power Supply for Set Top Box,” Reference
Design Documentation Package, TND334/D, ON Semiconductor,
www.onsemi.com.
Bassp. p.718.
“1 W, Dual Output, Off-Line Converter,” Design Note DN06002/D, ON
Semiconductor, www.onsemi.com.