Over the past several years, HBT-based (Hetero-Junction Bi-
Polar Technology) RF Integrated Circuits (RFICs) have gained wide acceptance by all major wireless and wire-line equipment suppliers as the preferred technology where performance, highlinearity, efficiency and price competitive-ness are a premium(1). These RFIC products include power amplifiers for handsets, driver amplifiers for CELLULAR/PCS base stations and CATV/FIBER line driver amplifiers. In these applications over the past two years, HBT-based products have overtaken many incumbent products based on GaAs MESFET and Silicon Bipolar technologies by providing price competitive, high performance solutions.
In this paper we examine trade-offs between various device technologies for high linearity base-station applications. Examples of RFICs fabricated using Aluminum Gallium Arsenide
(AlGaAs) GaAs, Indium Gallium Phosphide (InGaP) GaAs and
Silicon Germanium (SiGe) HBTs and their typical application in base-station Rx/Tx chains are discussed at a top level. As there are many possibilities we will address a simple example of the block diagram.
In order to achieve MTTF of greater than 20 years, we design our RFICs to operate at junction temperatures of less than
150 (C at an ambient package lead temperature of 85 (C. Extensive characterizations using IR imaging techniques confirms that these devices operate within the rated junction temperature range.
GaAs-based devices offer higher gain, frequency operation and IP3 than Silicon devices as a result of the superior carrier mobility and transport properties. However, with the addition of Silicon Germanium (SiGe) into the base of the conventional Silicon HBT, the advantages of the better transport properties of the base as well as the resultant base-emitter heterojunction, makes this device technology more GaAs-like in performance ( 2, 3 ). In the area of power density, the bipolar device (both GaAs & Silicon) has an inherent advantage due to the vertical nature of current flow in the device. FET-based devices (includes HEMT) typically have a 2 to 3 times larger die size for the same output power 4 . However, this higher power density also results in thermal management issues for GaAs
HBTs due to the low thermal conductivity of GaAs substrates
(0.4 W/cmK vs 1.1 W/cmK for Silicon).
In high linearity applications, a good figure of merit is the difference in output IP3 (OIP3) and the output 1.0 dB compression point (P1dB). This figure of merit is a measure of the efficiency of the amplifier. Typical values for MESFET and
HEMT are between 9.0dB -10.0 dB. GaAs HBTs typically have
OIP3-P1dB greater than 15.0 dB. This excellent figure of merit has been attributed to the cancellation of base-emitter heterojunction.
Table 1: Comparison of competing technologies for high linearity RFIC
Non-linearities, which results in a suppression of intermodulation products of multi-tone signals. Although earlier published results of SiGe HBT indicate a figure of merit of about 9.0 dB, we have recently measured numbers comparable to GaAs HBT, realizing greater than 14.0 dB.
For devices operating under large signal conditions, high device breakdown is required. GaAs devices typically have higher breakdown voltage than Silicon devices for comparable doping concen-trations. This is a result of the wider bandgap for GaAs compared to Silicon. Breakdown voltage is less of a concern for handset power amplifiers where the recent trends have driven devices towards 3.0 V operation.
However, in the case of base-stations, the supply rail is typically greater than 10V and a higher operating bias point is preferred because of IR losses associated with high current, low voltage operation. A summary of competing RFIC device technologies (Table 1, above page 2) is compared with each of the major performance requirements necessary for trade-off and selection.
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In the typical transmit and receive chain (see Fig. 1 below) the designer is faced with systems level decisions regarding the gain, efficiency, Noise figure and power output. In situations where performance is the main consideration the designer is usually looking for high 3rd order intercept (IP3) or linearity, efficiency and P1dB on the transmit side. On the receive side low noise figure dominates the equation, along with efficiency and gain. For low cost, high volume micro/pico systems the choices were few for the manufacturing engineer concerned about price over performance.
Until recently amplifier choices were limited to discretes and in some cases prematched circuits. Now the choices have broadened to gain blocks and multi-stage high-gain amplifiers due to the availability of SiGe and GaAs HBT technology. Products made possible by innovative pioneers are now available in high volumes as the output power amplifier in many handsets and power driver in base-stations. These choices enable the designer to make decisions which can provide the highest possible power, and IP3 for performance situations and still allow for competitive pricing decisions where performance is not the main driver.
A generic example of a transmit/receive block diagram is provided (in Figure 1) to give the designer a pictorial concept of how and where Stanford Microdevices’ products can be used as pre-amplifier, amplifier, power amplifier stages in a basestation and repeater system. In the block diagram shown the designer for a typical 2.0 GHz high performance design can now choose between the GaAs HBT for a power stage; +19dBm
P1dB, 38dBm IP3, 15 dB gain or the SiGe HBT; +20dBm P1dB,
34dBm IP3, 16dB gain. Both choices will consume 0.4W DC power; the GaAs will have the higher IP3 and frequency range.
The selection is somewhat easier for the pre-driver stages where shown in Figure 2. Input matching and feedback stabilization was done on-chip, output matching was left off-chip due to limitations of on-chip lumped element losses. The final stage size was chosen to provide good P1dB while simultaneously matched for highest linearity. Inter-stage matching on the two stage designs was chosen to provide good gain, P1dB, and IP3 response across the bands of interest. Input matching was synthesized for good input return loss. R-C feedback was added around each stage to help guarantee unconditional stability.
The amplifiers uses an on-chip self-bias circuit that provides excellent performance over temperature and as well as process. Similar to a standard current mirror, the power transistors track a small mirroring transistor in the bias circuit. The bias circuit was designed to hold device current flat over temperature and process variation. This helps keep gain, P
1dB
, and
IP
3
flat over these conditions as well.
Table 2: Typical performance of GaAs HBT PA’s at: 850Mhz, 1.95 and 2.15Ghz
As shown in Figure 1 , there may be both a mixer and a modulator/demodulator in the transmit and receive chains of a typical radio architecture. We are presently developing high linearity parts in SiGe HBT for these applications.
Figure 2: Two Stage Power Amplifier Topology
Figure 1: Typical Tx/Rx Block Diagram the SiGe has a distinct advantage due to its price-performance ratio. As the power requirements increase the InGaP/GaAs HBT will have the distinct advantage over the SiGe due its inherent power advantages at frequencies over 2.0 - 6.0 GHz.
A family of power amplifiers for 850, 1950, and 2150 MHz has been developed for linear amplifiers used in the driver stages of wireless base-stations. These amplifiers are packaged in thermally enhanced exposed pad 8-SOIC packages and have excellent IP3 performance as shown in Table 2. The designs use standard matching topologies and stabilization techniques as
Similar to amplifiers, the choices for high linearity mixers have typically been discrete devices, or modules. Passive FET mixers, often called resistive FET mixers have been the most popular method of achieving high linearity mixers, typically fabricated in MESFET or pHEMT processes. A simple block diagram of a single device FET mixer is shown in Figure 3. The gate is commonly biased near pinch-off and the LO is applied to the gate, so the channel acts as a gate-controlled resistor. The RF is applied to the drain through a high pass filter, and the IF is extracted from the drain through a low-pass filter. The passive
FET mixer is inherently linear, and may consist of a single transistor, but it requires a large LO signal, and as shown in Figure
3, matching and /or filtering on all ports. As well, passive mixers typically have 7 to 10 dB of insertion loss.
Active mixers based on the Gilbert cell (6) analog multiplier have existed for many years. This type of active mixer is typically implemented in, but not limited to bipolar technologies in either Si or GaAs. A simple block diagram of a balanced
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downconverting Gilbert cell is shown in Figure 4. In this configuration, the RF signal drives the lower differential pair. The
LO drives the quad pairs, which switches the current between the arms of the mixer, resulting in the multiplying of the RF and the LO. The balanced IF is taken off the collectors and is normally fed into an output buffer stage. The cross connection of the collectors of the quad pairs results in a virtual ground for the LO and the RF as well as even order spurious modes and intermod products.
As a result this type of mixer does not need filtering between the RF and IF ports. In addition this mixer may easily be matched to 50 ohms over a frequency range on-chip, which results in an overall size reduction on the circuit board. Because the mixer is active, the noise figure may be higher than the passive mixer, but it also has gain, which may balance out the extra noise figure in the system design.
Figure 5: Block diagram of SiGe receive mixer with LO and IF buffer amplifiers
The mixer conversion gain varies by less than 0.5 dB over the entire DCS band. The intermod products from a simulated
2-tone test (f
1
= 1750.0 MHz and f
2
= 1750.8MHz) vary linearly with input power. At the highest plotted input power (-4 dBm) the intermod products are approximately 52 dB below the output signal. Similar performance is achieved across the
PCS band.
Figure 3: Simplified block diagram of a single transistor passive FET mixer
In recent years, active Si mixer have become popular in wireless terminals as the power levels are suitable for this type of device, and integrated Si mixers have the advantage of low price and small size. However, until recently active Si mixers have not had sufficient linearity for most basestation applications.
New SiGe processes with ft ~ 50 GHz, are making SiGe circuits competitive with GaAs for certain applications at higher and higher frequencies as the technology improves. The combination of the faster process and some proprietary design techniques has enabled Stanford Microdevices to design very high linearity mixers operating in the 2 GHz band of frequencies for basestation applications.
Figure 6: Simulation of conversion gain of the receive mixer in the DCS1800 band.
Based on this mixer performance, a comparison of commercially available passive mixers and the SiGe HBT mixer is presented in Table 4. As previously mentioned, the active mixer has gain, whereas the passive mixer has loss. The input IP3 of the passive mixer looks much higher than the active mixer, but when the gain is taken into account, the output IP3 is in fact higher than the passive mixer. The LO drive for the active mixer is much lower than the passive mixer, as the LO buffer is on chip. As well, the integrated IF buffer makes the active mixer a much smaller solution, occupying less PCB real estate.
Figure 4: Simplified schematic of a balanced Gilbert cell down-converting mixer
A block diagram of a SiGe receive mixer is shown in Figure
5. The IC includes the mixer, an LO buffer and an IF buffer. It has broadband 50 ohm balanced ports, and has the ability to drive an IF SAW filter without additional buffer circuits. This circuit has been designed to operate in the 1700 to 2000 MHz band to address DCS, PCS and 3G applications. The key target specifications for this device are presented in Table 3. Some simulation results are shown in Figures 6 and 7 for conversion gain and intermod performance in the DCS band.
Table 3: Typical target specifications for SiGe receive mixer.
To summarize the mixer discussion, active mixers, which offer comparable performance to passive FET mixers are emerging in SiGe HBT. SiGe HBT offers size, cost and performance advantages over GaAs MESFET and pHEMT for
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Figure 7: Simulation- Pout and IMD3 vs Pin for the receive mixer. (Middle of DCS band) basestation applications leading to an overall superior system implementation.
Measured data for this mixer will be included in our presentation, along with examples of other SiGe mixers presently in development.
neering teams at Stanford Microdevices Inc. (SMDI) have successfully developed techniques using infrared thermal scans to accurately measure the hot spots of our RFICs in “real time” operation (Figure 8). Using a quiescent current of between 10 to 25 KA/cm 2 and adequate device spacing, we have achieved a junction temperature of less than 150
°
C, for our AlGaAs/ GaAs and InGaP/GaAs HBT RFICs at a package lead temperature of
85
°
C.
This is a major enhancement over previous and competitive product performance and designs for improved reliability. The result is that at these temperatures, the mean time to failure
(MTTF) of these devices is greater than 1 million hours (114 years). SMDI consider the 1 million hour MTTF point optimal for operational reliability.
Figure 8: IR-Scan measurement of InGaP HBT (Ta = 85oC, Tj (Peak
Junction Temp.) = 141.5
°
C)
Table 4: Active versus Passive Mixers
RFICs used in base-stations receive and transmit chains are typically used under CW, saturation as in GSM, operation as well as conditions where input drive is heavily backed-off, as CDMA/
WCDMA is, to meet distortion and linearity requirements. The system’s operational needs together with the high ambient operating temperatures and demanding environments places strict requirements on the proper thermal design of RFICs, taking into consideration both the die as well as the package. The high power density capability GaAs HBTs together with the low thermal conductivity of GaAs substrate places additional thermal consid-erations during the design and layout of the die. These are the critical factors taken under consideration by the engineering team to design and manufacture an RFIC for commercial use.
Although GaAs HBTs typically can operate up to 50 KA/ cm 2 with excellent performance, the reliability of the dies will be severely impacted as a result of the high junction temperature. The physical separation of the devices can also result in excessive adjacent device heating of multiple finger devices.
Careful trade offs between low current density operation, optimum performance and overall die size must be made in order to design a device that meets the overall requirements for basestations.
As a way to ensure the quality of product once the engi-
One of the major advantages of using our RFICs manufactured using SiGe HBTs is cooler operation. They typically have junction temperatures less than 125
°
C at a package lead temperature of 85
°
C. The cooler operating junction temperature is the result of the 3X improvements in thermal conductivity of the Silicon substrate as compared to GaAs and proprietary techniques developed by our design team. At this junction temperature (125
°
C), the devices have MTTF of greater than 1 million hours of operation. An example of a SiGe HBT technology life test curve is provided in Figure 9.
Figure 9: Typical Arhenius Plot For SiGe HBT
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HBT technology is proving itself to be well suited for high linearity, low cost RFIC product applications. Both GaAs and
Silicon based HBT technologies provide higher IP3 than their counterpart GaAs MESFET/HEMT and Silicon BJT technologies. For high linearity amplifiers AlGaAs/GaAs and InGaP/
GaAs HBTs offers “Better” performance, IP3, output power and frequency range, as well as higher voltage operation than
SiGe HBTs. For high linearity mixers SiGe HBT is becoming the technology of choice. SiGe HBT can provide comparable mixer performance to GaAs FETs for basesation applications in a more compact, lower cost implementation.
Another factor, which should not be overlooked, is the ubiquity of the HBT for integration as MMICs, as well as discrete components, and their subsequent use in multi-function micromodules. Micro-modules will need to make use of many technologies, GaAs and SiGe HBT, GaAs PHEMT, and FET, to capitalize on the performance and price points that will be required, demanded, by the growing basestation market.
(1) Allied World Report “Wireless Power Devices, Transistors,
ICs, and Power Modules: Strategies, Technologies and
Trends”, Dec 99.
(2) A. Schuppen, et. al., “A SiGe RF Technology for Mobile
Communications Systems”, pp.39-46, Microwave Engineering Europe, June 1998
(3) U. Erben, et. al., “ Applications of SiGe HBTs in 5.8 and
10GHz LNAs, pp. 1498-1500, Electronic Letters, Vol 34,
No. 15, 1998.
(4) Dimitris Parlidis, “HBT vs. PHEMT vs. MESFETs: What’s
Best and Why”, GaAs Mantech Digest, 1999.
(5) CIBC Oppenheimer, “GaAs Semiconductor Industry Update”, Ref. # 98-2635, March 20, 1998
(6) B. Gilbert, “ A Precise Four-Quadrant Multiplier with
Subnanosecond Response”, IEEE J. Solid-State Circuits,
Vol. SC-3, Dec. 1968, p 365
Kin L. Tan
Stanford Microdevices Inc.
522 Almanor Ave., Sunnyvale, CA 94086
Phone: 1 562 627 0745
Fax: 1 562 627 0704
Email: ktan@stanfordmicro.com
Bob Pinato
Stanford Microdevices Inc.
Phone: 1 562 627 0725
Fax: 1 562 0704
Email: bpinato@stanfordmicro.com
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