Journal of Electrical Engineering & Technology Vol. 5, No. 2, pp. 319~328, 2010 319
An Optimal Design Methodology of an Interleaved
Boost Converter for Fuel Cell Applications
Gyu-Yeong Choe*, Jong-Soo Kim*, Hyun-Soo Kang** and Byoung-Kuk Lee†
Abstract – In this paper, an optimal selection methodology for the number of phases will be proposed
for an interleaved boost converter (IBC). Also, the analysis of the input current ripple according to
CCM and DCM is carried out. The proposed design methodology will be theoretically analyzed, and
its validity verified by simulation as well as with experimental results. Moreover, a comparison of cost
and efficiency based on a 600W laboratory prototype using the Ballard NEXA 1.2kW PEMFC system
is demonstrated.
Keywords: Optimal Design Methodology, Interleaved Boost Converter, Fuel Cell Converter, Ripple
Analysis
1. Introduction
Recently, the interest in fuel cells has been increasing
due to the potential exhaustion of fossil fuel, concerns over
global warming and increases in power demands. An annual meeting for fuel cell systems has been initiated by the
U.S Department of Energy (DOE), and this has resulted in
much information relating to the increase of fuel cell projects [1]-[2]. These fuel cell systems mainly consist of a
fuel cell stack, an EBOP (Electrical BOP), and a MBOP
(Mechanical BOP).
In an EBOP, a PCS (Power Conditioning System) plays
an important role in order to deliver high quality and stable
power from the fuel cell to various loads. The PCS of a
fuel cell system is composed of a dc-dc converter and a dcac inverter. In fuel cell systems designed for small/medium
power, the dc-dc converter topologies are found to be either a full-bridge, a half-bridge or a push-pull type [3].
However, in high power fuel cell systems, such as those
that operate over 250kW MCFC, the dc-dc converter needs
to handle a much higher current, therefore an interleaved
boost converter (IBC) is considered a good solution with
respect to efficiency and current distribution.
As mentioned above, the IBC has been much studied
due to its merits, such as a high efficiency, a high current
distribution, a low ripple reduction and a small component
capacity. Previous research on the IBC can be summarized
as follows: In order to decrease the input current ripple and
improve the inductor utilization, a coupling inductor is
used [4]-[5]. Power factor correction has been studied employing an IBC [6]-[8]. An IBC was operated by DCM
(Discontinuous Conduction Mode) in order to reduce the
inductor size and to improve efficiency as a result of a de†
Corresponding Author: School of Information and Communication
Engineering, Sungkyunkwan University, Korea. (bkleeskku@skku.edu)
*
School of Information and Communication Engineering, Sungkyunkwan University, Korea.
** Korea Railroad Research Institute (hanmin@krri.re.kr,
cmlee@krri.re.kr, gdkim@krri.re.kr)
Received : January 13, 2010; Accepted : March 31, 2010
crease in diode reverse recovery loss [8]. Also, there has
been research into improving the control performance of
the converter by the analysis of a small signal model [9]
and by analyzing the input current ripple and output voltage ripple according to inductor coupling [10]-[12]. Current distribution control for solving the current imbalance
of each phase has also been studied [13]-[14].
For the optimal design of an IBC, based on these previous works, the decision on the number of phases needs to
be considered in the first design stages of the IBC. Therefore, in this paper, an optimal selection methodology for
the number of phases is proposed for the IBC with respect
to the input current and the output voltage ripples, taking
into account the duty ratio at the CCM and the DCM, the
switching frequency, and the inductance. Also, design
guidelines to select the passive components of the IBC,
such as the inductor and the capacitor are presented. The
proposed design methodology is theoretically analyzed,
and its validity is verified by simulation and experimental
results based on a 600W laboratory prototype using the
Ballard NEXA 1.2kW PEMFC system.
2. Ripple Analysis of Interleaved Boost Converters
2.1 Configuration and Operation Principle of the
IBC
Fig. 1 shows the circuit configuration of a 3-phase IBC
Fuel Cell
iin
Vin
i3
L3
D3
i2
L2
D2
i1
L1
D1
SW3
SW 2
SW1
iO
C
Vo LOAD
Fig. 1. Configuration of 3-phase interleaved boost converter.
320
An Optimal Design Methodology of an Interleaved Boost Converter for Fuel Cell Applications
where it can be seen that the inductors, switches and diodes
are connected in parallel and then to a single output capacitor. Each phase’s switching frequency is identical and each
switch has the same phase shift angle, namely, 360°/N.
According to the duty ratio, the switching sequences of
each phase can either be overlapped or not, resulting in
variations of the input current ripple; it is evident that the
input current ripple frequency will be N times the frequency of the inductor current.
The analysis of the current and voltage ripples according
to the duty ratio is derived under the following assumptions:
1) The resistances of the inductors and capacitors are negligible.
2) Stray inductances and capacitances are negligible.
3) The switches are ideal.
The input current ripple of the single-phase boost converter is expressed by multiplying the time by the rising
slope, or the falling slope, of the inductor current. The rising slope of the inductor current in the boost converter, and
the falling slope of the inductor current are expressed by
(1) and (2).
(1)
VoT
DD '
L
360°
OFF
Switching
Pattern
ON
S2
S3
I1
Inductor
Current
D
I2
D
I3
D'
T
d
tr
Iin
d'
ΔIin
τ
ΔVout
Vo
Output
Voltage
t1 t2
t0
(a)
0°
240°
120°
S1
ON
Switching
Pattern
360°
OFF
S2
S3
D
Inductor
Current
Input
Current
Output
Voltage
(2)
t4 t5
t3
(b)
where, D is the ON duty ratio of the switch, D ' is the OFF
duty ratio of the switch, and iL is the inductor current.
The input current ripple of the single-phase boost converter is expressed as shown in (3) [15].
ΔI in =
240°
120°
S1
Input
Current
2.2 Analysis of the Input Current Ripple at CCM
diL Vin
=
dt
L
diL Vin − Vo − DVin
=
=
dt
L
LD '
0°
(3)
Unlike the single-phase boost converter, the slope of the
input current ripple of a 3-phase IBC consists of three parts
according to the duty ratio. Fig. 2 depicts the input current
and the output voltage waveforms according to the duty
ratio where τ is the period of the input current, d and d '
are the rising and falling period of the input current , ret
spectively ( d = r , d ' = 1 − d ).
τ
In the case of 0<D<0.33, as shown in Fig. 2(a), only one
of the inductor currents is increased during t0 ~ t1 because
only one switch is turned on. As all the switches are turned
off during t1 ~ t2 , the inductor currents and the input currents are seen to decrease. In the other cases, such as
0.34<D<0.66 and 0.67<D<1, the variation of the input current ripple can be explained in similar ways. As a result,
0°
240°
120°
S1
360°
ON
Switching
Pattern
OFF
S2
S3
D
Inductor
Current
Input
Current
Output
Voltage
t6
t7 t8
(c)
Fig. 2. Input current and output voltage waveforms of 3phase IBC according to duty ratio: (a) 0<D<0.33
(b) 0.34<D<0.66 (c) 0.67<D<1.
the input current ripple can be expressed by (4), (5) and (6)
according to their duty ratios, and equations (7) and (8) can
be generalized by using (4), (5) and (6) as the functions of
input and output voltages, respectively.
Gyu-Yeong Choe, Jong-Soo Kim, Hyun-Soo Kang and Byoung-Kuk Lee
ΔI in =
Vin ⎛ 1 − 3D ⎞ T
d (0 < D < 0.33)
L ⎜⎝ D ' ⎟⎠ N
(4)
Vin ⎛ 2 − 3D ⎞ T
d (0.34 < D < 0.66)
L ⎜⎝ D ' ⎟⎠ N
V ⎛ 3 − 3D ) ⎞ T
ΔI in = in ⎜
d (0.67 < D < 1)
L ⎝ D ' ⎟⎠ N
ΔI in =
⎛ N on _ sw − ND ⎞ T
⎜⎜
⎟⎟ d
D'
⎝
⎠N
V
T
ΔI in = o N on _ sw − ND
d
L
N
ΔI in =
Vin
L
(
be depicted as shown in Fig. 4, and can be expressed by
(10) [12].
{
(5)
(6)
(7)
321
QC = ( N off _ sw + 1) I L − I o
} TN d '
(10)
Vo
Vo
is the output current, I L =
is the
RD ' N
R
average current of the inductor, and N off _ sw is the number
where, I o =
of OFF switches during τ .
)
(8)
Equation (11) represents the charge of the output current
ripple, and can be calculated using (12).
where, N is the number of the phase, T is the switching
period, and N on _ sw is the number of ON switches during τ .
Fig. 3 shows the variation of the input current ripple according to the duty ratio under the same inductors. The
input current ripple becomes zero at D/N times duty (for
example, at 0.33, 0.66 duty in 3ph-IBC), and is proportionally reduced according to the increase in the number of
phases. In the case of an even number of phases, the input
current ripple becomes zero at a 0.5 duty ratio in common
and above 3-phase, it is considerably reduced compared
with a single-phase boost converter.
Input c urrent ripple ac c ording to duty ratio
QC =
TVo dd '
(11)
N 2 RD '
Q
TVo dd '
Δvo = C =
C
RCN 2 D '
(12)
Fig. 5 shows the variation of the output voltage ripple
according to the number of phases under the same capacitance. From (12), it is noted that the output voltage ripple is
decreased by 1/N2 times according to the increase in the
number of phases, and it becomes zero at specific duty
ratios (at the D/N times). Therefore, in the case of the IBC
1
1 -P h
2 -P h
3 -P h
4 -P h
5 -P h
0.9
0.8
Input current ripple
0.7
V o C o n s ta n t
IB C
IB C
IB C
IB C
IB C
( N off _ sw + 1) I L
Io
QC
0.6
dτ
Io
0.5
0.4
τ
0.3
t
0.2
0.1
0
Fig. 4. Charge of output voltage.
0
0.1
0.2
0.3
0.4
0.5
0.6
Duty ratio
0.7
0.8
0.9
1
Output voltage ripple acc ording to duty ratio
0.35
Fig. 3. Input current ripple variation according to duty ratio.
1-Ph IBC
2-Ph IBC
3-Ph IBC
4-Ph IBC
5-Ph IBC
0.3
2.3 Analysis of the Output Voltage Ripple at CCM
Ic = I L − Io
(9)
where, I c is the capacitor current, and I o is the average
output current. The charge of the output capacitor, QC , can
Output voltage ripple
0.25
The output voltage ripple is determined by the output current
charge (Q), which is caused by the charging/discharging
currents of the capacitor. Even though the shape of the output voltage of the N-phase IBC is different, the average
charge is identical. The capacitor current can be obtained
by (9) which is the difference between the average current
of the inductor and the average output current.
Vo Constant
0.2
0.15
0.1
0.05
0
0
0.1
0.2
0.3
0.4
0.5
0.6
Duty ratio
0.7
0.8
0.9
1
Fig. 5. Variation of output voltage ripple according to duty
ratio.
322
An Optimal Design Methodology of an Interleaved Boost Converter for Fuel Cell Applications
with multiple phases, the output voltage ripple becomes
dramatically reduced compared with a conventional singlephase boost converter.
0°
240°
120°
S1
360°
OFF
Switching
Pattern
S2
2.4 Analysis of the Input Current Ripple at DCM
S3
The rising slope of inductor current at the DCM is same
as the rising slope of inductor current of single boost converter at the CCM. However, the falling slope of inductor
current is different and is expressed by (13).
− DVin
diL Vin − Vo
=
=
dt
L
L(t RF − D)
ON
(13)
Inductor
Current
t RF
Input
Current
ΔI in
ΔVout
Output
Voltage
(a)
where, t RF is the duration of the inductor current.
Using (13), the input current ripple of the single-phase
boost converter at the DCM is represented by (14).
VT
ΔI in = o D (t RF − D)
L
0°
S1
Switching
Pattern
(14)
Inductor
Current
Input
Current
Output
Voltage
(b)
0°
120°
S1
Vin ⎛ 2t RF − 3D ⎞ T
⎜
⎟ d
L ⎝ (t RF − D) ⎠ N
(16)
D
S3
Input
Current
Output
Voltage
(c)
0°
(17)
S1
Switching
Pattern
120°
ON
240°
OFF
S2
D
S3
(18)
Inductor
Current
Input
Current
• Case of constant falling slope at 0.67<D<1
V ⎛ 2t − 3D ⎞ T
ΔI in = in ⎜ RF
⎟ d'
L ⎝ (t RF − D) ⎠ N
S2
Inductor
Current
• Case of constant rising slope at 0.34<D<0.66
ΔI in =
360°
OFF
(15)
• Case of constant falling slope at 0.34<D<0.66
V ⎛ t − 2D ⎞ T
ΔI in = in ⎜ RF
⎟ d'
L ⎝ (t RF − D) ⎠ N
240°
ON
Switching
Pattern
• Case of constant rising slope at 0<D<0.33
V ⎛ t − 3D ⎞ T
ΔI in = in ⎜ RF
⎟ d
L ⎝ (t RF − D) ⎠ N
ON
S2
• Case of constant falling slope at 0<D<0.33
⎛ −2 D ⎞ T
⎜
⎟ d'
⎝ (t RF − D) ⎠ N
360°
OFF
S3
As shown in Fig. 6, the input current ripple can be categorized as five different cases according to duty ratio and
shape of rising and falling slope. Therefore, the input current
ripple should be calculated at a constant slope to obtain
accurate input current ripple. As a result, the input current
ripple at the DCM can be expressed by (15)-(19) according
to duty ratio.
V
ΔI in = in
L
240°
120°
(19)
Output
Voltage
(d)
360°
Gyu-Yeong Choe, Jong-Soo Kim, Hyun-Soo Kang and Byoung-Kuk Lee
0°
S1
Switching
Pattern
240°
120°
Input c urrent ripple ac c ording to duty ratio
360°
ON
323
1.5
1-Ph IBC: L
2-Ph IBC: L/2
3-Ph IBC: L/3
4-Ph IBC: L/4
5-Ph IBC: L/5
OFF
S2
Input current ripple
S3
Inductor
Current
Input
Current
1
0.5
Output
Voltage
0
(e)
The calculation method of output voltage ripple at DCM
is almost same as method at CCM. Equation (20) represents the output voltage ripple at DCM.
{
Vo d ' T t RF N off − (t RF − D) N
Q
Δvo = C =
C
RC (t RF − D) N 2
}
(20)
2.5 Design of the Inductor and Capacitor Elements at
the CCM
The input current ripple at the CCM of the IBC is the
sum of each inductor current. Therefore, a specific design
guideline is needed for selecting the inductor, compared to
the conventional single-phase boost converter. From (8),
the input current ripple of the IBC is inversely proportional
to the number of phases, and it can be noted that during the
duty ratio 0.4<D<0.7, the inductor at the L/N times is used
and the input current ripple becomes almost the same as
the single phase boost converter.
Therefore, the inductor of the IBC can be designed by
using the L/N times. Fig. 7 shows the variation of the input
current ripple for the N-phase IBC with 1/N times the inductance. On the other hand, during the duty ratio
0.4<D<0.7, the output voltage ripple is almost identical to
the single-phase boost converter.
With the same capacitance, in the IBC, the output voltage ripple is decreased by 1/N2 times from (12). Therefore,
in the IBC, the output capacitor can be reduced by 1/N2
times. Fig. 8 shows the variation of the output voltage ripple for the N-phase IBC with 1/N2 times capacitance.
0.1
0.2
0. 3
0.4
Duty ratio
0.5
0.6
0.7
0.8
Fig. 7. Variation of input current ripple according to inductance.
Output voltage ripple according to capacitor
0.45
1-Ph IBC: C
2-Ph IBC: C/4
3-Ph IBC: C/9
4-Ph IBC: C/16
5-Ph IBC: C/25
0.4
0.35
Output voltage ripple
Fig. 6. Input current and output voltage waveforms of 3phase IBC according to duty ratio at DCM: (a)
Case of constant falling slope at 0<D<0.33 (b)
Case of constant rising slope at 0<D<0.33 (c) Case
of constant falling slope at 0.34<D<0.66 (d) Case
of constant rising slope at 0.34<D<0.66 (e) Case of
constant falling slope at 0.67<D<1.
0
0.3
0.25
0.2
0.15
0.1
0.05
0
0
0.1
0.2
0. 3
0.4
Duty ratio
0.5
0.6
0.7
0.8
Fig. 8. Variation of output voltage ripple according to capacitance.
3. Design Example of the IBC
Based on the analysis presented in section II, a design
example of the IBC for fuel cell power generation is explained as follows where the design specification is that
output voltage of fuel cell stack is 27V~40V, the output
voltage of the IBC is 90V, and the power rating is 600W.
The duty ratio of the IBC can be calculated as 0.44~0.7
from (21).
D = 1−
Vin
Vo
(21)
Considering input current ripple and output voltage ripple during this specific duty ratio, one can select 2-phase,
3-phase, and 4-phase as shown in Figs. 9 and 10. For a the
fuel cell normally operated fuel cell at the rated power, the
3-phase and 4-phase IBCs have a lower minimum input
current ripple than the 2-phase IBC.
An Optimal Design Methodology of an Interleaved Boost Converter for Fuel Cell Applications
324
Consequently, for this application, 3-phase IBC can be
selected with respect of cost reduction.
Input current ripple variation (0.44<D<0.7)
1.5
Input current ripple
1-Ph IBC: L
2-Ph IBC: L/2
3-Ph IBC: L/3
4-Ph IBC: L/4
5-Ph IBC: L/5
1
0.5
Rating Power
0
0.45
0.5
0.55
Duty ratio
0.6
0.65
0.7
Fig. 9. Variation of input current ripple during 0.44<D<0.7.
Output voltage ripple variation (0.44<D<0.7)
0.3
Output voltage ripple
0.25
1-Ph IBC: C
2-Ph IBC: C/4
3-Ph IBC: C/9
4-Ph IBC: C/16
5-Ph IBC: C/25
0.2
0.15
0.1
0.05
Rating Power
0
0.45
0.5
0.55
Duty ratio
0.6
0.65
0.7
Fig. 10. Variation of output voltage ripple during
0.44<D<0.7.
error signal becomes the current reference through the
voltage PI controller. The final reference is generated after
the same sequence, and then compared with a 120° shifted
saw-toothed waveform, from which the 120° shifted PWM
of the 3-phase IBC is generated. The EPLD makes the 120°
shifted PWM of the 3-phase IBC.
Figs. 12 and 13 represent the simulation and experimental results of the 3-phase IBC, respectively, at 0.5, 0.6, and
0.66 duty ratios. As expected from Fig. 9, the input current
ripple is reduced, and it is dramatically decreased at the
0.66 duty ratio. Fig. 14 represents the experimental results
of the 3-phase IBC applied to the proposed optimal design
methodology where the voltage of the fuel cell is 27V and
current is at 23A and the output voltage and current of the
3-phase IBC is 90V and 6.6A respectively.
Fig. 15 shows the simulation and experimental results of
the inductor and input current ripple at 600W. Compared
with Figs. 15(a) and (b), it is noted that the simulation results have a good agreement with the experimental ones.
Finally, in order to certify the validity of the proposed
analysis and design methodology, all the results from the
theoretical analysis, simulations and experimental studies
have been compared, as shown in Fig. 16.
Using a YOKOGAWA WT3000 Power Analyzer, the efficiency of the 3-phase IBC is also measured and compared
with the single-boost converter under several operational
conditions and the results are summarized in Fig. 17.
As the result of the efficiency comparison, the efficiency
of the 3-phase IBC is about 2% higher than that of the single-phase boost converter. Fig. 18 shows the efficiency
map according to the each duty ratio in case of the singlephase boost converter and the 3-phase IBC. Table II shows
the estimated cost of the 3-phase and the single phase boost
converter.
As shown in Table 2, the estimated total cost of the 3phase IBC is more economical than that of the single boost
converter. From these comparisons, it is possible to verify
that the analysis and the proposed design methodology has
been well developed and can be utilized for the various
applications of designing and specifying the required IBCs.
4. Simulation and Experimental Results
In order to verify the analysis of the input ripple current
for the IBC, simulations and experimental studies are performed. A PSIM 6.0 is used as the simulation tool, with the
system parameters shown in Table 1. A Ballard NEXA
1.2kW PEMFC system is utilized to implement the experimental testbed.
Fig. 11 shows the 600W rated 3-phase IBC laboratory.
The controller of the 3-phase IBC consists of an inner current loop and an outer voltage loop configuration. The
outer voltage loop controls the voltage level to stay constant at 90V.
First, the output voltage is sensed and this becomes the
real voltage in the controller, and then it is compared with
the voltage reference to generate the voltage error. This
3-phase IBC
Control board
Fig. 11. Experimental testbed of 3-phase IBC.
Gyu-Yeong Choe, Jong-Soo Kim, Hyun-Soo Kang and Byoung-Kuk Lee
(a)
325
(b)
(c)
△Iin = 0.2A
△Iin = 0.315A
△Iin = 0.04A
Duty ratio = 0.5
Duty ratio = 0.6
Duty ratio = 0.66
Fig. 12. Simulation waveforms of ripple variation of the input current: (a) 0.5 duty ratio (b) 0.6 duty ratio (c) 0.66 duty ratio
(250mA/div, 50us/div).
(a)
(b)
(c)
△Iin = 0.334A
△Iin = 0.218A
△Iin = 0.05A
Duty ratio = 0.6
Duty ratio = 0.5
Duty ratio = 0.66
Fig. 13. Experimental waveforms of ripple variation of the input current: (a) 0.5 duty ratio (b) 0.6 duty ratio (c) 0.66 duty
ratio (200mA/div, 20us/div).
Input current
Table 1. Simulation Parameter
Parameter
Value
Fuel Cell Voltage
23-40 [V]
Output Voltage of 3-phase IBC
90 [V]
Rating Power
600 [W]
Switching Frequency
20 [kHz]
Inductor
1.19 [mH]
Capacitor
940 [uF]
Inductor current
Fuel Cell
Voltage
(a) Simulation waveforms
Fuel Cell
Current
23A
Converter
Output Voltage
7.6A
Input current
(500mA/div, 20us/div)
Inductor current
(1A/div, 20us/div)
Converter
Output Current
Fig. 14. Current and voltage waveforms of 600W rated 3phase IBC (Above: 10V/div., 10A/div., 20us/div.
Below: 50V/div., 5A/div., 20us/div.).
(b) Experimental waveforms
Fig. 15. Input current and inductor current waveforms.
An Optimal Design Methodology of an Interleaved Boost Converter for Fuel Cell Applications
326
Input current ripple
1.5
1.4
1.3
1.2
1.1
1
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
Experiment 1ph
Simulation 1ph
Theory 1ph
Experiment 3ph
Simulation 3ph
Theory 3ph
0
0.05
0.1
0.15
0.2
0.25
0.3
0.33 0.35 0.4 0.45
Duty ratio
0.5
0.55
0.6
0.66
0.7
0.75
0.8
Fig. 16. Comparison of theory, simulation, and experimental results.
Efficiency
Efficiency
100
Efficiency
100
98
100
3-phase IBC
98
3-phase IBC
96
94
96
94
92
Single-phase Boost
90
92
90
90
88
88
86
86
86
84
84
Duty = 0.74
80
60
120
180
240
300
360
420
480
540
84
82
82
Duty = 0.5
80
600
Single-phase Boost
94
Single-phase Boost
92
88
82
3-phase IBC
98
96
60
120
180
240
Power[W]
300
360
420
480
540
Duty = 0.35
80
600
60
120
Power[W]
(a)
180
240
300
360
420
480
540
600
Power[W]
(b)
(c)
Fig. 17. Efficiency comparison: (a) Duty = 0.74 (b) Duty = 0.5 (c) Duty = 0.35.
Table 2. Cost Comparison
Efficiency
100
98
3-phase IBC
96
94
92
Cost
(Unit: US$)
Inductor
3
15
Diode
3
2.19
Switch
3
Single-phase Boost
90
88
86
Single-phase Boost Converter
Quantity
Quantity
Cost
(Unit: US$)
Inductor
1
14.29
Diode
1
5.75
13.54
Switch
1
22.5
1
6.17
4
48.7
84
82
80
60
120
180
240
300
D=0.35
D=0.5
360
Power[W]
420
480
540
D=0.74
600
(a)
Efficiency
100
Capacitor
1
6.17
Capacitor
Total
10
36.9
Total
98
96
Source - http://dkc1.digikey.com/kr/digihome.html
94
3-phase IBC
92
90
88
5. Conclusion
86
84
82
80
60
120
180
240
300
D=0.35
D=0.5
360
Power[W]
420
480
540
D=0.74
600
(b)
Fig. 18. Efficiency mapping according to duty ratio: (a)
Single-phase boost converter (b) 3-phase IBC.
This paper has presented a detailed analysis and an optimal design methodology for effectively utilizing interleaved boost converters in fuel cell applications. Based on
the proposed method, the IBC with a minimum number of
switches and other parasitic components can be designed
and successfully operated to minimize input current and
output voltage ripples.
Gyu-Yeong Choe, Jong-Soo Kim, Hyun-Soo Kang and Byoung-Kuk Lee
Therefore, the proposed design methodology can be
used in various applications that use fuel cells and photovoltaic devices with broad system specifications from low
to high power situations.
References
J. M. Carrasco et al., “Power-electronic systems for
the grid integration of renewable energy sources: A
survey,” IEEE Trans. Ind. Electron., Vol. 53, Issue 4,
pp. 1002-1016, Jun., 2006.
[2] U.S Department of Energy, Summary of annual merit
review fuel cells subprogram, [Online]. Available:
http://www.hydrogen.energy.gov/pdfs/review07/4207
2-05_fuel_cells.pdf
[3] S. J. Jang, T. W. Lee, K. S. Kang, S. S. Kim, C. Y.
Won, “A new active clamp sepic-flyback converter
for a fuel cell generation system,” in Proc. Ind.
Electron. Soc., pp. 2538-2542, 2005.
[4] T. F. Wu, Y. S. Lai, J. C. Hung, Y. M. Chen, “Boost
converter with coupled inductors and buck-boost type
of active clamp,” IEEE Trans. Ind. Electron., Vol. 55,
Issue 1, pp. 154-162, Jun., 2008.
[5] P. W. Lee, Y. S. Lee, D. K. Cheng, X. C. Liu,
“Steady-state analysis of an interleaved boost
converter with coupled inductors,” IEEE Trans. Ind.
Electron., Vol. 47, Issue 4, pp. 787-795, Aug., 2000.
[6] L. Huber, B. T. Irving, M. M. Jovanovic, “Open-loop
control methods for interleaved DCM/CCM
boundary boost PFC converters,” IEEE Trans. on
Power Elect., Vol. 23, Issue 4, pp. 1649-1657, July,
[7] Y.
2008.
Chen, K. M. Smedley, “Parallel operation of onecycle controlled three-phase PFC rectifiers,” IEEE
Trans. Ind. Electron., Vol. 54, Issue 6, pp. 3217-3224,
Dec., 2007.
[8] B. A. Miwa, D. M. Dtten, M. F. Schlecht, “High
efficiency power factor correction using interleaving
technique,” IEEE Pro. APEC'92, Vol. 1, pp. 557-568,
Feb., 1992.
[9] H. B. Shin, E. S. Jang, J. K. Park, H. W. Lee, T. A.
Lipo, “Small-signal analysis of multiphase
interleaved boost converter with coupled inductor,”
IEE Proc. Electr. Power Appl., Vol. 152, No. 5, pp.
1161-1170, Sept., 2005.
[10] J. J. Lee, B. H. Kwon, “Active-clamped ripple-free
DC/DC converter using an input-output coupled
inductor,” IEEE Trans. Ind. Electron., Vol. 55, Issue 4,
pp. 1842-1854, April, 2008.
[11] R. J. Wai, C. Y. Lin, R. Y. Duan, Y. R. Chang, “Highefficiency DC-DC converter with high voltage gain
and reduced switch stress,” IEEE Trans. Ind.
Electron., Vol. 54, Issue 1, pp. 354-364, Feb., 2007.
[12] H. B. Shin, J. G. Park, S. K. Chung, H. W. Lee, T. A.
Lipo, “Generalized steady-state analysis of
multiphase interleaved boost converter with coupled
inductors,” IEE Electr. Power Appl., Vol. 152, Issue 3,
pp. 584-594, May, 2005.
[1]
327
[13] M. H. Todorovic, L. Palma, P. N. Enjeti, “Design of a
wide input range DC-DC converter with a robust
power control scheme suitable for fuel cell power
conversion,” IEEE Trans. Ind. Electron., Vol. 55, Issue
3, pp. 1247-1255, March, 2008.
[14] C. T. Pan, Y. H. Liao, “Modeling and control of
circulating currents for parallel three-phase boost rectifiers with different load sharing,” IEEE Trans. Ind.
Electron., Vol. 55, Issue 7, pp. 2776-2785, July, 2008.
[15] N. Mohan, T. M. Undeland, W. P. Robbins, “Power
electronics converter,” application and design, 3rd ed.
JOHN WILEY & SONS, INC., ch.7, 2003
Gyu-Yeong Choe received the M.S.
degrees from Sungkyunkwan University,
Suwon, Korea, in 2008, in electrical
engineering.
From 2008, he is working toward the
Ph.D. degree at electrical engineering,
Sungkyunkwan University. His research
interests include renewable energy
source modeling, renewable energy hybrid system, and
battery charger for PHEV/EV and interleaved dc-dc converters.
Jong-Soo Kim received the B.S.
degrees from Seoul National University
of Technology, Seoul, Korea, in 2006,
and received the M.S. degree from
Sungkyunkwan University, Suwon,
Korea, in 2008, all in electrical
engineering.
From 2008, he is working toward the
Ph.D. degree at electrical engineering, Sungkyunkwan
University. His research interests include eco-friendly vehicle technologies, power conditioning systems for renewable energy and PM motor drives.
Hyun-Soo Kang received the B.S. and
the M.S. degrees from Hanyang University, Seoul, Korea, in 1994 and
1996, respectively, and the Ph.D.
degree from Sungkyunkwan University,
Suwon, Korea, in 2008, all in electrical
engineering.
From 1996 to 1999, he has been an
Associate Research Engineer at Power Electronics Lab.,
LGIS R&D Center, Anyang, Korea. From 2000 he joins at
ADT co., Ltd, and now he is a Principal Engineer in R&D
Center, ADT co., Ltd. His research interests include sensorless drives for IM and PM motor drives, power conditioning systems for renewable energy sources and power
electronics.
328
An Optimal Design Methodology of an Interleaved Boost Converter for Fuel Cell Applications
Byoung-Kuk Lee received the B.S.
and the M.S. degrees from Hanyang
University, Seoul, Korea, in1994 and
1996, respectively and the Ph.D. degree from Texas A&M University,
College Station, TX, in 2001, all in
electrical engineering.
From 2003 to 2005, he has been a Senior Researcher at Power Electronics Group, KERI, Changwon, Korea. From 2006 Dr. Lee joins at School of Information and Communication Engineering, Sungkyunkwan
University, Suwon, Korea, as an Assistant Professor. His
research interests include electric vehicles, sensorless
drives for high speed PM motor drives, power conditioning
systems for renewable energy, modeling and simulation,
and power electronics.
Prof. Lee is a recipient of Outstanding Scientists of the
21stCentury from IBC and listed on 2008 62nd Ed. of
Who’s Who in America. and 2009 26th Ed. of Who's Who
in the World. Prof. Lee is an Associate Editor in the IEEE
Transactions on Industrial Electronics and is the IEEE Senior Member.