Robot and Servo Drive Lab.
Sensorless Control of BLDC Motor Drive
for an Automotive Fuel Pump Using
a Hysteresis Comparator
Tae-Won Chun, Member, IEEE, Quang-Vinh Tran, Hong-Hee Lee, Senior Member, IEEE,
and Heung-Geun Kim, Senior Member, IEEE
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 3, MARCH 2014
Advanced Servo Control
11/13/2014
Student: T A R Y U D I / 林運昇
Advisor: Prof. Ming Shyan Wang
Department of Electrical Engineering
Southern Taiwan University of Science and
Technology
Outline
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
Abstract
Introduction
Sensorless control using
hysteresis comparator
Start-up technique
Alignment of Rotor Position
 Start-Up Procedure
Hardware and Software
Configurations
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Experimental results
Conclusion
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Abstract




The brushless dc (BLDC) motor sensorless control system for an automotive
fuel pump that are based on a hysteresis comparator and a potential startup method with a high starting torque.
The hysteresis comparator is used to compensate for the phase delay of
the back EMFs due to a low-pass filter (LPF) and also prevent multiple
output transitions from noise or ripple in the terminal voltages.
The rotor position is aligned at standstill for maximum starting torque
without an additional sensor and any information of motor parameters. Also,
the stator current can be easily adjusted by modulating the pulse width of the
switching devices during alignment.
Some experiments are implemented on a single chip DSP controller to
demonstrate the feasibility of the suggested sensorless and start-up
techniques
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Introduction


The brushless dc (BLDC) motor is receiving much interest in automotive
applications especially on vehicle fuel pumps due to its high efficiency, compact
size, and lower maintenance when compared to a brush dc motor [1], [2].
The requirements:


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
Good reliability
wide speed range from 3000 to 9000 rpm
fast start up
high starting torque.
In order to obtain an accurate and ripple-free instantaneous torque of the BLDC
motor, the rotor position information for stator current commutation must be
known, which can be obtained using hall sensors mounted on a rotor [3], [4]
 This results in a high costs as well as poor reliability, which are serious
problems at the vehicle applications.
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Introduction
To cope with the aforementioned restriction, many position sensorless
algorithms have been considered as potential solutions

The zero-crossing of the back EMF
measured from the stator winding is
detected and the commutation points
can be estimated by shifting 30° from
the zero crossing of the back EMFs [5].

The performance of the sensorless
drive deteriorates with the phase
shifter in the transient state. Also, it
is sensitive to the phase delay of the
low-pass filter (LPF) especially at
the high speed.
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Conventional sensorless commutation [30]
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Phase terminal voltage and back EMF waveform [30]
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Introduction

Several phase shifters to compensate for phase error induced by
the LPF of back EMFs are proposed [6], [7], [8], [9].


They require an additional compensation circuit including
the timers.
The position information is extracted by integrating the back
EMF of the silent phase [1], [10], [11].

This method has an error accumulation problem at low speed.
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Introduction

The sensorless control techniques using the phase-locked loop (PLL)
and the third-harmonic back EMF are suggested [12], [13].
 The motor commutation drifts away from the desired phase angle
due to the conduction of the freewheel diode.

Furthermore, the drift angle varies as the motor parameters, speed, and
load conditions change. The improved sensorless controller by
removing the effect of the freewheel diode conduction is suggested
[14].
 Access to the motor's neutral point is required, which will
complicate the motor structure and increase the cost.
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Introduction


Some approaches use the zero crossing points of three-phase line-toline voltages, so that they coincide to six commutation points [15],
[16], [17].
Although the commutation signals can be obtained without any
phase shifter, the phase delay due to the LPF could not be
considered and the multiple output transitions of the comparator
may occur from the high frequency ripple or noise in the back EMFs.
The zero-crossing point of the back EMF for generating proper
commutation control of the inverter is calculated by sampling the
voltage of the floating phase without using current and position
sensors [18], [19].
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Introduction

Most sensorless techniques are based on back EMF estimation.
However, when a motor is at standstill or very low speed, it is well
known that the back EMF is too small to estimate a precise rotor
position.

Therefore, a specific start-up process in sensorless
drive systems is required.
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Introduction

The general solution to the problem is the open-loop start-up method
named “align and go” [5], [20].
 The procedure is to excite two phases of the three-phase windings
for a preset time. The permanent magnet rotor will then rotate to
align to a specific position. With a known initial rotor position and
a given commutation logic, an open-loop control scheme is then
applied to accelerate the motor from a standstill. Although this
technique can be applied to certain automotive applications, it
causes a large instantaneous peak current and generates a
temporary vibration.
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Introduction

In addition, the rotor position of the BLDC motor can be identified
and driven smoothly from standstill without any position sensors by
utilizing the inductance variation technique [21], [22], [23].

This is done by monitoring the current responses to the inductance
variation on the rotor position. Although these methods can detect a
precise rotor position at standstill, they result in a complex control
algorithm and an increase of the system costs due to an additional
current sensor.

Instead of the detection of current, only terminal voltage level is used
for detection of the initial position of the permanent magnet [24].
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Introduction


Some drawbacks of the aforementioned sensorless and start-up
techniques are restricted in an automotive fuel pump application
which requires the good reliability, a wide speed range from
3000 to 9000 rpm, fast start up, and high starting torque for the
sensorless BLDC motor drive systems.
To satisfy these requirements, this paper presents a sensorless control based on a
hysteresis comparator of terminal voltage and a potential start-up method with a
high starting torque.
The hysteresis comparator is used to compensate for the phase lag due to the LPF
and also to prevent multiple output transitions from noise or ripple in the terminal
voltages.
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SENSORLESS CONTROL USING
HYSTERESIS COMPARATOR
As only two phases of the BLDC motor are energized at any time, the back EMF can be
measured from its terminal voltage in the period of an open phase (60°).
During the two-phases conduction period (120°), the only difference between the back
EMF and its terminal voltage is a stator impedance voltage drop, which may be
considerably small compared with the dc voltage source.
Therefore, the waveform of the
terminal voltage is nearly the same as
that of the back EMF.
The terminal voltages can be used to
detect the commutation points of the
BLDC motor instead of the back
EMFs at the proposed sensorless
control.
Block diagram of sensorless control by using
a hysteresis comparator
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Hysteresis Comparator


The hysteresis comparator is used to
compensate for the phase lag of the back
EMFs due to the LPF in order to
determine the proper commutation
sequence of the inverter according to the
rotor position.
Also, it can prevent multiple output
transitions by high frequency ripples in the
terminal voltages. The outputs of the
three-phase hysteresis comparators
become three commutation signals
(Za,Zb,Zc), and then six gating signals can
be generated through some logic
Block diagram of sensorless control by
equations.
using a hysteresis comparator
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Plots of phase lag caused by LPF

The lag will disturb current
alignment with the back EMF
and will cause serious problems
for commutation at high speed.

The phase lag in commutation
can produce significant pulsating
torques in such drive which may
cause oscillations of the rotor
speed, and generate extra
copper losses.
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Plots of phase lag to various rotor speeds under a
variation of the cut-off frequency
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The operation of the a-phase hysteresis comparator

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The filtered a-phase terminal voltage is applied
to the inverting input, and the filtered c-phase
terminal voltage is applied via R1 to the non
inverting input.
where Voa is the output voltage of a-phase
hysteresis comparator and resistance
ratio n=R2/R1.
At Voa = +Vsat, the output voltage Voa switches
to (-Vsat) , when the differential
1
n
voltage Vda at (1) is negative, i.e., the
Vda  uaf 
Voa 
ucf (1)
n 1
n 1
following condition is satisfied for the
filtered c-phase terminal voltage ucf :


1
1
ucf   1  uaf   Vsat  (2)
n
n


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Vda : A differential voltage
of a-phase hysteresis
comparator
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The operation of the a-phase hysteresis comparator
a-phase hysteresis comparator

At Voa=(-Vsat) , the output voltage Voa switches back
to +Vsat, when the differential voltage Vda at (1) is positive,
i.e., the following condition is satisfied for ucf :


1
1
ucf   1  uaf   Vsat  (3)
n
n


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The operation of the a-phase hysteresis comparator

The term (1/n)Vsat in (2) and (3) is defined as a hysteresis band,
which is determined by the output voltage level Vsat and resistance
ratio n. The right sides of (2) and (3) are the lower threshold
voltage VLT and the upper-threshold voltage VUT for the a-phase
terminal voltage multiplied by (1+1/n), respectively [25]. Both
threshold voltages can be given by


1
1
ucf   1  uaf   Vsat  (2)
n
n


VUT


1
1
ucf   1  uaf   Vsat  (3)
n
n


1

1
VLT  1  uaf   Vsat (5)
n

n
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1

1
 1  uaf   Vsat (4)
n

n
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The operation of the a-phase hysteresis comparator

When the output voltage is at +Vsat and ucf is smaller than VLT, the output
voltage switches to -Vsat. Then, the output voltage stays at -Vsat as long
as is below with respect to VUT. When ucf is higher than VUT, the output
voltage switches back to +Vsat. Because the original commutation signal Za′
is generated by the filtered terminal voltages, it lags to the proper
commutation point. The a-phase commutation signal Za extracted from the
output of the hysteresis comparator leads to the original commutation
signal Za’ by time ta. The equation of VLT can be expressed as a function of
time as
12Vp 
1 1
VLT (t ) 
1   t   Vsat (6)
Ts 
n n

where Ts is one period of the terminal voltage, and Vp is the peak of the
terminal voltage.
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The operation of the a-phase hysteresis comparator

Using (6), the time tc at the cross
point between VLT and ucf can be
derived as follows:
T  n
1 Vsat 
 (7)
tc  s 


12  n  1 n  1 Vp 

The time ta is the interval from
( Ts/12) to tc as
Ts
t a   tc
(8)
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By substituting (7) into (8), the
advanced angle θa can be derived as

Ts 1  Vsat 
1 

 a   ta 
12 n  1 
Vp 
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(9)
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The operation of the a-phase hysteresis comparator



The advanced angle θa is determined by the
values of n, Vsat, and Vp, where the Vp is nearly
proportional to the rotor speed. This figure plots
of the angle θa for various rotor speeds under
various resistance ratios, when the Vsat is
+1.2 V.
The θa increases as the resistor ratio decreases
or the rotor speed decreases. Therefore, the
phase lag by LPF at the overall speed range can
be compensated by adjusting the resistance
ratio of the hysteresis comparator.
The resistance ratio is determined to 1.2 in order
that the gating signal can be nearly kept in
phase with the back EMF when the motor is at
the nominal speed, 6000 rpm.
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Plots of advanced angle to various rotor
speeds under various resistanceratios.
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The operation of the a-phase hysteresis comparator

This Figure shows the plots of the phase delay
due to the LPF with 2.5 kHz cut-off frequency,
the advanced angle by the hysteresis
comparator with 1.2 resistance ratio, and the
phase shift of the terminal voltage after
compensating for the phase lag by the
hysteresis comparator. It can be seen that
although the phase lag by the LPF ranges from
−4.5° to −13°, the phase shift after
compensation ranges only from −3° to +2°.
Thus, the maximum commutation delay at the
9000 rpm rotor speed is significantly reduced
from −13° to −3°.
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Plot of the phase delay, the advanced
angle, and the phase shift after
compensation
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The operation of the a-phase hysteresis comparator

The logic equations for
generating six gating signals of
three-phase PWM inverter from
three commutation signals can
be derived as
A  ( Z a  Zb )  Z a , A  ( Z a  Zb )  Z a
B   ( Zb  Zc )  Zb , B   ( Zb  Zc )  Zb
(10)
C   (Zc  Z a )  Zc , C   (Zc  Z a )  Zc
Timing diagram for commutation signals and threephase gating signals relative to the terminal voltages
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
The a-phase filtered terminal voltage
lags to the unfiltered terminal voltage by
the phase θd. Three commutation signals
are advanced by the angle θa to the
commutation points estimated by the
three-phase filtered terminal voltages,
and they can be compensated for the
phase delay θd by the LPF. Because the
waveform of the filtered terminal
voltage is nearly the same as that of the
filtered back EMF, the gating signal of
each phase bridge of the inverter is
almost in phase with the each phase of
the back EMF.
Timing diagram for commutation signals and threephase gating signals relative to the terminal voltages
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Block diagram for adjusting Vsat

This figure shows the circuit block diagram for adjusting the output voltage of
hysteresis comparator Vsat linearly to the battery output voltage change in order that
the θa maintains constant. The circuit consists of an inverter amplifier with
−0.1amplication ratio, an inverting amplifier with minus unity gain for converting
from -Vsat to +V sat, and two buffers.
Block diagram for adjusting Vsat linearly to the battery voltage change
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START-UP TECHNIQUE

When the motor is at standstill or
very low speed, the back EMF is too
small to estimate a precise rotor
position. Therefore, a specific startup process in sensorless drive
systems is needed
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Alignment of rotor position. (a) Switching states
of the inverter. (b) Initial rotor position.
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A. Alignment of Rotor Position



In the BLDC motor, only two phases of
the three-phase stator windings are
excited at any time by utilizing
alternative six excited voltage
vectors V1 – V6,
That is why the current can flow into
only two of the three windings and
commutated every 60° of electrical
angle.
At standstill, the initial rotor position is
aligned into one of the six positions that
are determined by the six excited
voltage vectors to energize two phases
of the BLDC motor [20], [26], [27], [28]
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Alignment of rotor position.
(a) Switching states of the inverter.
(b) Initial rotor position.
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A. Alignment of Rotor Position

all three stator windings are
energized in the case of the proposed
start-up scheme by using a specific
initial voltage vector (1,0,0). As the
rotor is located between voltage
vector and , the voltage vector is
orthogonal to . It is chosen as the
next applied voltage vector in order
to achieve maximum starting motor
torque at startup.
Alignment of rotor position.
(a) Switching states of the inverter.
(b) Initial rotor position.
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A. Alignment of Rotor Position

It should be noted that the amplitude
of the stator current for alignment of
the rotor position can be easily
adjusted by modulating the pulse
width of the switching devices

This Figure depicts the three-phase
current responses of the BLDC
motor, when turning on switch + and
modulating the pulse-width of the
Stator currents responses under alignment
two switches B− and C− at the same of rotor position
time
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A. Alignment of Rotor Position



This Figure shows minimum, maximum,
and average values of the a-phase stator
current with a variation of the duty cycles
of PWM signals when a switching
period is 100 μs.
Obviously, the magnitude of the stator
current can be easily governed by
adjusting the duty cycle, which can be
decided by considering the initial torque
required at alignment.
This method can prevent a surge of
current that may damage the motor as in
the case of utilizing the conventional
method, and also it is robust with motor
parameter changes.
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Magnitude of stator current to various duty
cycles
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B. Start-Up Procedure




After aligning the rotor position, the start-up procedure is
considered for accelerating the BLDC motor from standstill up to
a specific speed, 3000 rpm which is a minimum speed at the
automotive fuel pump application.
As the sensorless scheme is not self-starting, the motor should be
started and can be brought to a certain speed at which the zerocrossing point of the back EMF can be detected.
The v/f starting method is used at this paper, which is suitable for
the BLDC motor drive.
The open loop starting based on / control is accomplished by
producing the rotating electric field with a specific relationship to
the reference voltage in terms of a rotor speed.
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B. Start-Up Procedure


As the frequency is gradually increased, the rotor speed also
increases. The magnitude of a reference voltage is adjusted as
proportional to the rotor speed.
A phase angle can be obtained from integrating the rotor speed
and the pulse width of the gating signals is modulated with the
reference voltage magnitude. The six PWM signals with 60° phase
displacement are generated corresponding to the phase angle
without any rotor position information. When the rotor speed
reaches at 3000 rpm, the back EMF can be sensed to provide the
rotor position information and the system is switched to the
sensorless control.
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Experimental Setup
(a)
Experimental setup. (a) Hardware configuration.
(b) Photograph of the experimental setup
(b)
The control system is implemented by a 16-bit DSP type TMS320LF2406 operating with a clock
frequency of 40 MHz and the sampling interval is 50 μs for both the start-up and sensorless
controls. The DSP generates six PWM signals and also measures a rotor speed by using the threephase commutation signals. The reference speed can be changed from the host computer through
a RS232 C serial port. a 12-bit 4-channel D/A converter and displayed on an oscilloscope.
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BLDC Parameter
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Software Flowchart

The rotor is aligned to the initial
position for a time interval of
80 ms with constant duty cycle,
and then the motor is accelerated
to 3000 rpm by the proposed
start-up method. When the rotor
speed is greater than 3000 rpm, a
sensorless control scheme for the
BLDC motor is applied
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Experiment results

This Figure shows the a-phase and c-phase filtered terminal voltages, Za,
and a stator current at the light load condition when the rotor speed is 3000
(a) and 6000 (b)
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Experiment results

This Figure shows the a-phase and c-phase filtered terminal
voltages, Za =10.7º, and a stator current at the light load condition
when the rotor speed is 9000 rpm (c)
Plot of the phase delay, the advanced
angle, and the phase shift after
compensation
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Experiment results

This figure shows the unfiltered
and filtered a-phase terminal
voltages and gating signals of
the a-phase switching devices
at = 6000 rpm. Although the
filtered terminal voltage lags
the nonfiltered terminal
voltage, A+ and A− both are
nearly in phase with the actual
terminal voltage
A-phase terminal voltages and gating signals at
ωr = 6000 rpm.
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Experiment results

This Figure shows the unfiltered
and filtered a-phase terminal
voltages and stator current at
heavy load when the rotor speed
is 6000 rpm. The stator current is
increased and the conducting
period of the freewheeling diode
is extended a little bit.
Terminal voltages and stator current at heavy
load when the rotor speed is 6000 rpm
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Experiment results
PWM signals and stator current response under aligning rotor position.
(a) At 7% duty cycle. (b) At 15% duty cycle

The switching device A+ is always conducted and both switches B− and C−
are modulated by 7% and 15% duty cycles, respectively. The average values
of stator current are about 0.8 and 4 A when the duty cycle is 7% and 15%,
respectively.
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Experiment results

It can be seen that both of
average values of stator current
are similar with those as shown
in this Figure and the magnitude
of stator current can be easily
controlled by the duty cycle for
aligning a rotor position.
Magnitude of stator current to various duty
cycles
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Experimental result
Start-up currents with a variation of the initial angle between stator
and rotor fluxes under the light load condition. (a) At initial angle = 90◦. (b)At
initial angle = 60◦. (c) At initial angle = 0◦.
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Experimental result
Start-up currents with a variation of the initial angle between stator
and rotor fluxes under the light load condition. (c) At initial angle = 0◦.
The system is switched from the
start-up mode to the sensorless
control mode, when the rotor speed
reaches at 1500 rpm. It can be seen
that the start-up current at the initial
angle = 90° is lowest, and the start-up
current increases as the initial angle
decreases to 0° under both load
conditions. The start-up current at the
same angle is higher under the heavy
load condition
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Experimental result
Start-up currents with a variation of the initial angle between stator and rotor fluxes
under the heavy load condition. (a) At initial angle = 90◦. (b) At initial angle = 60◦. (c)
At initial angle = 0◦.
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Experimental result
Start-up currents with a variation of the initial angle between stator and rotor fluxes
under the heavy load condition .(c) At initial angle = 0◦.
The system is switched from the start-up
mode to the sensorless control mode,
when the rotor speed reaches at
1500 rpm. It can be seen that the start-up
current at the initial angle = 90° is
lowest, and the start-up current increases
as the initial angle decreases to 0° under
both load conditions. The start-up
current at the same angle is higher under
the heavy load condition
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Experimental result




The responses of the reference and rotor
speeds, reference voltage, and -phase
current in order to verify the start-up
technique.
At first, the rotor is aligned to the initial
position for a time interval of 80 ms by
adjusting the duty cycle to 15%.
Then, the motor is accelerated to
3000 rpm by the proposed start-up
method.
Subsequently, a sensorless control
scheme for the BLDC motor is applied
for speeding up the motor to 6000 rpm.
The start-up time is about 1.2 s, which is
acceptable for vehicle fuel pump
application.
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Fig. 19. Transient responses from the startup mode to the sensorless mode
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CONCLUSION



The maximum commutation phase lag is significantly reduced
from −13° to −3° by adjusting both the resistance ratio and the
output voltage level of the hysteresis comparator,
The commutation signal is nearly in phase with the back EMF.
If a peak of ripple voltage in the terminal voltage is within the
hysteresis band +1 V regardless of magnitude of the terminal
voltage, it can prevent multiple output transitions at a
hysteresis comparator by high frequency ripples in the
terminal voltage.
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CONCLUSION



After aligning the rotor position for achieving the maximum
starting torque, the BLDC motor accelerates from a standstill
up to a nominal speed within 1.2 s.
The magnitude of the stator current for aligning the rotor
position can be easily controlled by modulating the pulse
width of specific switching devices.
Through the experimental results, it can be seen that the
proposed sensorless and start-up techniques are ideally suited
for the automotive fuel pump application.
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Thank you for your attention
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Department of Electrical Engineering
Southern Taiwan University