LEFT-HANDED METAMATERIAL INCORPORATED WITH
MICROSTRIP ANTENNA
HUDA BIN A. MAJID
UNIVERSITI TEKNOLOGI MALAYSIA
i
LEFT-HANDED METAMATERIAL INCORPORATED WITH MICROSTRIP
ANTENNA
HUDA BIN A. MAJID
A thesis submitted in fulfilment of the
requirements for the award of the degree of
Master of Engineering (Electrical)
Faculty of Electrical Engineering
Universiti Teknologi Malaysia
FEBRUARY 2010
iii
To my beloved father and mother for their understanding and support throughout my
quest in completing my Master
iv
ACKNOWLEDGEMENT
First of all, thanks to our creator for the continuous blessing and for giving
me the strength and chances in completing this project.
Special thanks to my project supervisor, Associate Prof. Dr. Mohamad Kamal
A. Rahim, for his guidance, support and helpful comments in doing this project.
My family deserves special mention for their constant support and for their
role of being the driving force towards the success of my project. My friends
deserve recognition for lending a helping hand when I need them. I would also like
to thank the wonderful members of P18; Dr Thelaha Masri, Mr. Mohd Nazri A
Karim, Mr. Osman Ayop, Mr Farid Zubir, Mrs. Maisarrah Abu, and my middle east
friends; Mrs. Mai Abdul Rahim, Mr. Alii, Mr. Hassien, Mr. Hithem, Mr. Akram, Mr.
Suhil and Mr. Kusay, who have also been extremely kind and helpful throughout my
stay. “We don’t remember days, but we remember moments” and my moments with
these guys are very fruitful during my study in UTM.
My sincere appreciation also goes to everyone whom I may not have
mentioned above who have helped directly or indirectly in the completion of my
project.
v
ABSTRACT
Left-handed metamaterial (LHM) is an artificial material where the
permittivity and permeability are simultaneously negative at a certain range of
frequency. One of the unique properties of the LHM is its negative refraction index
which produces focusing effect to the wave propagating through the LHM. With this
unique property, the LHM structures are used to increase the low gain of the
microstrip antenna. In this work, an LHM structure consists of a modified split ring
resonator (MSRR) and two capacitance loaded strip (CLS) is proposed. The MSRR
has four slots in the middle of the structure which create wider range of negative
permittivity and permeability. The MSRR exhibits negative permeability while the
CLS exhibits negative permittivity. The well-known modified Nicolson-Ross-Wier
(NRW) approach has been used to determine the values of permittivity and
permeability. Parametric study on the parameters of the LHM has been carried out.
The gap between the CLS – MSRR, the MSRR length and the CLS width show
strong influence to the resonant frequency and the range of negative permittivity and
permeability. A series of LHM structures are then incorporated with different
antenna type such as single patch antenna, linear polarized 2x2 array patches antenna
and circular polarized 2x2 array patches antenna at operating frequency of 2.4 GHz.
The simulation and measurement results such as return loss, bandwidth, gain, half
power beamwidth and radiation pattern are analyzed. The gain of the antennas
increased upto 4 dB while the half power beamwidth decreased upto 37 % and
became directional. The bandwidth of the antennas also increased upto 60 %.
vi
ABSTRAK
Metabahan tangan kiri (LHM) adalah satu bahan buatan yang mana
kebolehtelusan dan kebolehtelapan bahan itu adalah negatif pada julat frekuensi.
Salah satu sifat unik LHM adalah indeks pembiasan negatif yang menghasilkan
kesan fokus terhadap gelombang yang melalui LHM. Dengan sifat unik ini, LHM
digunakan untuk mengatasi gandaan rendah pada antena mikrojalur. Struktur LHM
yang terdiri daripada satu penyalun cincin terpisah (MSRR) yang telah dimodifikasi
dan dua kapasitans jalur muatan (CLS) dicadangkan. MSRR mempunyai empat
celahan di pertengahan struktur dan ia mewujudkan julat kebolehtelusan dan
kebolehtelapan negatif yang lebih luas. MSRR mempamerkan kebolehtelapan negatif
sementara CLS mempamerkan kebolehtelusan negatif. Pendekatan terkenal
Nicolson-Ross-Wier (NRW) yang dimodifikasi telah digunakan untuk menetukan
nilai kebolehtelusan dan kebolehtelapan. Kajian parametrik pada parameter LHM
telah dilakukan. Lebar celahan antara CLS - MSRR, panjang MSRR dan lebar CLS
menunjukkan
pengaruh
yang
kuat
kepada
frekuensi
resonans
dan
julat
kebolehtelusan dan kebolehtelapan negatif. Satu rangkaian struktur LHM
kemudiannya digabungkan dengan beberapa jenis antena seperti satu antena tampal,
2x2 antena tampal tatasusun linear berkutub dan 2x2 antena tampal tatasusun
berkutub bulat beroperasi pada frekuensi 2.4 GHz. Hasil keputusan dari simulasi dan
pengukuran seperti kehilangan balikan, lebarjalur, gandaan, lebaralur setengah kuasa
dan corak sinaran dianalisis. Gandaan antena meningkat sehingga 4 dB manakala
lebaralur setengah kuasa menurun sehingga 37 % dan menjadi terarah. Lebarjalur
antena juga meningkat sehingga 60 %.
vii
TABLE OF CONTENTS
CHAPTER
TITLE
DECLARATION
ii
DEDICATION
iii
ACKNOWLEGMENT
iv
ABSTRACT
v
ABSTRAK
vi
TABLE OF CONTENTS
vii
LIST OF TABLES
xii
LIST OF FIGURES
xv
LIST OF SYMBOLS
LIST OF ABBREVIATIONS
1
2
PAGE
xxvi
xxviii
INTRODUCTION
1.1
Introduction
1
1.2
Problem Statement
3
1.3
The Scope of Research
3
1.4
The Objective of Research
4
1.5
Organisation of Thesis
4
LITERATURE REVIEW ON LEFT HANDED
METAMATERIAL AND INCORPORATION WITH
MICROSTRIP ANTENNA
2.1
Introduction
2.2
Definition & Background of
Left-Handed Metamaterial
6
6
viii
2.3
History of Left-Handed Metamaterial (LHM)
7
2.4
Left-Handed Metamaterial Structure
8
2.4.1
Split Ring Resonator (SRR)
8
2.4.2
Capacitance Loaded Strip (CLS)
9
and Thin Wire (TW)
2.5
2.6
Left-Handed Metamaterial Characteristics
11
2.5.1
Negative Refraction
11
2.5.2
Backward-wave Propagation
13
Method to Determine the Value of
Permittivity and Permeability Using Modified
2.7
Nicolson-Ross-Wier (NRW) Approach
16
Different Types of Left-Handed Metamaterial
19
2.7.1
Split Ring Structure
19
2.7.2
Symmetrical Ring Structure
20
2.7.3
Omega Structure
21
2.7.4
S-Shape Structure
22
2.7.5
Split Ring with Capacitance Loaded
2.7.6
2.8
Strip Structure
23
Summary
24
LHM Application in Antenna Technology
2.8.1
25
Theoretical Investigation of a Circular
Patch Antenna in the presence of a
Left-Handed Medium
2.8.2
Metamaterial Enhanced Patch Antenna
for WiMAX Application
2.8.3
25
28
A Study of Using the Double Negative
Structure to Enhance the Gain of
2.8.4
2.9
Rectangular Waveguide Antenna Array
31
Summary
34
Chapter Summary
35
ix
3
DESIGN OF MICROSTRIP ANTENNA
& LEFT-HANDED METAMATERIAL
3.1
Introduction
35
3.2
Flow Chart of the Design Process
36
3.3
Methodology
38
3.4
LHM Design and Configuration
39
3.5
Boundary Condition for the Simulation Setup
42
3.6
Parametric Studies and Analysis of the
Dependence between the Resonant Frequency
and the Parameters of the Unit Cell
3.6.1
Varying the Gaps,G 1 and Width,W 2
of the MSRR
3.6.2
3.7
3.8
and the CLS, G 2
45
3.6.3
Varying the Length of outer MSRR, L 2
46
3.6.4
Varying the Width of CLS, W 1
49
3.6.5
Parametric Studies Conclusion
50
Simulation of the LHM unit cells in Different
Size of Air Gap
50
Microstrip Antenna Design
57
3.8.1
Single Patch Microstrip Antenna
57
3.8.2
Linear Polarized 2x2 Array Patch
3.8.3
61
Circular Polarized 2x2 Array Patch
Microstrip Antenna
4
43
Varying the Gap between the MSRR
Microstrip Antenna
3.9
43
Chapter Summary
65
69
SIMULATION AND MEASUREMENT OF MICROSTRIP
ANTENNA INCORPATED WITH LEFT-HANDED
METAMATERIAL
4.1
Introduction
4.2
Simulation of Single Patch Microstrip Antenna
Incorporated with LHM
70
70
x
4.3
Simulation of Linear Polarized 2 x 2 Array
Patch Microstrip Antenna Incorporated with LHM
4.4
4.5
76
Simulation of Circular Polarized 2 x 2 Array
Patch Microstrip Antenna Incorporated with LHM
78
Measurement Result
83
4.5.1
Measurement of Single Patch Microstrip
Antenna Incorporated with LHM
4.5.2
85
Measurement of Single Linear Polarized
2 x 2 Array Patch Microstrip Antenna
Incorporated with LHM
4.5.3
88
Measurement of Single Circular Polarized
2 x 2 Array Patch Microstrip Antenna
Incorporated with LHM
4.6
5
Chapter Summary
92
96
ANALYSIS AND DISCUSSION
5.1
Introduction
5.2
Analysis and Discussion on Simulation
97
of the Single Patch Microstrip Antenna
Incorporated with LHM
5.2.1
97
E-field analysis in the present of
LHM in front of the single
patch microstrip antenna
5.3
100
Analysis and Discussion on Simulation
of Linear Polarized 2x2 Array Patch
Microstrip Antenna Incorporated with LHM
5.4
102
Analysis and Discussion on Simulation
of Circular Polarized 2x2 Array Patch
Microstrip Antenna Incorporated with LHM
5.5
105
Analysis and Discussion on Measurement
of the Single Patch Microstrip Antenna
Incorporated with LHM
108
xi
5.6
Analysis and Discussion on Measurement
of the Linear Polarized 2x2 Array Patch
Microstrip Antenna Incorporated with LHM
5.7
113
Analysis and Discussion on Measurement
of the Circular Polarized 2x2 Array Patch
5.8
6
Microstrip Antenna Incorporated with LHM
117
Chapter Summary
124
CONCLUSION
6.1
Overall Conclusion
126
6.2
Key Contribution
128
6.3
Future Work
129
REFERENCES
Appendices A - E
130
136 - 146
xii
LIST OF TABLES
TABLE NO.
TITLE
PAGE
2.1
Comparison between five LHM designs
25
2.2
Comparison between three discussed papers
35
3.1
Comparison between different MSRR
40
3.2
Dimension of LHM
42
3.3
Correlation between frequency range of negative
45
permittivity, ε r and permeability, µ r with gap, G 2
3.4
Correlation between gap, G 2 and length, L 1
46
3.5
Correlation between gap, G 2 and length, L 4
46
3.6
Correlation between frequency range of negative
47
permittivity, ε r and permeability, µ r with length, L 2
3.7
Correlation between length, L 2 and length, L 3
48
3.8
Correlation between length, L 2 and length, L 1
48
3.9
Correlation between length, L 2 and length, L 4
48
xiii
3.10
Correlation between frequency range of negative
49
permittivity, ε r and permeability, µ r with width, W 1
3.11
Comparison between 4 mm, 6 mm and 8 mm air gap
55
3.12
Comparison between unit cell with and without air gap
56
4.1
The results of the measured single patch microstrip
86
antenna
4.2
The results of the measured single patch microstrip
87
antenna incorporated with LHM
4.3
The results of the measured linear polarized 2x2 array
89
patch microstrip antenna
4.4
The results of the measured linear polarized 2x2 array
90
patch microstrip antenna incorporated with LHM
4.5
The results of the measured circular polarized 2x2 array
92
patch microstrip antenna
4.6
The results of the measured circular polarized 2x2 array
93
patch microstrip antenna incorporated with LHM
4.7
The results of the measured circular polarized 2x2 array
94
patch microstrip antenna in cross polar position
4.8
The results of the measured circular polarized 2x2 array
patch microstrip antenna incorporated with LHM in
cross polar position
94
xiv
5.1
Comparison of the antenna’s performance between
100
single patch microstrip antenna with and without LHM
5.2
Comparison of the antenna’s performance between
105
linear polarized 2x2 array patch microstrip antenna with
and without LHM
5.3
Comparison of the antenna’s performance between
108
circular polarized 2x2 array patch microstrip antenna
with and without LHM
5.4
Comparison of the antenna’s performance between
111
single patch microstrip antenna with and without LHM
5.5
Comparison between simulated and measured single
112
patch microstrip antenna incorporated with LHM
5.6
Comparison of the antenna’s performance between
116
linear polarized 2x2 array patch microstrip antenna with
and without LHM
5.7
Comparison between simulated and measured linear
polarized
2x2
array
patch
microstrip
116
antenna
incorporated with LHM
5.8
Comparison of the antenna’s performance between
122
circular polarized 2x2 array patch microstrip antenna
with and without LHM
5.9
Comparison between simulated and measured circular
polarized
2x2
array
incorporated with LHM
patch
microstrip
antenna
123
xv
LIST OF FIGURES
FIGURE NO.
TITLE
PAGE
1.1
Different type of material
2
2.1
First experimental LHM structure
7
2.2
(a) Circular split ring resonator and (b) Square split ring
8
resonator.
2.3
(a) Capacitance loaded strip (CLS) and (b) Thin wire
9
(TW)
2.4
(a) TW (solid) and (b) CLS (dotted)
10
2.5
(a) The refracted wave in a RH Medium and (b) The
11
refracted wave in a LH Medium
2.6
The negative refractive index obtained from experiment
12
2.7
The refocused wave after passing through the LHM Slab
12
2.8
(a) RH medium triad and (b) LH medium triad
15
2.9
Split ring resonator (SRR) and a single thin wire (TW)
20
2.10
(a) Value of permittivity and (b) Value of permeability
20
xvi
2.11
Symmetrical ring structure
21
2.12
(a) Value of permittivity and (b) Value of permeability
21
2.13
Omega shape structure
22
2.14
(a) Value of permittivity and (b) Value of permeability
22
2.15
S-Shape Structure
23
2.16
(a) Value of permittivity and (b) Value of permeability
23
2.17
Split ring resonator (SRR) and four capacitance loaded
24
strip (CLS)
2.18
Value of permittivity and permeability
24
2.19
(a)Unit cell of the LHM consisting of SRR and electrical
26
wires and (b) Dimensions of the SRR.
2.20
LH MTM incorporated with the circular patch antenna
26
2.21
Value of permittivity and permeability
26
2.22
(a) S 11 of the patch antenna and (b) S 11 of the patch
27
antenna incorporated with LHM
2.23
(a) Radiation pattern of the patch antenna and (b)
Radiation pattern of the patch antenna incorporated with
LHM
28
xvii
2.24
Antenna incorporated with metamaterials (a) perspective
29
view, (b) side view and (c) dimension of the unit cell
2.25
Measured return loss between ordinary patch antenna and
30
the antenna incorporated with metamaterials (proposed
antenna)
2.26
Measured (solid line) and simulated (dotted line) peak
30
antenna gain across WiMAX 3.5 GHz band
2.27
Measured radiation patterns for E-plane (y-z plane) and
31
H-plane (x-z plane)
2.28
DNG structure consist of SRR and SW
31
2.29
S 11 and S 21 results
32
2.30
Value of permittivity and permeability
32
2.31
Rectangular waveguide antenna incorporated with DNG
33
structure
2.32
(a) Radiation pattern in E-plane and (b) Radiation pattern
33
in H-plane
2.33
Fabricated rectangular waveguide antenna incorporated
34
with DNG structure
2.34
(a) Radiation pattern in E-plane and (b) Radiation pattern
34
in H-plane
3.1
Flow Chart of Designing LHM
37
xviii
3.2
(a) Side view of the LHM (b) Top view of the LHM and
39
(c) Perspective view of the LHM proposed by
Ziolkowski [10]
3.3
Proposed LHM structure
40
3.4
The dimension of the LHM structure
41
3.5
Boundary condition for simulation setup
42
3.6
Results of S 11 and S 21 of the LHM unit cell
44
3.7
Value of ε r and µ r
44
3.8
Correlation between gap, G 2 and resonant frequency
45
3.9
Correlation between length, L 2 and resonant frequency
47
3.10
Correlation between width, W 1 and resonant frequency
49
3.11
Simulation on single cell without air gap
51
3.12
Value of S 11 and S 21 of the single cell without air gap
51
3.13
Value of ε r and µ r of the single cell without air gap
52
3.14
Simulation on single cell with air gap
52
3.15
Value of S 11 of the single cell with different size of air
53
gap
3.16
Value of S 21 of the single cell with different size of air
gap
53
xix
3.17
Value of ε r of the single cell with different size of air gap
54
3.18
Value of µ r of the single cell with different size of air gap
54
3.19
Value of ε r , µ r and refractive index for unit cell with 8
56
mm air gap
3.20
Layout of single patch microstrip antenna
58
3.21
Coaxial port coordinate
59
3.22
Return loss, S 11 of the single patch microstrip antenna
59
3.23
3D radiation pattern at 2.4 GHz
60
3.24
(a) Polar plot of radiation pattern at 2.4 GHz in H-plane
61
and (a) Polar plot of radiation pattern at 2.4 GHz in Eplane
3.25
Layout of the transmission line feeding technique
62
3.26
Layout of linear polarized 2x2 array microstrip patch
63
antenna
3.27
Return loss, S 11 of the 2x2 array patch microstrip antenna
63
3.28
3D radiation pattern at 2.4 GHz
64
3.29
(a) Polar plot of radiation pattern at 2.4 GHz in E-plane
65
and (b) Polar plot of radiation pattern at 2.4 GHz in Hplane
xx
3.30
Layout of Circular Polarized 2x2 Array Patch Microstrip
66
Antenna
3.31
(a) E-field at 00, (b) E-field at 900, (c) E-field at 1800 and
67
(d) E-field at 3600
3.32
Return loss, S 11 of the 2x2 array patch circular polarized
67
microstrip antenna
3.33
3D radiation pattern at 2.4 GHz
67
3.34
(a) Polar plot of radiation pattern at 2.4 GHz in E-plane
68
and (b) Polar plot of radiation pattern at 2.4 GHz in Hplane
4.1
(a) Dimension of the microstrip antenna incorporated
71
with LHM with three different views where (a) (b) front
view, (c) side view and (d) back view
4.2
Return loss, S 11
72
4.3
(a) Simulated radiation patterns at 2.45 GHz, (b)
73
Simulated radiation patterns at 2.48 GHz and (c)
Simulated radiation patterns at 2.4 GHz
4.4
(a) Polar plot of the radiation pattern in H-plane at 2.4
74
GHz and (a) Polar plot of the radiation pattern in E-plane
at 2.4 GHz
4.5
LHM structure
74
4.6
(a) Focusing effect of 12.5 mm gap and (b) Focusing
75
effect of larger than 14.55 mm gap
xxi
4.7
Resonant frequency shift as the gap between the antenna
75
and the LHM structure varies
4.8
Perspective view on the 2x2 array patch microstrip
76
antenna incorporated with LHM
4.9
Return loss, S 11
77
4.10
(a) Simulated radiation patterns at 2.4 GHz and (b)
77
Simulated radiation patterns at 2.45 GHz
4.11
(a) Polar plot of the radiation pattern in H-plane at 2.4
78
GHz and (b) Polar plot of the radiation pattern in E-plane
at 2.4 GHz
4.12
Perspective view of the 2x2 array circular patch
79
microstrip antenna
4.13
Return loss, S 11
80
4.14
Return loss altered as the gap between the antenna and
80
the LHM varies
4.15
(a) Simulated radiation patterns at 2.35 GHz, (b)
82
Simulated radiation patterns at 2.4 GHz and (c)
Simulated radiation patterns at 2.45 GHz
4.16
(a) Polar plot of the radiation pattern in H-plane at 2.4
82
GHz and (b) Polar plot of the radiation pattern in E-plane
at 2.4 GHz
4.17
(a) Measurement setup using network analyzer and (b)
Layout of the measurement setup
84
xxii
4.18
(a) Measurement equipment for radiation pattern
84
measurement (from left; signal generator, spectrum
analyzer, antenna measurement rotator and rotator within
an anechoic chamber) and (b) Layout of the measurement
setup
4.19
(a) The fabricated single patch microstrip antenna and (b)
85
the fabricated LHM
4.20
Perspective view of the single patch microstrip antenna
86
incorporated with LHM
4.21
Measured radiation pattern of the single patch microstrip
87
antenna
4.22
Measured radiation pattern of the single patch microstrip
88
antenna with LHM
4.23
Perspective view of the 2x2 array patch microstrip
89
antenna with and without LHM
4.24
Measured radiation pattern of the 2x2 array patch
90
microstrip antenna
4.25
Measured radiation pattern of the 2x2 array patch
91
microstrip antenna with LHM
4.26
Perspective view of the 2x2 array patch circular polarized
92
microstrip antenna with and without LHM
4.27
Measured radiation pattern of the 2x2 array patch circular
polarized microstrip antenna
95
xxiii
4.28
Measured radiation pattern of the 2x2 array patch circular
95
polarized microstrip antenna with LHM
5.1
Return loss, S 11 of the single patch microstrip antenna
98
incorporated with LHM
5.2
Radiation Patterns in E-plane
99
5.3
Radiation Patterns in H-plane
99
5.4
(a) Observation on E-field in E-plane for the single patch
101
microstrip antenna and (b) Observation on E-field in Eplane for the single patch microstrip antenna incorporated
with LHM
5.5
(a) Observation on E-field in H-plane for the single patch
101
microstrip antenna and (b) Observation on E-field in Hplane for the single patch microstrip antenna incorporated
with LHM
5.6
Return loss, S 11 of the linear polarized 2x2 array
102
mircostrip patch antenna
5.7
(a) 3D radiation pattern of the linear polarized 2x2 array
103
patch microstrip antenna incorporated with LHM and (b)
3D radiation pattern of the linear polarized 2x2 array
patch microstrip antenna
5.8
Radiation pattern in E-plane
104
5.9
Radiation pattern in H-plane
104
xxiv
5.10
Return loss, S 11 of the circular polarized 2x2 array patch
106
microstrip antenna
5.11
Radiation pattern in E-plane
106
5.12
Radiation pattern in H-plane
107
5.13
Return loss, S 11 of the single patch microstrip antenna
109
incorporated with and without LHM
5.14
Transmission coefficient, S 21 of the single patch
109
microstrip antenna incorporated with and without LHM
5.15
Radiation pattern in E-plane
110
5.16
Radiation pattern in H-plane
110
5.17
(a) Comparison between simulated and measured
112
radiation patterns in E-plane and (b) Comparison
between simulated and measured radiation patterns in Hplane
5.18
Return loss, S 11 of the linear polarized 2x2 Array Patch
113
Microstrip Antenna Incorporated with and without LHM
5.19
Transmission coefficient, S 21 of the linear polarized 2x2
114
Array Patch Microstrip Antenna Incorporated with and
without LHM
5.20
Radiation pattern in E-plane
115
5.21
Radiation pattern in H-plane
115
xxv
5.22
(a) Comparison between simulated and measured
117
radiation patterns in E-plane and (b) Comparison
between simulated and measured radiation patterns in Hplane
5.23
Return losses, S 11 circular polarized 2x2 array patch
118
microstrip antenna incorporated with and without LHM
5.24
Transmission coefficient, S 21 in co-polar position of the
119
circular polarized 2x2 array patch microstrip antenna
incorporated with and without LHM
5.25
Transmission coefficient, S 21 in cross-polar position of
119
the circular polarized 2x2 array patch microstrip antenna
incorporated with and without LHM
5.26
Co-polar radiation patterns in E-plane
120
5.27
Cross-polar radiation patterns in H-plane
120
5.28
Co-polar radiation patterns in E-plane
121
5.29
Cross-polar radiation patterns in H-plane
121
5.30
(a) Comparison between simulated and measured
124
radiation patterns in E-plane and (b) Comparison
between simulated and measured radiation patterns in Hplane
xxvi
LIST OF SYMBOLS
E
-
Electric Field
H
-
Magnetic Field
D
-
Electric Flux Density
B
-
Magnetic Flux Density
ρ
-
Charge Density
S
-
Poynting Vector
P0
-
Power Flow
ε
-
Permittivity
μ
-
Permeability
εr
-
Relative Permittivity
μr
-
Relative Permeability
n
-
Refractive Index
c
-
Speed of Light
ω
-
Radian Frequency
ωp
-
Plasma Radian Frequency
k
-
Complex wavenumber
f
-
Frequency
λ
-
Wavelength
ξ
-
Damping Coefficient
Z
-
Impedance
β
-
Propagation Constant
σ
-
Conductivity of Metal
η
-
wave Impedance
T
-
Transmission Coefficient
Γ
-
Reflection Coefficient
υp
-
Phase Velocity
xxvii
f pm
-
Magnetic Plasma Frequency
f pe
-
Electric Plasma Frequency
S 11
-
Return Loss
S 21
-
Insertion loss
d
-
Thickness of the slab (LHM)
xxviii
LIST OF ABBREVIATIONS
DNG
-
Double Negative
LHM
-
Left Handed Metamaterial
NRI
-
Negative Refractive Index
NRW
-
Nicolson-Ross-Weir
TW
-
Thin Wire
SRR
-
Split Ring Resonator
MSRR
-
Modified Split Ring Resonator
CLS
-
Capacitance Loaded Strip
FSS
-
Frequency Selective Surface
xxix
LIST OF APPENDICES
APPENDIX
TITLE
PAGE
A
List of Publications
136
B
Example of the Calculation on the modified
138
NRW method using MathCAD
C
Refraction Index
141
D
Scaling the Dimension of the Left Handed
145
Metamaterial Structure
E
Wet Etching Process
146
CHAPTER 1
INTRODUCTION
1.1
Introduction
Recently, there have been frequent study and research by researchers around
the world regarding the left handed metamaterial that contradict with a lot of the
physics law. The unusual characteristics of these materials have yet to be found in
any natural material and are considered as a new material studied in the 21st century.
Left-handed metamaterial (LHM) is an artificial material (periodic metallic
structure) where the permeability and permittivity were simultaneously negative at a
certain range of frequency [1]. Before venturing deeper into this topic, a brief look
into the material terminology would help in understanding this peculiar artificial
material. Figure 1.1 shows the terminology of the materials.
2
Figure 1.1: Different type of material
From the figure, the material terminology is divided into 4 groups. Group I
shows the Double Positive (DPS) material which have positive value of permittivity
and permeability. Almost all existing materials are DPS material and one of the
examples is dielectric. For group II, Epsilon Negative (ENG) material has only
permeability in positive value but the permittivity is negative. On the other hand,
group III represents Miu Negative (MNG) material which is opposite of ENG
material where the value of permittivity is positive and the value of permeability is
negative. Lastly, the group IV shows the Double Negative (DNG) material and also
known as Left Handed Metamaterial (LHM). This material has both permittivity and
permeability in negative value. The works in this thesis focus on group IV material
where the value of the permittivity and permeability are negative.
Left Handed Metamaterial has a few unique and unusual properties due to
negative value permittivity and permeability of the structure itself. Therefore, a fair
amount of explanation has to be presented in order to showcase this newly
discovered material. Their properties are discussed in chapter 2. One of the unique
properties of Left Handed Metamaterial is negative refraction which will produce the
focusing effect. With this property, LHM will be used to focus the radiation of an
antenna thus should increased the gain of the antenna.
3
1.2
Problem Statement
The study of the Left-Handed Metamaterial is carried out due to escalating
interest in this unique material and its immense potential in various applications.
The material with negative value of permittivity and permeability is not found
in nature. By applying the concept, theory and design of the Left-Handed
Metamaterial, the material with negative value of permittivity and permeability can
be created artificially.
Microstrip antenna suffered low gain. A comment technique to overcome this
drawback is using array of patch antenna. However, this technique has drawbacks
which are high feed network losses and produce mutual coupling [2]. Another
method to overcome this disadvantage is by using the Left Handed Metamaterial.
The integration of the Left Handed Metamaterial with the microstrip antenna will
increased the gain of the microstrip antenna. With this feature, the antenna that
integrates with the Left Handed Metamaterial structure can be used for longer range
point to point or point to multipoint wireless communication or to extend wireless
coverage.
1.3
Scope of Research
The main scopes of the research are:
i.
Study the Left-Handed Metamaterial and microstrip antennas.
ii.
Design, simulate and analyze Left-Handed Metamaterial using Computer
Simulation Technology (CST) CAD tools.
iii.
Simulate the Left-Handed Metamaterial incorporate with the single patch,
2x2 array patch and circular polarized 2x2 array patch microstrip antenna.
iv.
Measurement the Left-Handed Metamaterial incorporate with the single
patch, 2x2 array patch and circular polarized 2x2 array patch microstrip
antenna.
4
v.
Analyze and compare the results between simulation and measurement.
vi.
Thesis write up.
1.4
Objective of Research
The main objectives of this research are:
i.
To study, design, simulate and analyze the new structure of Left-Handed
Metamaterial.
ii.
To simulate, fabricate and measure the Left-Handed Metamaterial
incorporated with the single patch microstrip antenna, 2x2 array patch
microstrip antenna and 2x2 array circular polarized patch microstrip
antenna.
iii.
To compare and analyze the behavior and properties of the microstrip
antennas incorporate with Left Handed Metamaterial.
1.5
Organisation of Thesis
Chapter 1 discusses the brief introduction of Left Handed Metamaterial, the
problem statement, the objective and scope of the research as well as the
organization of the thesis.
Chapter 2 explains the basic concepts and theories of the Left Handed
Metamaterial. The unique properties of this material are elaborated thoroughly in this
chapter.
Chapter 3 discusses on the design of the Left Handed Metamaterial using
Computer Simulation Technology (CST) and MathCAD software. For the
completeness of the thesis, the design equation of the single patch microstrip
5
antenna, 2x2 array patch microstrip antenna and 2x2 array patch circular polarized
microstrip antenna are also presented.
The simulation and measurement results of the Left Handed Metamaterial
incorporated with the antenna are discusses in Chapter 4.
In Chapter 5, the results of the antennas with and without Left Handed
Metamaterial are analyzed in terms of return loss, gain, directivity, half power beam
width and others antenna’s parameters. The results are presented in a form of tables
and graphs.
Lastly, Chapter 6 concludes the finding of the project, key contributions and
recommendations for future research.
In addition, the list of references and appendices were listed at the end of this thesis.
CHAPTER 2
LITERATURE REVIEW ON LEFT HANDED METAMATERIAL AND
INCORPORATION WITH MICROSTRIP ANTENNA
2.1
Introduction
The Left-Handed Metamaterial (LHM) has a few unique properties such as
negative refraction and backward wave. In this chapter, the basic theories behind
their unique properties are presented and some applications of LHM toward the
antenna application are discussed.
2.2
Definition & Background of Left-Handed Metamaterial
Electromagnetic metamaterial can be defined as artificial effective
homogenous electromagnetic structures with unusual properties not readily found in
nature [3]. A Left-Handed Metamaterial (LHM) or Double Negative Metamaterial
(DNG) is an electromagnetic metamaterial that exhibit negative permittivity and
permeability. This phenomenon can be characterized by the negative refraction index
and the anti-parallel phase velocity which is also known as backward wave. The
backward wave propagation has been verified in [4] and the negative refraction has
been proven in [5].
7
2.3
History of Left-Handed Metamaterial (LHM)
The initial work on LHM was started by V. G. Veselago from the Lebedjev
Physical Institute in Moscow when he made a theoretical speculation of this artificial
material that exhibit negative permittivity and negative permeability [6]. Veselago
speculation remain silent for 29 years until 1996, J. B. Pendry from Imperial College
London and his co-author form GEC-Marconi published a paper about artificial
metallic construction which exhibit negative permittivity and negative permeability
[7-8].
Following this interesting discovery, in 2001, the first experimental
verification was made by Shelby, Smith and Schultz at the University of California
[9]. The left handed material structure consists of split ring resonator and thin wire
inspired by J. B. Pendry as shown in Figure 2.1.
Figure 2.1: First experimental LHM structure
Since the introduction of LHM twelve years ago, many researchers were
interested in investigating this artificial material and several of them was using LHM
to improve the properties of the microwave devices such as antennas and filters.
Many papers have been published regarding the LHM integrated with antennas and
their properties have been analyzed [10 - 12]. The focusing affect of LHM has made
a low gain antenna becomes directive and have an increment of gain.
8
2.4
Left-Handed Metamaterial Structure
The first LHM structure consists of split ring resonator (SRR) and thin wire
(TW) or capacitance loaded strip (CLS) [13]. The SRR exhibit the negative value of
permeability and the CLS and TW exhibit the negative value of permittivity in a
certain range of frequency [13].
2.4.1
Split Ring Resonator (SRR)
(a)
(b)
Figure 2.2: (a) Circular split ring resonator and (b) Square split ring resonator
A split ring resonator (SRR) as shown in Figure 2.2 is part of the LHM
structure that exhibit negative value of permeability. If the excitation of the magnetic
field is perpendicular to plane of the structure, this will generate the magnetic dipole
moment [13].
The SRR is a highly conductive structure in which the capacitance between
the two rings balances its inductance [14]. The SRR induces high current density
structure which creates a large magnetic moment. This magnetic moment will exhibit
a plasmonic-type of frequency in the form of [3];
2


ω pm


µ (ω ) = µ0 1 −

(
)
ω
ω
j
ξ
−
m 

(2.1)
9
Where ω pm = 2πf pm
f pm = Magnetic plasma frequency
ξ m = Damping coefficient
µ 0 = Permeability in free-space
ω = 2πf
f = frequency of the electromagnetic wave
It is clear that the SRR would yield a negative value of permeability when the
ω < ω pm.
Plasma frequency can be interpreted as the frequency of the collective vibration of
the electron cloud [15]. It also can be called as Plasmon with a frequency f p.
2.4.2
Capacitance Loaded Strip (CLS) and Thin Wire (TW)
(a)
(b)
Figure 2.3: (a) Capacitance loaded strip (CLS) and (b) Thin wire (TW)
Figure 2.3(a) shows the capacitance loaded strip (CLS) and Figure 2.3(b)
shows the thin wire (TW). CLS and TW would produce strong dielectric like
response. As electric field propagates parallel through the TW or CLS, it will induce
a current along them. This will generate an electric dipole moment to the structure
and exhibit a plasmonic-type of permittivity frequency in a function of [3];
2


ω pe


ε (ω ) = ε 0 1 −

(
)
ω
ω
j
ξ
−
e 

(2.2)
10
Where
ω pe = 2πf pe
f pe = Electric plasma frequency
ξ e = Damping coefficient
ε 0 = Permittivity at free-space
ω = 2πf
f = frequency of the electromagnetic wave
It is clear that the SRR would yield a negative value of permittivity when the
ω < ω pe.
The inclusions in the CLS act as an extra capacitance and these criteria will
make the resonant frequency shift to a lower region compared to the resonant
frequency of the TW. Figure 2.4 shows the differential between CLS and TW in term
of transmissions. CLS has a lower stop band than TW due to the extra capacitance
exists in CLS [16].
Figure 2.4: (a) TW (solid) and (b) CLS (dotted) [16]
11
2.5
Left-Handed Metamaterial Characteristic
2.5.1
Negative Refraction
Due to its peculiarity in its DNG values, where the ε and μ are both negative,
many other properties of this material are changed altogether. The most obvious
alteration is the refractive index of the material where it takes on a negative value as
given by the formula [17];
n = ± ε r µr
(2.3)
On the other hand, the Snell’s law shown that the wave that propagates
through the LHM bends the ‘wrong’ way as shown in Figure 2.5.
(a)
(b)
Figure 2.5: (a) The refracted wave in a RH Medium and (b) The refracted wave in a
LH Medium
In Figure 2.5(b), the refractive index of n 2 ’ = - n 2 and the wave is refracted to
the opposite side compared to the ray propagating in Right-Handed (RH) Medium. In
addition, even though the wave bends the opposite direction, the Snell’s Law is still
been satisfied when a negative value of n is substituted into the equation,
n 1 sin θ 1 = n 2 sin θ 2
(2.4)
12
and θ 2 < 0 is obtained. Figure 2.6 shows the experiments done by [18] in order to
prove the negative refraction,
RHM
LHM
Figure 2.6: The negative refractive index obtained from experiment [18]
Due to its negative refractive index, it is apparent that wave propagating
through a LHM slab would be focused internally inside the slab and create a refocus
point after leaving the slab. This is shown in Figure 2.7 and if this idea is applied into
the microwave engineering, an enhancement of the directivity and gain can be
obtained from a particular antenna.
Figure 2.7: The refocused wave after passing through the LHM Slab
13
2.5.2
Backward Wave
To prove one of the unique properties of the LHM, the backward wave
phenomena. It starts with the Maxwell equations [19].
∇× E = −
∇× H =
∂B
−MS
∂t
∂D
+JS
∂t
(2.5)
(2.6)
∇ ⋅ D = ρe
(2.7)
∇ ⋅ B = ρm
(2.8)
Where E = Electric field density
H = Magnetic field density
D = Electric flux density
B = Magnetic flux density
M s = Magnetic current density
J s = Electric current density
ρ e = Electric charge density
ρ m = Magnetic charge density
The Maxwell equations and the constitutive equation can then be presented as:
∇ × E = − jωµ H − M s
∇ × H = jωε E + J s
(2.9)
(2.10)
∇ ⋅ D = ρe
(2.11)
∇ ⋅ B = ρm
(2.12)
14
and
D =εE
(2.13)
B = µH
(2.14)
To make it simple, assume that the lossless medium ( ε " = µ " = 0 ) in region without
sources ( M s = J s = 0 ). In the case of ordinary RH medium ( ε , µ > 0 ), therefore
[13]:
β × E = +ωµ H
(2.15)
β × H = −ωε E
(2.16)
Where β = wave vector
On the other hand, in the case of LHM medium ( ε , µ < 0 ), and since ε = −ε > 0
and µ = − µ > 0 , therefore [13]:
β × E = −ωµ H
(2.17)
β × H = +ωε E
(2.18)
This builds the LHM arrangement as shown in Figure 2.8(b) while Figure 2.8(a)
shows the RH arrangement.
15
β
β
(a)
(b)
Figure 2.8: (a) RH medium triad and (b) LH medium triad
Where S = Poynting vector
β = Wave number
E = Electric field
H = Magnetic field
As can be seen, the wave number, β is anti-parallel with the power flow in the
LH medium.
The phase velocity then,
νp =
Where β =
ω
β
β
(2.19)
β
β
The phase velocities of the RH medium are reverse compared to the LH
medium. Moreover, because the wave number, β is known positive in RH medium
(outward propagation from the source), it is negative in LH medium (inward
propagation from the source). It can be summarized as:
16
RH medium: β > 0
LH medium: β < 0
(ν
(ν
p
> 0)
p
< 0)
As for the poynting vector, the power flows of the LH medium are similar to the RH
medium. It can be defined as:
S = E×H *
P0 =
1
E × H * ⋅d s
2 ∫S
(2.20)
(2.21)
Where S = Poynting vector
P0 = Power flow
On the other hand, the phase velocity is rather opposite to the oriented
direction due to the fact that phase velocity is simply corresponding to the
propagation of perturbation and not of energy [13].
In conclusion, we can see the effect of the backward wave phenomenon due
to the opposite orientation of the phase velocity. Although the phase velocity is
reverse oriented, the power flow and the group velocity is not affected.
2.6
Method to Determine the Value of Permittivity and Permeability Using
Modified Nicolson-Ross-Wier (NRW) Approach [20]
In order to get more accurate approximation of the permittivity and
permeability, the modified NRW Approach were studied and applied in this project
[20]. NRW approach is commonly used technique to determine the value of
permittivity and permeability.
17
The NRW method begins the expression of the transmission term, T from equation
below;
Τ=
V1 − Γ
1 − ΓV1
(2.22)
Where the expression of the reflection coefficient, Γ ;
Γ=
Τ − V2
1 − ΤV2
(2.23)
Where;
V1 = S 21 + S11
(2.24)
V2 = S 21 − S11
(2.25)
From (2.22) and (2.23), we can obtain the equation below;
1− Τ =
η=
(1 + Γ )(1 − V1 )
1 − ΓV1
1 + Γ 1 + Τ 1 − V2
=
1 − Γ 1 − Τ 1 + V2
(2.26)
(2.27)
Where T = Transmission coefficient
Γ = Reflection coefficient
η = Wave impedance
Assuming that the electrical thickness of the LHM slab is not large (i.e., k real d < 1)
and aware that the wave number;
k=
ω ε r µr
c
Where ω = 2πf
c = Speed of light, 2.998x108 m/s
= k0 ε r µr
(2.28)
18
k0 =
ω
c
The transmission term can be written as Τ ≈ 1 − jkd to obtain the approximate
results of permittivity and permeability from equation (2.26) and (2.27), respectively;
µr ≈
2 1 − V2
jk 0 d 1 + V2
 k
ε r = 
 k0
(2.29)
2
 1

 µr
(2.30)
Where d = thickness of substrate
k0 =
ω
c
V2 = S 21 − S11
Where the refraction index is simply obtained as;
n = ε r µr =
k
k0
(2.31)
And the wave impedance as;
µ r 1 + V1 1 − V2 (S11 + 1)2 − S 212
=
=
η =
ε r 1 − V1 1 + V2 (S11 − 1)2 − S 212
2
(2.32)
To avoid the square root issues in the permittivity expression, the equation (2.32) and
(2.29) below has been used to get the equation of permittivity;
εr ≈
2 1 − V1
jk 0 d 1 + V1
(2.33)
19
Where d = thickness of substrate
k0 =
ω
c
V1 = S 21 + S11
MathCAD software is used in the calculation of the permittivity and
permeability of the LHM structure. The NRW Approach (Equation 2.29 and 2.33)
are used to calculate the permittivity and permeability of the LHM. This was done by
exporting the S-Parameters from CST Microwave Studio software to MathCAD
software.
Once the LHM region was obtained, the exact dimensions of the structure
were incorporated with antennas via CST Microwave Studio and for subsequent
fabrication processes.
2.7
Previous Research on of Left-Handed Metamaterial
2.7.1
Split Ring Structure
The LHM consist of a split ring resonator (SRR) and a single thin wire (TW)
as shown in Figure 2.9 [21]. The results are shown in Figure 2.10 where Figure
2.10(a) shows the value of permittivity and the value of permeability is shown in
Figure 2.10(b).
20
Thin wire
Dielectric substrate
Split ring
resonator
Figure 2.9: Split ring resonator (SRR) and a single thin wire (TW)
εr
µr
(a)
(b)
Figure 2.10: (a) Value of permittivity and (b) Value of permeability [21]
Figure 2.10 (a) shows that the negative value of permittivity exists in region
below 11.2 GHz while Figure 2.10(b) shows that the negative value of permeability
exists from 11 GHz to 11.2 GHz. In conclusion, the range of negative permittivity
and permeability is from 11 GHz to 11.2 GHz. The value of imaginary part of
permittivity and permeability is near to zero therefore the structure has small losses.
2.7.2
Symmetrical Ring Structure
Symmetrical ring structure as shown in Figure 2.11 is one of the classic
structure that exhibit negative permittivity and permeability [21]. The value of
permittivity and permeability of the structure are shown in Figure 2.12(a) and
2.12(b).
21
Thin wire
Dielectric substrate
Symmetrical
ring
Figure 2.11: Symmetrical ring structure
εr
µr
(a)
(b)
Figure 2.12: (a) Value of permittivity and (b) Value of permeability [21]
Respectively from observation, the permittivity has a negative value from 8.5
GHz to 12 GHz and below 8 GHz while the negative value of permeability occurs at
around 8 GHz. The range of simultaneously negative permittivity and permeability is
from 8 GHz to 8.2 GHz. At that region, the imaginary part of permittivity and
permeability is near to zero.
2.7.3
Omega Structure
Figure 2.13 shows the Omega shape structure. This structure is a complex
design where the rod and the ring are coupled together which means that the
permittivity and permeability are coupled [21]. Figure 2.14(a) shows the value of
permittivity while Figure 2.14(b) shows the value of permeability.
22
Omega
shape
Dielectric substrate
Figure 2.13: Omega shape structure
εr
µr
(a)
(b)
Figure 2.14: (a) Value of permittivity and (b) Value of permeability [21]
The result shows that the negative permittivity and permeability occurs from 11 GHz
to 15 GHz. The value of imaginary part of permittivity and permeability is near to
zero start from 12.5 GHz above.
2.7.4
S-Shape Structure
Figure 2.15 shows another LHM consist of a coupled “S” shaped structure.
There are no obvious rings or rod parts any more, but it still has the properties of
having an electric plasma frequency and a magnetic resonant frequency [21]. Figure
2.16(a) shows the value of permittivity while Figure 2.16(b) shows the value of
permeability.
23
Dielectric substrate
S shape
Figure 2.15: S-Shape Structure
εr
µr
(a)
(b)
Figure 2.16: (a) Value of permittivity and (b) Value of permeability [21]
Two negative permittivity regions exist from 6.5 GHz to 9.4 GHz and 10 GHz to 19
GHz. The result shows that the negative permeability region occurs from 10 GHz to
12.5 GHz. The negative permittivity and permeability region are from 10 GHz to
12.5 GHz.
2.7.5
Split Ring with Capacitance Loaded Strip Structure
The structure consists of a split ring resonator (SRR) and four capacitance
loaded strip (CLS) as shown in Figure 2.17. Figure 2.18 shows the value of
permittivity and the permeability of the structure.
24
Dielectric substrate
Capacitance
loaded strip
Split ring
resonator
Figure 2.17: Split ring resonator (SRR) and four capacitance loaded strip (CLS)
εr
µr
Figure 2.18: Value of permittivity and permeability [20]
The result shows that the negative permeability region occurs from 9.5 GHz to 9.9
GHz while the negative permittivity region occurs from 6.3 GHz to 9.3 GHz and
9.55 GHz to 15 GHz. The region with negative permittivity and permeability is from
9.55 GHz to 9.9 GHz.
2.7.6
Summary
Five LHM designs have been discussed. Table 2.1 shows the comparison of
those five designs. S-shape structure and Omega structure have wider range of
negative permittivity and permeability than the others. Split ring structure and
25
Symmetrical structure have the smallest bandwidth of negative permittivity and
permeability which is around 0.2 GHz. Split ring with Capacitance loaded strip has a
bandwidth of negative permittivity and permeability up to 0.35 GHz.
Table 2.1: Comparison between five LHM designs
Split Ring
LHM /
Split Ring
Symmetrical
Omega
S-Shape
with
parameters
Structure
Ring
Structure
Structure
Capacitance
Structure
Loaded Strip
Structure
Range of
11 GHz to
8 GHz to
11 GHz to
10 GHz to
9.55 GHz to
negative
11.2 GHz
8.2 GHz
15 GHz
12.5 GHz
9.9 GHz
0.2 GHz
0.2 GHz
4 GHz
2.5 GHz
0.35 GHz
µr & εr
Bandwidth
of
negative
µr & εr
2.8
LHM Application in Antenna Technology
2.8.1
Theoretical Investigation of a Circular Patch Antenna in the presence of
a Left-Handed Medium [22]
This paper describes the influence of a LHM on the performances of a
circular patch antenna. The aim of this paper is to show that the performances of a
patch antenna can be increased by incorporating the LHM without the use of an array
of antennas. Roger RT/duroid 5880 with a permittivity of 2.2, thickness of 0.254
mm and a tangential loss of 0.0009 has been used to construct the patch antenna and
LHM. The construction of the LH MTM is shown in Figure 2.19 which represents
the unit cell of LH MTM and the patch antenna incorporated with the LHM is
depicted in Figure 2.20.
26
(a)
(b)
Figure 2.19: (a)Unit cell of the LHM consisting of SRR and electrical wires and (b)
Dimensions of the SRR.
Figure 2.20: LH MTM incorporated with the circular patch antenna
The S-parameters of the LHM that have been obtained from the numerical
calculation by using the finite element-based computation in Ansoft’s HFSS and the
data was then been extracted to get the permittivity and permeability. The extracted
data are shown in Figure 2.21.
Figure 2.21: value of permittivity and permeability
27
From Figure 2.21, the interesting area in the range of the negative refraction
index is lies from 12.25 GHz to 13.3 GHz. Notes that at 12.4 GHz, both permittivity
and permeability are negative and the refraction index is -1.55. The grey area shows
in Figure 2.21 represent the frequency band where Re(n) saturated due to the positive
value of |Re(n)| attained.
The circular patch antenna was then been designed at 12.4 GHz with the
patch radius of 5.28 mm. The dielectric board that was used is made of foam having
a relative permittivity of 1.45 and tangential loss of 0.0058. The results of S 11 for the
patch antenna is shown in Figure 2.22(a) while Figure 2.22(b) shows the results of
S 11 of the patch antenna incorporated with the LHM. With the integration of LHM,
the S 11 obtained was slightly different due to the frequency-varying effective
parameters of the LHM.
(a)
(b)
Figure 2.22: (a) S 11 of the patch antenna and (b) S 11 of the patch antenna
incorporated with LHM
The radiation pattern of the antenna with and without the LHM is measured.
Figure 2.23(a) shows the radiation pattern of the patch antenna in E-plane while
Figure 2.23(b) shows the radiation pattern of the patch antenna incorporated with
LHM in E-plane. From observation, it was seen that the maximum power received
for the E-plane increased approximately 2 dB respectively.
28
(a)
(b)
Figure 2.23: (a) Radiation pattern of the patch antenna and (b) Radiation pattern of
the patch antenna incorporated with LHM
As a conclusion, an improvement of gain up to 2 dB was achieved and a more
directional antenna was obtained when the LHM was placed on top of the antenna. A
more directional antenna was obtained even without the use of an array of several
elements.
2.8.2
Metamaterial Enhanced Patch Antenna for WiMAX Application [23]
This paper shows an air-layered patch antenna incorporated with
metamaterial super-strate structure. According to the measured result, the
metamaterial antenna radome can improve the antenna gain to about 2.0 dB. Figure
2.24 below shows the antenna incorporated with the metamaterials and the
dimension of the unit cell itself.
29
Figure 2.24: Antenna incorporated with metamaterials (a) perspective view, (b) side
view and (c) dimension of the unit cell
The size of the substrate is 110 mm x 55 mm covering the patch antenna
while the ground plane of the antenna is 100 mm x 85 mm. The gap between
metamaterials to the antenna is impedance sensitive and has been tune to a height of
3.5 mm from the ground plane. The substrate used was FR4 board with a dielectric
constant of 4.4. The metamaterials consist of 4 layered S-shape metallic rings
sandwiched to each other where the dimension has been tuned to operate at the
WiMAX 3.5 GHz band.
The antenna has been fabricated and measured. The return loss and the
antenna’s gain vary across 3.4 GHz to 3.6 GHz band and the results are shown in
Figure 2.25 and Figure 2.26. The bandwidth of the antenna incorporated with
metamaterials improved to 520 MHz which is 14.2 %. In other hand, the measured
antenna’s gain has been effectively improved around 1.9 – 2.6 dB in the whole
operating band while 1.6 – 3.0 dB in simulated results.
30
Figure 2.25: Measured return loss between ordinary patch antenna and the antenna
incorporated with metamaterials (proposed antenna)
Figure 2.26: Measured (solid line) and simulated (dotted line) peak antenna gain
across WiMAX 3.5 GHz band
Figure 2.27 shows the radiation pattern of the antenna incorporated with
metamaterials has been studied at 3.6 GHz. The 3 dB beam-width in E-plane is 510
and the H-plane is 370. From the measured E-plane and H-plane, a decent broadside
radiation pattern is observed and it is similar to the ordinary patch antenna.
31
Figure 2.27: Measured radiation patterns for E-plane (y-z plane) and H-plane (x-z
plane)
2.8.3
A Study of Using the Double Negative Structure to Enhance the Gain of
Rectangular Waveguide Antenna Array [24]
This paper describes a method to enhance the gain of an array of rectangular
waveguide antenna using double negative medium (DNG). The DNG structure
consists of split ring resonator (SRR) and strip wire (SW) as shown in Figure 2.28.
SRR
SW
Dielectric
substrate
Figure 2.28: DNG structure consist of SRR and SW
The structure uses the strip wire to produce the negative permittivity while
the SRR which consists of two concentric square rings is producing the negative
32
permeability. The value of permittivity and permeability is negative when the
frequency of the propagating electromagnetic wave is below the plasma frequency.
The DNG structure is placed inside the rectangular waveguide. The
waveguide is filled with a dielectric substrate. The dielectric constant of the substrate
is 2.65. The S-parameters data is retrieved and the value of permittivity and
permeability can be obtained using modified Nicolson-Ross-Wier (NRW) approach.
Figure 2.29 shows the S 11 and S 21 results while Figure 2.30 shows the value of
permittivity and permeability. As can be seen, it resonates at 12 GHz and the value of
permittivity and permeability are simultaneously negative at that region.
Figure 2.29: S 11 and S 21 results
Figure 2.30: Value of permittivity and permeability
A single rectangular waveguide antenna (BJ-100) is used and the size is 22.86
mm x 10.16 mm. The frequency chosen is 12 GHz with a 35 mm x 35 mm ground
plane. The dielectric board has a dielectric constant of 2.65 and 0.5 mm thickness.
33
The DNG structure is placed 15 mm from the antenna aperture. Figure 2.31 shows
the antenna incorporated with the DNG structure.
Figure 2.31: Rectangular waveguide antenna incorporated with DNG structure
(a)
(b)
Figure 2.32: (a) Radiation pattern in E-plane and (b) Radiation pattern in H-plane
Figure 2.32(a) and 2.32(b) shows the radiation pattern of the antenna with
and without DNG structure in E-plane and H-plane. The gain of the antenna
increased after the introduction of the DNG structure. The gain enhance from 6.86
dB to 11.68 dB while the back lobe of the antenna is reduced. The gain of the
antenna increases up to 4.82 dB.
The improvement of the gain of the antenna is due to DNG structure can
congregate the radiation energy. The rectangular waveguide antenna with the DNG
structure is fabricated and measured in order to prove the validity of the simulation
results. Figure 2.33 shows the fabricated rectangular waveguide antenna incorporated
34
with DNG structure and the results of the measured radiation pattern in E-plane and
H-plane are shown in Figure 2.34(a) and 2.34(b).
Figure 2.33: Fabricated rectangular waveguide antenna incorporated with DNG
structure
Figure 2.34: (a) Radiation pattern in E-plane and (b) Radiation pattern in H-plane
From the results, the gain of the antenna improved up to 4 dB and the
magnitude of the back lobe is reduced. The experimental results are consistent to the
simulation results and this shows that this method is effective to improve the gain of
the antenna.
2.8.4
Summary
Three papers have been discussed in this sub-chapter. The incorporation of
the LHM improves the performance of the antenna. The main improvement of the
35
antenna’s performance is the gain. Table 2.2 shows the performance of the antenna
incorporated with LHM in those three papers.
Table 2.2: Comparison between three discussed papers
Paper title
Theoretical
A Study of Using the
Investigation of a
Metamaterial
Double Negative
Circular Patch
Enhanced Patch
Structure to Enhance
Antenna in the
Antenna for WiMAX
the Gain of
presence of a Left-
Application [18]
Rectangular
Handed Medium [17]
Waveguide Antenna
Array [19]
12.25 GHz
3.5 dB
12 GHz
2.8 dB
2 dB
4 dB
Half power
E-plane: 450
E-plane: 510
E-plane: 400
beamwidth
H-plane:450
H-plane: 370
H-plane: 400
Resonant
frequency
Gain
increment
2.9
Chapter Summary
The Definition and background of the LHM has been presented in this
chapter. The first LHM consist of split ring resonator (SRR) and thin wire (TW)
which product negative permeability and negative permittivity has been discussed.
The unique properties of the LHM such as negative refraction and backward wave
have been elaborated thoroughly and the method and approach to obtain the value of
permittivity and permeability has been presented. A few recently publish papers on
the LHM toward the antenna application is also discussed. The incorporation of the
LHM increases the antenna’s gain.
CHAPTER 3
DESIGN OF LEFT-HANDED METAMATERIAL &
MICROSTRIP ANTENNAS
3.1
Introduction
In Chapter 2, the mystery of the LHM has been revealed and the method to
determine the value of permittivity and permeability has been discussed. In this
chapter, the design of the LHM is discussed and the procedure in the simulation of
the LHM using CST software is elaborated thoroughly. Besides that, the design of
the single patch, linear polarized 2x2 array patch and circular polarized 2x2 array
patch microstrip antennas are also elaborated.
36
3.2
Flow Chart of the Design Process
Start
Design the LHM structure using
CST Microwave Studio
Obtain the SParameters
Export the S-Parameters
to MathCAD
Calculate Effective Negative
Parameters using NRW Approach
No
Simultaneous
Negative ε & µ
Achieved?
Yes
Design Single Patch, Linear
polarized 2x2 Array Patch &
Circular Polarirzed 2x2 Array Patch
Microstrip Antennas Operating at
2.4 GHz
2
1
37
1
2
Obtain Return Loss,
Radiation Pattern and Other
Parameters of the Antennas
Integrate LHM to
Antennas
Obtain Return Loss,
Radiation Pattern and Other
Parameters of the Antennas
integrate with LHM
Compare the
Gain/Directivity and
Return Loss
Analysis & Discussion
No
Gain/Directivity,
Improved?
Return Loss
acceptable?
Yes
Compile & write report
End
Figure 3.1: Flow Chart of Designing LHM
38
3.3
Methodology
The main area of works and research approach include:
i.
Study, design and simulate the Left Handed Metamaterial structure and
incorporated with microstrip antenna.
•
Study and understand the techniques of designing the Left Handed
Metamaterial structure and the microstrip antenna.
•
Investigate and categorize the Left Handed Metamaterial structure by
looking into their various designs.
•
Design the Left Handed Metamaterial structure and incorporated with
microstrip antenna
•
ii.
Simulation using Computer Simulation Technology (CST) software.
Fabricate the Left Handed Metamaterial structure and microstrip antenna.
•
Develop and construct models using inexpensive printed circuit board
construction techniques. The process involved layout fabrication and
photo etching.
iii.
Measure and test the structure
•
Analyze, evaluate and compare the results (S 11 , S 21 and radiation pattern)
from simulations and fabrication.
iv.
Report
•
Finalize the designs and compile thesis.
39
3.4
LHM Design and Configuration
The designed LHM is inspired by the LHM designed by Ziolkowski [20].
Figure 3.2(a), 3.2(b) and 3.2(c) shows the initial design of the LHM unit cells
proposed by Ziolkowski and the detail of the structure have been elaborated in
Chapter 2. As can be seen, the LHM structure has four CLSs and a single SRR and it
is placed in a substrate with a dielectric constant of 2.2. The dimension of LHM unit
cell structure is 4.318 mm high, 2.3622 mm wide and 7.366 mm long.
(a)
(b)
(c)
Figure 3.2: (a) side view of the LHM (b) top view of the LHM and (c) perspective
view of the LHM proposed by Ziolkowski [10]
In order to use less substrate and reduce the cost of the project, a planar
structure is proposed in the designed LHM. The designed of LHM structure is shown
in Figure 3.3 and it consist of two CLSs and a SRR.
40
Figure 3.3: Proposed LHM structure
A few modifications on the SRR have been done in order to analyze the
effect of permittivity and permeability. Table 3.1 shows the proposed LHM structure
where the SRR of the structure is modified. The first LHM structure is the initial
structure design by Ziolkowski while the second LHM structure is modified in such a
way that the SRR has four gaps and the gaps are placed at the center of the structure.
The SRR of the third LHM structure also has four gaps and the gaps are placed
perpendicular to the structure while the SRR of the fourth LHM structure has eight
gaps.
Table 3.1: Comparison between different MSRR
No
LHM structure
Value of permittivity, ε r and
permeability, µ r
1
150
100
50
0
-50
-100
-150
0
1
2
3
4
5
6
7
8
Frequency, GHz
Permittivity
Permeability
2
60
40
20
0
-20
-40
-60
0
1
2
3
4
5
Frequency, GHz
Permittivity
Permeability
6
7
8
41
3
60
40
20
0
-20
-40
-60
-80
0
1
2
3
4
5
6
7
8
6
7
8
Frequency, GHz
Permittivity
Permeability
4
100
80
60
40
20
0
-20
-40
-60
-80
0
1
2
3
4
5
Frequency, GHz
Permittivity
Permeability
The first structure was structure designed by Ziolkowski. In order to produce
new design structure, the initial design will be neglected from the analysis. The
second proposed LHM structure has a wide band of negative ε r and µ r from around
4.08 GHz to 4.67 GHz and the third proposed LHM structure also has a wide band of
negative ε r and negative µ r from 3.50 GHz to 4.02 GHz. As can be seen, the fourth
LHM structure has almost non-existing band of negative ε r and negative µ r .
Subsequently, the only structures that can be chosen are the second and third
proposed structure. Although the second proposed structure is operating at higher
frequency than the third proposed structure, it has been chosen for further analysis
because the second proposed structure has a wider frequency range of negative ε r and
µ r compared to the third proposed structure.
The initial dimension of the LHM unit cell is shown in Figure 3.4. It consists
of one SRR between two pairs of CLSs in planar form. Table 3.2 shows the
dimensions of the LHM. The dielectric constant of the substrate is 4.7 with a
thickness of 1.6 mm and a tangential loss of 0.019.
42
Figure 3.4: The dimension of the LHM structure
Table 3.2: Dimension of LHM
3.5
Parameters
Dimension (mm)
W1
1
W2
0.5
G1
0.5
G2
2
G3
1
L1
15.1
L2
9.1
L3
7.1
L4
13.1
L5
6.55
Boundary Condition for the Simulation Setup
The simulation of LHM has been done using Computer Simulation
Technology (CST) software. Perfect magnetic conductor (PMC) boundary condition
is set on the front and back faces of the block in z-axis and perfect electric conductor
(PEC) boundary condition is set on the top and bottom of the block in the y-axis. The
43
E-field of the incident wave is polarized along y-axis while the H-field of the
incident wave is polarized along z-axis and the wave propagates in x-axis direction.
Figure 3.5 illustrates the simulated structure. Those boundaries are configured in
such a way so that zero reflection occurs in the waveguide. The boundary setting for
simulation of LHM has been used in papers such as [20], [21], [24] and [25].
Figure 3.5: Boundary condition for simulation setup
Through this configuration, the S-Parameters (S 11 and S 21 ) data were
collected and exported to MathCAD for the calculation of the LHM region using the
modified NRW Approach.
3.6
Parametric Studies and Analysis of the Dependence between the
Resonant Frequency and the Parameters of the Unit Cell
In this section, the second proposed LHM structure is chosen and the
parameters such as the gaps, lengths and widths of the unit cell are varied in order to
study the influence in the determination of the resonant frequency and the value of ε r
and µ r .
3.6.1
Varying the Gaps,G 1 and Width,W 2 of the MSRR
The gaps, G 1 and width, W 2 of the MSRR are varied from 0.5 to 1.25 mm for
both gap and width respectively.
44
0
S11 & S21, dB
-10
-20
-30
-40
-50
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
Frequency, GHz
S11 at
S21 at
S11 at
S21 at
S11 at
S21 at
S11 at
S21 at
0.5 mm
0.5 mm
0.75 mm
0.75 mm
1.0 mm
1.0 mm
1.25 mm
1.25 mm
Figure 3.6: Results of S 11 and S 21 of the LHM unit cell
40
30
20
10
εr, µr
0
-10
-20
-30
-40
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
Frequency, GHz
Permittivity at 0.5 mm
Permeability at 0.5 mm
Permittivity at 0.75 mm
Permeability at 0.75 mm
Permittivity at 1.0 mm
Permeability at 1.0 mm
Permittivity at 1.25 mm
Permeability at 1.25 mm
Figure 3.7: Value of ε r and µ r
It was observed that the gap, G 1 and width, W 2 did not have any significant
to the resonant frequency and the value of ε r and µ r as shown in Figure 3.6 and
Figure 3.7. In this case, only both parameters have been varied and the others were
remained with the same dimensions that were fixed in the original structure.
45
3.6.2
Varying the Gap between the MSRR and the CLS, G 2
In this case, the gap between the MSRR and the CLS, G 2 is varied to observe
the effect of the resonant frequency and the value of ε r and µ r . The dimension of the
MSRR is fixed as an initial structure shown in Figure 3.4. The results are plotted and
shown in Figure 3.8 and Table 3.3.
4.2
4.0
3.8
Frequency, GHz
3.6
3.4
3.2
3.0
2.8
2.6
2.4
2.2
2.0
1
2
3
4
5
6
7
8
9
Gap, mm
Resonant frequency vs Gap
Figure 3.8: Correlation between gap, G 2 and resonant frequency
Table 3.3: Correlation between frequency range of negative permittivity, ε r and
negative permeability, µ r with gap, G 2
Gap (G 2 )
Frequency range of negative permittivity &
permeability (GHz)
2mm
4.0 - 4.224
4mm
3.104 - 3.264
6mm
2.592 - 2.736
8mm
2.224 - 2.336
Referring to Figure 3.8, the resonant frequency is shifted by varying the gap,
G 2 . When the gap, G 2 increase, the resonant frequency becomes lower while the
range of negative ε r and negative µ r are shifted to the lower region as shown in Table
3.2. Extra capacitance produces when the gap, G 2 increase and shifted the resonant
frequency to lower region. Note that, by varying the gap between the SRR and CLS,
G 2 the CLS inclusion, L 1 and CLS length, L 4 also varies. Table 3.4 shows the
46
correlation between the gap, G 2 and CLS inclusion, L 1 . Meanwhile, Table 3.5 shows
the correlation between the gap, G 2 and the CLS length, L 4 . The increment of the
length, L 1 and L 4 produce extra inductance, therefore shifted the resonant frequency
even lower.
Table 3.4: Correlation between gap, G 2 and length, L 1
Gap (G 2 )
CLS inclusion length (L 1 )
2 mm
15.1 mm
4 mm
19.1 mm
6 mm
23.1 mm
8 mm
27.1 mm
Table 3.5: Correlation between gap, G 2 and length, L 4
3.6.3
Gap (G 2 )
CLS strip length (L 4 )
2 mm
13.1 mm
4 mm
17.1 mm
6 mm
21.1 mm
8 mm
25.1 mm
Varying the Length of outer MSRR, L 2
In this simulation, the length of the outer MSRR, L 2 is varied to observe the
effect of the resonant frequency and the value of ε r and µ r . As the L 2 varied, other
parameters such L 1 , L 3 and L 4 are altered as those parameters are related to L 2 .
47
4.8
4.6
Frequency, GHz
4.4
4.2
4.0
3.8
3.6
3.4
3.2
6
7
8
9
10
11
12
13
Length, mm
Resonant frequency vs Length
Figure 3.9: Correlation between length, L 2 and resonant frequency
Table 3.6: Correlation between frequency range of negative permittivity, ε r and
negative permeability, µ r with length, L 2
MSRR outer length (L 2 )
Frequency range of negative permittivity
& permeability (GHz)
7.1 mm
4.762 - 4.936
8.1 mm
4.426 - 4.552
9.1 mm
4.084 - 4.264
10.1 mm
3.808 - 4.036
11.1 mm
3.568 - 3.808
12.1 mm
3.358 - 3.736
Figure 3.9 shows the correlation between L 2 and the resonant frequency. As
the length of the outer MSRR increase, the resonant frequency goes to the lower
region. Consequently, the range of negative ε r and µ r also goes to the lower
frequency region as the value of L 2 increase as shown in Table 3.6. Extra inductance
produces when the length, L 2 increase and shifted the resonant frequency to lower
region. As a result on varying the value of L 2 , other parameters also are altered.
Table 3.7 shows the correlation between the MSRR outer length, L 2 with the MSRR
inner length, L 3 . Table 3.8 shows the correlation between MSRR outer length, L 2
with the CLS inclusion length, L 1 . While, Table 3.9 shows the relationship between
the MSRR outer length, L 2 with the CLS strip length, L 4 . As can be seen from these
48
three tables, the value of L 3 , L 1 and L 4 increases as the MSRR outer length, L 2
increase. This shows that increasing the size of the structure will make the resonant
frequency and the range of negative ε r and µ r shift to the lower region.
Table 3.7: Correlation between length, L 2 and length, L 3
MSRR outer length (L 2 )
MSRR inner length (L 3 )
7.1 mm
5.1 mm
8.1 mm
6.1 mm
9.1 mm
7.1 mm
10.1 mm
8.1 mm
11.1 mm
9.1 mm
12.1 mm
10.1 mm
Table 3.8: Correlation between length, L 2 and length, L 1
MSRR outer length (L 2 )
CLS inclusion length (L 1 )
7.1 mm
13.1 mm
8.1 mm
14.1 mm
9.1 mm
15.1 mm
10.1 mm
16.1 mm
11.1 mm
17.1 mm
12.1 mm
18.1 mm
Table 3.9: Correlation between length, L 2 and length, L 4
MSRR outer length (L 2 )
CLS strip length (L 4 )
7.1 mm
11.1 mm
8.1 mm
12.1 mm
9.1 mm
13.1 mm
10.1 mm
14.1 mm
11.1 mm
15.1 mm
12.1 mm
16.1 mm
49
3.6.4
Varying the Width of CLS, W 1
In this case, the width of the CLS, W 1 is varied to observe the effect to the
resonant frequency and the value of permittivity and permeability.
The diagram in Figure 3.10 shows that when the parameter becomes larger,
the resonant frequency becomes lower which also affect the frequency range of ε r
and µ r as shown in Table 3.10. The frequency range of negative permittivity and
permeability shift to the lower region as the width of the CLS, W 1 increases.
4.2
4.0
Frequency, GHz
3.8
3.6
3.4
3.2
3.0
2.8
2.6
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
Width, mm
Resonant frequency vs Width
Figure 3.10: Correlation between width, W 1 and resonant frequency
Table 3.10: Correlation between frequency range of negative permittivity, ε r and
negative permeability, µ r with width, W 1
Width (W 1 )
Frequency range of negative permittivity &
permeability (GHz)
1 mm
4.0 - 4.224
2 mm
3.392 - 4.016
3 mm
3.04 - 4.048
4 mm
2.784 - 3.744
50
3.6.5
Parametric Studies Conclusion
From observation, there are some parameters that have a very strong
influence to the resonant frequency and others are not significant. As examples, the
gap, G 1 and width, W 2 is varied from 0.5 mm to 1.25 mm. From observation, the
resonant frequency and the frequency range of negative ε r and µ r did not shift as the
parameters varies. It is due to the small variation of steps used in the study. If larger
steps are used, it will shift the resonant frequency and the frequency range of
negative ε r and µ r . The gap, G 2 is varied form 2 mm to 8 mm. It shows that it shifts
the resonant frequency from 4 GHz to 2.2 GHz. The frequency range of negative ε r
and µ r is also shifted from 4 GHz to 2.2 GHz. The variation of G 2 also varies other
parameters such as the length, L 1 and L 4 . The resonant frequency is shifted from 4.7
GHz to 3.3 GHz linearly after the length, L 2 is varied. The frequency range of
negative ε r and µ r is also shifted with a similar range to the resonant frequency. The
variation of L 2 also varies other parameters such as the length, L 1 , L 3 and L 4 . The
width, W 1 is varied from 1 mm to 4 mm and it shifts the resonant frequency from 3.9
GHz to 2.8 GHz. The frequency range of negative ε r and µ r is also shifted from 4.2
GHz to 2.7 GHz. In conclusion, the parameters of G 2 , L 2 and L 1 have a very strong
influence in the resonant frequency and the frequency range of the negative value of
ε r and µ r . The parameters of G 2 and L 2 and L 1 play important roles as they denote
the capacitance and inductance values that determine the operating frequencies of the
structures. If a large change in the resonant frequency needed, those three parameters
should be varied accordingly.
3.7
Simulation of the LHM unit cells in Different Size of Air Gap
The introduction of the air gaps between each cell effect the resonant
frequency and the value of ε r and µ r . In this section, an analysis has been done to
determine the consequences of the resonant frequency and the value of ε r and µ r .
The first model was simulated with a single cell without air gaps. Figure 3.11
shows the configuration of a single cell without air gap. The resonant frequency for
51
the structure is at 1.15 GHz and 4.1 GHz as shown in Figure 3.12. Figure 3.13 shows
the graph of the permittivity and permeability. The negative range of ε r and µ r
extracted from S 11 and S 21 is from 4.08 GHz to 4.67 GHz. The first resonant
frequency only produced negative value of ε r while the value of µ r is positive.
Figure 3.11: Simulation on single cell without air gap
0
S11 & S21, dB
-10
-20
-30
-40
-50
-60
0
1
2
3
4
5
6
frequency, GHz
S11
S21
Figure 3.12: Value of S 11 and S 21 of the single cell without air gap
52
60
40
20
εr, µr
0
-20
-40
-60
0
1
2
3
4
5
6
7
8
Frequency, GHz
Permittivity
Permeability
Figure 3.13: Value of ε r and µ r of the single cell without air gap
Subsequently, the size of the air gap is increased to observe its effect on the
S 11 and S 21 and the value ε r and µ r .
Figure 3.14: Simulation on single cell with air gap
53
0
S11, dB
-5
-10
-15
-20
-25
-30
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
Frequency, GHz
4 mm air gap
6 mm air gap
8 mm air gap
12 mm air gap
Figure 3.15: Value of S 11 of the unit cell with different size of air gap
0
S21, dB
-10
-20
-30
-40
-50
-60
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
Frequency, GHz
4 mm air gap
6 mm air gap
8 mm air gap
12 mm air gap
Figure 3.16: Value of S 21 of the unit cell with different size of air gap
54
8
6
value of permittivity
4
2
0
-2
-4
-6
-8
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
Frequency, GHz
4 mm air gap
6 mm air gap
8 mm air gap
12 mm air gap
Figure 3.17: Value of ε r of the unit cell with different size of air gap
1
value of permeability
0
-1
-2
-3
-4
-5
-6
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
Frequency, GHz
4 mm air gap
6 mm air gap
8 mm air gap
12 mm air gap
Figure 3.18: Value of µ r of the unit cell with different size of air gap
Figure 3.14 shows the simulated unit cell structure with an air gap. The
results of the S 11 and S 21 are shown in Figure 3.15 and Figure 3.16 respectively
55
while the results of ε r and µ r are shown in Figure 3.17 and Figure 3.18 respectively.
The simulation of the unit cell structure starts with an air gap of 4 mm for both side
of the structure. As can be seen, the structure is resonating at 1.9 GHz and after
extraction using the NRW approach, the negative value of ε r and µ r appear from
1.912 GHz to 2.74 GHz. The second simulation has been done with an air gap of 6
mm for both side of the structure. From the observation, the structure is resonating at
2.05 GHz and the range of negative value of ε r and µ r start from 2.053 GHz to 2.734
GHz. The third simulation has been done with a single unit cell with an air gap of 8
mm for both side of the structure. From the observation, the structure is resonating at
2.18 GHz and the range of negative value of ε r and µ r start from 2.11 GHz to 2.7
GHz. The final simulation has been done with an air gap of 12 mm for both side of
the LHM unit cell structure. From the observation, the structure is resonating at 2.28
GHz and the range of negative value of ε r and µ r start from 2.29 GHz to 2.575 GHz.
Five simulations on the LHM unit cells in different air gap size have been
simulated. The results show that all the simulation has a range of a negative ε r and µ r
from around 1.9 GHz to 2.7 GHz.
The unit cells with air gap of 12 mm has a smaller negative range of
permittivity and permeability compared to the 4 mm, 6 mm, and 8 mm air gap. In
order to construct the LHM structure using less dielectric substrate therefore unit cell
with 8 mm air gap is chosen. Table 3.11 shows the difference between 4 mm, 6 mm
and 8 mm air gap effect the usage of the dielectric substrate / LHM slab.
Table 3.11: Comparison between 4 mm, 6 mm and 8 mm air gap
w
Air gap = 8 mm
LHM Slab = 3
w
Air gap = 6 mm
LHM Slab = 4
w
Air gap = 4 mm
LHM Slab = 6
56
The structures have similar width, w which is 24 mm. As can be seen, the 8
mm air gap uses less dielectric substrate / LHM slab than the others. Due to this
reason, the LHM is designed with 8 mm air gap. Figure 3.19 shows the value of
permittivity, permeability and refractive index for LHM unit cell with 8 mm air gap.
4
3
2
1
Refraction
0
index, εr & µr
-1
-2
-3
-4
2.0
2.2
2.4
2.6
2.8
3.0
Frequency, GHz
Permittivity
Permeability
Refractive Index
Figure 3.19: Value of ε r , µ r and refractive index for unit cell with 8 mm air gap
From figure above, the range of negative value of ε r and µ r start from 2.11
GHz to 2.7 GHz. The refractive index at 2.4 GHz shows a value of -0.67. The
refractive index with a value of -1 occurs at 2.36 GHz. Although the refractive index
at 2.4 GHz is not met the desired value, which is -1, the LHM with -0.67 refraction
index is still considered to be integrates with the microstrip antenna and the reason
behind it is elaborated in Appendix C.
Table 3.12: Comparison between unit cell with and without air gap
Parameters
unit cell without air gap
unit cell with air gap
of 8 mm
Resonant frequency
4.1 GHz
2.18 GHz
Range of negative ε r and µ r
4.08 GHz - 4.67 GHz
2.11 GHz - 2.7 GHz
57
The comparison between simulated unit cell with and with out air gap are
shown in Table 3.12. As can be seen, the introduction of an air gap between unit cells
will shift the resonant frequency to a lower region. In conclusion, a smaller unit cell
structure size can be used for a lower frequency by simply introducing an air gaps
between unit cells.
3.8
Microstrip Antenna Design
An antenna is a device to transmit or receive electromagnetic wave.
Microstrip antenna has been chosen in this research due to it low profile,
conformable to planar and non-planar surfaces, simple and inexpensive to
manufacture using modern printed circuit technology. They are mechanically robust
when mounted with rigid surfaces, compatible with MMIC designs. When the
particular shape and mode are selected, they are very versatile in terms of resonant
frequency, polarization, pattern and impedance [2]. Despite the advantages,
microstrip antenna has a few disadvantages where they have narrow bandwidth and
low gain.
A single patch, linear polarized 2x2 array patch and circular polarized 2x2
array patch microstrip antennas is used to measure and prove the unique property of
the LHM. Appropriate to this reason, the topic onward will revealed the design
equation and procedure of the single patch microstrip antenna, linear polarized 2x2
array patch microstrip antenna and the circular polarized 2x2 array patch microstrip
antenna.
3.8.1
Single Patch Microstrip Antenna
The single patch microstrip antenna as shown in Figure 3.20 has been
designed using CST software. The antenna is fabricated on FR4 board with the
58
relative dielectric constant, ε r = 4.7, substrate thickness of 1.6 mm with tangential
loss of 0.019.
Single patch
(copper)
Substrate
(FR4 board)
Coaxial
connector
Figure 3.20: Layout of single patch microstrip antenna
A widely used probe/coaxial feeding technique have been applied in the
antenna design. The microstrip antenna was designed to operate at 2.4 GHz with the
dimension of the patch after optimization is 27.5 mm x 37 mm where the dimension
of the substrate is 11 mm x 125.5 mm. With bigger ground plane, the magnitude of
the back lobe can be reduced while it increased the gain of the antenna. The
parameters of the single patch/element can be calculated using equations below [26]:
W =
L=
c
2f
2
εr +1
c
2 f r ε reff
ε reff =
(3.1)
− 2∆L
(3.2)
ε r +1 ε r −1
2
+
2
12h 

1 +
W 

−1 / 2

W
0.412h(ε reff + 0.3) + 0.264 

h
∆L =
(ε reff − 0.258) W + 0.8 

h
Where c = speed of light
f r = resonant frequency
W = width of the patch
(3.3)
(3.4)
59
L = length of the patch
h = substrate height
ε r = dielectric constant
ε reff = effective dielectric constant
In brief, the port is placed in the centre of X-axis and 1/3 from the bottom of
the patch in Y-axis as shown in Figure 3.21. Intended for a good return loss (S 11 )
which is below -10 dB, the position of the port is then been optimized by varying the
value of its position in Y-axis [26].
x
y
Figure 3.21: Coaxial port coordinate
Figure 3.22: Return loss, S 11 of the single patch microstrip antenna
60
Figure 3.23: 3D radiation pattern at 2.4 GHz
The result in Figure 3.22 shows that the value of the return loss (S 11 ) at 2.4
GHz is -17 dB. The antenna’s bandwidth is 2.5 % from 2.38 GHz to 2.44 GHz.
Figure 3.23 shows the radiation pattern of the antenna at 2.4 GHz where the value of
directivity is 7.281 dBi and the total efficiency is around 43 %. The low value of the
total efficiency is due to the losses of the substrate used. Figure 3.24(a) shows the
polar plot of the radiation pattern in H-plane where the 3 dB beam-width is 78.70 and
Figure 3.24(b) shows the E-plane of the radiation pattern where the 3 dB beam-width
approximately similar value to the H-plane.
(a)
61
(b)
Figure 3.24: (a) Polar plot of radiation pattern at 2.4 GHz in H-plane and (b) Polar
plot of radiation pattern at 2.4 GHz in E-plane
3.8.2
Linear polarized 2x2 Array Patch Microstrip Antenna
Microstrip antenna is often used as a single element but there are some cases
arrays are also used. This array is used to increase the gain and directivity and
perform various functions beyond the capability of a single element. The
transmission line feeding technique has been used to feed the arrays [2]. A simple
power divider and quarter-wave transformer method is used in the construction of the
transmission line. The power divider is used to divide the power equally to all
junctions meanwhile the quarter-wave transformer is used for the impedance
matching between two transmission lines.
There are two important things that need to be considered in designing the
quarter-wave transformer. The first is the characteristic impedance and the second is
the length of the transmission line. The length of the transmission line should be
λ g / 4 and the characteristic impedance can be calculated with this equation [27];
Z1 = Z 0 RL
(3.5)
62
Where Z 0 = input impedance
R L = load resistance
And;
λg =
λ0
ε reff
(3.6)
Where λ g = Wavelength in substrate
The power divider can be designed by using this equation [27];
1
1
1
=
+
Z 0 Z1 Z 2
(3.7)
Where Z 0 = input characteristic impedance
Z 1 = Z 2 = output characteristic impedance
Power divider
Quarter-wave transformer
50 ohm line
Coaxial feed
Figure 3.25: Layout of the transmission line feeding technique
63
Figure 3.25 shows the detail of the transmission line that has been designed.
Inset feed matching technique is used to match the patch to the 50 ohm transmission
line [2]. The 2x2 array microstrip patch antenna as shown in Figure 3.26 has been
designed to incorporate with the LHM. The size of the antenna is 115 mm x 102 mm
and the patches size is 29 mm x 29 mm. The patches are fed by a transmission line
feeding technique and the transmission line is connected to a single SMA port. The
substrate used is FR4 board with a dielectric constant of 4.7, thickness of 1.6 mm and
a tangential loss of 0.019.
Patches
(copper)
Substrate
(FR4 board)
Transmission
line
Coaxial
connector
Figure 3.26: Layout of linear polarized 2x2 array microstrip patch antenna
Figure 3.27: Return loss, S 11 of the 2x2 array patch microstrip antenna
64
Figure 3.28: 3D radiation pattern at 2.4 GHz
Figure 3.27 shows the return loss, S 11 of the linear polarized 2x2 array
microstrip patch antenna. The antenna is resonating at 2.374 GHz with a bandwidth
from 2.34 GHz to 2.41 GHz. The bandwidth percentage is 2.9 %. Therefore, the
antenna is acceptable because the required frequency which is 2.4 GHz is still in
frequency range of the antenna. Note that, the integration of the LHM will shift the
resonant frequency to higher region and due to this reason, the antenna is designed to
operate at a lower resonant frequency.
The radiation pattern of the antenna is shown in Figure 3.28. The gain of the
antenna is 9.963 dBi at 2.4 GHz and the total efficiency is 37.6 %. The low total
efficiency of the antenna is due to the substrate loss where the value of the tangential
loss is large. Meanwhile, Figure 3.29(a) and 3.29(b) show the E-plane and H-plane of
the radiation pattern of the antenna. The 3dB beam-width of the antenna in E-plane is
550 and at the H-plane, the 3 dB beam-width is 61.80.
65
(a)
(b)
Figure 3.29: (a) Polar plot of radiation pattern at 2.4 GHz in E-plane and (b) Polar
plot of radiation pattern at 2.4 GHz in H-plane
3.8.3
Circular Polarized 2 x 2 Array Patch Microstrip Antenna
In this section, circular polarized 2x2 array patch microstrip antenna is
discussed. The transmission line feed is taken from the previous 2x2 array patch
microstrip antenna discussed in Section 3.9.2. The circular polarization can be
obtained using different feed arrangement or slight modifications to the
patch/elements. A slight modification to the patch is done to get the circular
66
polarization where the patch corner is trimmed [2]. Figure 3.30 shows the circular
polarized 2x2 array patch microstrip antenna with a cut on both corners.
Figure 3.30: Layout of Circular Polarized 2x2 Array Patch Microstrip Antenna
Figure 3.31 shows the E-field on top of the antenna where Figure 3.31(a) is
the E-field at 00, Figure 3.31(b) is the E-field at 900, Figure 3.31(c) is the E-field at
1800 and Figure 3.31(d) is the E-field at 3600. From observation, the field is rotating
from 00 to 3600 and this proves that the antenna is circular polarized.
(a)
(b)
67
(c)
0
(d)
0
Figure 3.31: (a) E-field at 0 , (b) E-field at 90 , (c) E-field at 1800 and (d) E-field at
3600
Figure 3.32: Return loss, S 11 of the 2x2 array patch circular polarized microstrip
antenna
Figure 3.33: 3D radiation pattern at 2.4 GHz
68
Figure 3.32 shows the simulated results of S 11 . The antenna has a bandwidth
of 6.4 % from 2.326 GHz – 2.48 GHz. At 2.4 GHz, the return loss of the antenna is 12.9 dB. The result of the radiation pattern is shown in Figure 3.33 where the
directivity of the antenna is 9.826 dBi with the total efficiency of 42.3 % at 2.4 GHz.
The 3 dB beam-width in the E-plane is 61.30 and at the H-plane is 57.10. The
polar plot of the radiation pattern in E and H-plane are shown in Figure 3.34(a) and
3.34(b).
(a)
(b)
Figure 3.34: (a) Polar plot of radiation pattern at 2.4 GHz in E-plane and (b)
Polar plot of radiation pattern at 2.4 GHz in H-plane
69
3.9
Chapter Summary
The design of the LHM has been presented in this chapter. Parameter of G 2 ,
L 2 and L 1 of the LHM structure has a very strong influence in the resonant frequency
and the frequency range of the negative value of ε r and µ r . The introduction of the air
gap between the unit cells shifted the resonant frequency of the LHM structure to a
lower region while reducing the usage of the dielectric substrate. The design of the
single patch, 2x2 array patch and 2x2 array patch circular polarized microstrip
antenna has also been presented in this chapter. The single patch microstrip antenna
is fed by coaxial port. A transmission line feed technique using power divider and
quarter-wave transformer has been used to design the array antenna.
CHAPTER 4
SIMULATION AND MEASUREMENT OF MICROSTRIP ANTENNA
INCORPATED WITH LEFT-HANDED METAMATERIAL
4.1
Introduction
In Chapter 3, Left-Handed Metamaterial (LHM) has been designed and
elaborated while the design procedure of the microstrip antenna also has been
revealed thoroughly. In this chapter, the antenna’s performance such as the gain,
directivity, return loss as well as other supporting characteristics of the antenna (total
efficiency & 3 dB beam-width) of the microstrip antenna incorporated with LHM
were simulated, measured and plotted.
4.2
Simulation of Single Patch Microstrip Antenna Incorporated with
LHM
Figure 4.1 shows the single patch microstrip antenna incorporated with LHM.
The front, side and back view of the structure are shown in Figure 4.2. The
dimension of the structure is 117 mm x 122.5 mm x 41.635 mm. The gap between
the LHM and the ground plane of the microstrip antenna is 12.5 mm. The substrate
that is used to realize the structure is FR-4 (Fire Retarded No.4) with a thickness of
1.6 mm, dielectric constant of 4.7 and a tangential loss of 0.019.
71
12.5 mm
(a)
(b)
(c)
(d)
Figure 4.1: (a) Dimension of the microstrip antenna incorporated with LHM with
three different views where (b) front view, (c) side view and (d) back view
Figure 4.2 shows the return loss, S 11 results of the single patch microstrip
antenna incorporated with the LHM structure. The return loss at 2.4 GHz is -9.88 dB.
From this figure, it is noticed that the operating frequency is between 2.4 GHz and
2.5 GHz while the percentage of bandwidth is 4 %.
72
Figure 4.2: return loss, S 11
The results of the 3D radiation pattern of three different operating frequencies
are shown in figure 4.3(a) and 4.3(b) and 4.3(c). Figure 4.3(a) and 4.3(b) depicted the
3D radiation pattern at frequency of 2.45 GHz and 2.48 GHz respectively. It is
noticed that the radiation pattern are non directional where the beam were not
focused to one point and the directivities are low. It prove that the LHM structure is
not act as focusing device at 2.45 GHz and 2.48 GHz. Compared with the Figure
4.3(c), the radiation pattern is directional where the beam is focusing to one point
and the directivity is the highest between those three. The directivity is 11.5 dBi and
the total efficiency is 41 % at 2.4 GHz.
(a)
73
(b)
(c)
Figure 4.3: (a) Simulated radiation patterns at 2.45 GHz, (b) Simulated radiation
patterns at 2.48 GHz and (c) Simulated radiation patterns at 2.4 GHz
Figure 4.4(a) and 4.4(b) shows the polar plots of the radiation pattern at 2.4
GHz and it is observed that the 3dB beam-width of the E-plane is 48.50 and the Hplane is 34.30.
74
(a)
(b)
Figure 4.4: (a) Polar plot of the radiation pattern in E-plane at 2.4 GHz and (a) Polar
plot of the radiation pattern in H-plane at 2.4 GHz
Parametric study on the gap distance of the microstrip antenna and the LHM
structure has been carried out to study the effects on the return loss of the antenna.
From Figure 4.5, it is noticed that only at the gap of 12.5 mm, the return loss around
-10 dB from 2.4 GHz to 2.5 GHz can be acquired. At 2 mm gap and 4.5 mm gap, the
resonant frequency is shifted to 2.35 GHz and observed that at 8.5 mm gap, the
return loss is a bit distorted. At 10.5 mm gap, the return loss is getting better where
the operating frequency is from 2.41 GHz to 2.51 GHz. Larger gaps which are higher
than 12.5 mm products less gain for the antenna. The gap between the antenna and
LHM need to be less than half of the length of LHM. The value of half of the length
of LHM is 14.55 mm as shown in Figure 4.5. Figure 4.6(a) shows how the wave
focus with 12.5 mm gap while Figure 4.6(b) shows how the wave focus with the gap
larger than 14.55 mm. as can be seen, Figure 4.6(a) create bigger wave than Figure
4.6(b) due to the gap between the antenna and LHM. Bigger wave produced in front
of the LHM creates higher gain than the smaller one.
14.55 mm
Figure 4.5: LHM structure
75
(a)
(b)
Figure 4.6: (a) Focusing effect of 12.5 mm gap and (b) Focusing effect of larger than
14.55 mm gap
From the S 11 results in Figure 4.7, the gap of the antenna and LHM structure
certainly effect the return loss of the antenna and produced some minor shift to the
resonant frequency.
Figure 4.7: Resonant frequency shift as the gap between the antenna and the LHM
structure varies
76
4.3
Simulation of Linear Polarized 2 x 2 Array Patch Microstrip Antenna
Incorporated with LHM
A linear polarized 2x2 array patch microstrip antenna has been designed and
incorporated with the LHM. Figure 4.8 shows perspective and front view of the
antenna incorporated with LHM. The LHM is placed 12.5 mm in front of the
antenna. The gap between the microstrip antenna and the LHM is determined base
from the previous parametric studies in Section 4.1. The dimension of the whole
structure is 117 mm X 122.5 mm X 41.635 mm. The substrate that is used to realize
the structure is FR-4 board (Fire Retarded No.4) with a thickness of 1.6 mm,
dielectric constant of 4.7 and a tangential loss of 0.019.
12.5 mm
(a)
(b)
Figure 4.8: Linear polarized 2x2 array patch microstrip antenna incorporated
with LHM (a) Perspective view and (b) front view
77
Figure 4.9 shows the simulated return loss, S 11 of the antenna incorporated
with LHM. The deepest dip of return loss is at 2.433 GHz with -25 dB and the
bandwidth is around 2.4% from 2.4 GHz to 2.46 GHz.
Figure 4.9: Return loss, S 11
(a)
(b)
Figure 4.10: (a) Simulated radiation patterns at 2.4 GHz and (b) Simulated radiation
patterns at 2.45 GHz
78
The 3D radiation pattern of two different frequencies is shown in Figure
4.10(a) and 4.10(b). The 3D radiation pattern has been observed at the operating
frequency of 2.4 GHz and 2.45 GHz. The directivity of the antenna at 2.4 GHz is
higher then the directivity at 2.45 GHz which is 12.4 dBi compared to the 12.28 dBi
and noticed that the total efficiency of the antenna at 2.4 GHz is better then the total
efficiency at 2.45 GHz. The total efficiency at 2.4 GHz is 34.2% compared to the
total efficiency at 2.45 GHz which is 23%.
Figure 4.11(a) and 4.11(b) shows a 2D polar plot of the radiation pattern at
2.4 GHz and it is noticed that the 3dB beam-width of the E-plane is 38.20 and the Hplane is 42.10.
(a)
(b)
Figure 4.11: (a) Polar plot of the radiation pattern in E-plane at 2.4 GHz and (b)
Polar plot of the radiation pattern in H-plane at 2.4 GHz
4.4
Simulation of Circular Polarized 2 x 2 Array Patch Microstrip Antenna
Incorporated with LHM
The circular Polarized 2x2 array patch microstrip antenna incorporated with
LHM has been designed using CST software. The dimension of the whole structure
is 117 mm X 122.5 mm X 41.635 mm. The LHM is placed 12.5 mm in front of the
microstrip antenna. The substrate that is used to construct the LHM and the antenna
is FR-4 (fire retarded No.4) with a thickness of 1.6mm, dielectric constant of 4.7 and
79
a tangential loss of 0.019. Figure 4.12 shows the perspective and front view of the
structure itself.
12.5 mm
(a)
(b)
Figure 4.12: Circular polarized 2x2 array patch microstrip antenna incorporated
with LHM (a) Perspective view and (b) front view
The result of the return loss, S 11 shows that the antenna has two resonant
frequencies which are 2.35 GHz and 2.47 GHz with a return loss at -12.15 dB and 14 dB. Meanwhile, the return loss at 2.4 GHz is -8.32 dB and the bandwidth of the
antenna is from 2.3 GHz to 2.38 GHz and 2.42 GHz to 2.5 GHz.
80
Figure 4.13: Return loss, S 11
Due to the unsatisfied S 11 results obtained, a parametric study on the gap of
the microstrip antenna and LHM has been done to observe the effects of the return
loss, S 11 of the antenna. From Figure 4.14, it is noticed that only at 12.5 mm gap, the
lowest return loss can be acquired at 2.4 GHz which is around -8.3 dB. At 10.5 mm
gap, the resonant frequency is shifted to 2.49 GHz and noticed that at 1.635 mm gap,
the return loss is a bit distorted. At 7.135 mm gap, the return loss is better then others
where it resonating at 2.3 GHz and 2.48 GHz but at 2.4 GHz, the return loss is -7.3
dB. In conclusion, the 12.5 mm gap is chosen due to the lowest return loss at 2.4
GHz. The gaps with the value more than 12.5 mm are not chosen due to similar
reason discussed in sub-chapter 4.2.
Figure 4.14: Return loss altered as the gap between the antenna and the LHM varies
81
Figure 4.15(a), 4.15(b) and 4.15(c) show the 3D radiation pattern of the
antenna at 3 different frequencies. Figure 4.15(a) shows the radiation pattern at 2.35
GHz and note that the directivity of the antenna is 10.38 dBi with a total efficiency of
35.3 %. At 2.4 GHz, the radiation pattern shown in Figure 4.15(b) indicates that the
directivity of the antenna is 10.39 dBi with a total efficiency of 29 %. The radiation
pattern shown in Figure 4.15(c) is simulated at 2.45 GHz and the results of the
directivity is 8.879 dBi and the total efficiency of the antenna is 23.7 %
(a)
(b)
82
(c)
Figure 4.15: (a) Simulated radiation patterns at 2.35 GHz, (b) Simulated radiation
patterns at 2.4 GHz and (c) Simulated radiation patterns at 2.45 GHz
Figure 4.16(a) shows the polar plot of the radiation pattern in E-plane while
Figure 4.16(b) shows the polar plot of the radiation pattern in H-plane. Observation
from the polar plot indicates that at 2.4 GHz the 3 dB beam-width of the E-plane is
50.70 and the H-plane is 47.50.
(a)
(b)
Figure 4.16: (a) Polar plot of the radiation pattern in E-plane at 2.4 GHz and (b)
Polar plot of the radiation pattern in H-plane at 2.4 GHz
83
4.5
Measurement Result
After obtaining such encouraging results in the simulation, the design is
fabricated and measured. The fabrication process has been done using wet etching
technique and its procedure is discussed in Appendix E.
The measurement is taken place at the P18, Faculty of Electrical Engineering
(FKE), Universiti Teknologi Malaysia (UTM). The anechoic chamber available here
and others equipment such as vector network analyzer, signal generator, horn
antenna and spectrum analyzer has been used to perform this measurement. Figure
4.17 (a) shows the equipments used in the measurement and 4.17(b) shows the layout
of the measurement setup for S 11 and S 21 parameters. The S 21 parameter is measured
in order to obtain the gain of the antenna with and without LHM. Figure 4.18 (a)
shows the equipments setup for the radiation pattern measurement and 4.18 (b)
shows the layout of the radiation pattern measurement setup.
(a)
84
(b)
Figure 4.17: (a) Measurement setup using network analyzer and (b) Layout of the
S 11 and S 21 parameters measurement setup
(a)
(b)
Figure 4.18: (a) Measurement equipment for radiation pattern measurement (from
left; signal generator, spectrum analyzer, antenna measurement rotator and rotator
within an anechoic chamber) and (b) Layout of the radiation pattern measurement
setup
85
Vector network analyzer is used to measure S 11 and S 21 parameter. The
measurement has been done in an anechoic chamber in such a way it will reduces the
reflection and produce good measurement results. The radiation pattern is measured
in the same anechoic chamber using different equipments such as signal generator
and spectrum analyzer.
4.5.1
Measurement of Single Patch Microstrip Antenna Incorporated with
LHM
Figure 4.19(a) shows the fabricated single patch microstrip antenna and
Figure 4.19(b) shows the fabricated LHM. The fabricated single patch microstrip
antenna incorporated with LHM is shown in Figure 4.20. The substrate that is used to
construct the LHM and the antenna is FR-4 (fire retarded No.4) with a thickness of
1.6 mm, dielectric constant of 4.7 and a tangential loss of 0.019. The dimensions of
the structures are similar as in the simulation and the gap between the ground plane
of the microstrip antenna and the LHM is approximately 12.5 mm.
(a)
(b)
Figure 4.19: (a) The fabricated single patch microstrip antenna and (b) the fabricated
LHM
86
Figure 4.20: Single patch microstrip antenna incorporated with LHM
The measurement of the single patch microstrip antenna is carried out. The
measurement results of the S 11 and S 21 are shown in Table 4.1.
Table 4.1: The results of the measured single patch microstrip antenna
Parameters
S 11
S 21
Value
Minimum value
-18.68 dB at 2.39 GHz
Value at 2.4 GHz
-17.488 dB
bandwidth
2.36GHz - 2.43GHz
Maximum value
-35.7 dB at 2.36 GHz
Value at 2.4 GHz
-36.1 dB
From observation, it shows a good return loss, S 11 result where at 2.4GHz the
return loss is at -17.488 dB and the bandwidth of the single patch microstrip antenna
is between 2.36GHz and 2.43GHz. The percentage of the bandwidth is around 2.9 %.
At 2.4 GHz, the transmission coefficient, S 21 result is -36.1 dB.
The measurement results of the LHM incorporated with the single patch
microstrip antenna are shown in Table 4.2.
87
Table 4.2: The results of the measured single patch microstrip antenna incorporated
with LHM
Parameters
S 11
S 21
Value
Minimum value
-17.835 dB at 2.42 GHz
Value at 2.4 GHz
-15.71 dB
bandwidth
2.35 GHz to 2.47 GHz
Maximum value
-31.32 dB at 2.38 GHz
Value at 2.4 GHz
-31.734 dB
Table 4.2 shows the return loss, S 11 at 2.4 GHz is at -15.71 dB. The
bandwidth of the antenna is around 4.98 % which is from 2.35 GHz to 2.47 GHz.
The transmission coefficient, S 21 result at 2.4 GHz is -31.734 dB. It shows that the
S 21 results improved up to 4.3 dB after the incorporation of LHM.
single patch microstrip antenna
0
-40
330
30
-45
-50
-55
300
60
-60
-65
-70
-75
270
-80
90
-40 -45 -50 -55 -60 -65 -70 -75 -80 -75 -70 -65 -60 -55 -50 -45 -40
-75
-70
-65
-60
240
120
-55
-50
-45
210
150
-40
180
E-co
E-crross
H-co
H-cross
Figure 4.21: Measured radiation pattern of the single patch microstrip antenna
88
single patch microstrip antenna incorparated with LHM
0
330
30
-40
-45
-50
-55
300
60
-60
-65
-70
-75
270
-80
-40 -45 -50 -55 -60 -65 -70 -75 -80 -75 -70 -65 -60 -55 -50 -45 -40
-75
90
-70
-65
-60
240
120
-55
-50
-45
210
-40
150
180
E-co
E-cross
H-co
H-cross
Figure 4.22: Measured radiation pattern of the single patch microstrip antenna with
LHM
Figure 4.21 shows the measured radiation pattern of the single patch
microstrip antenna. The power received at the antenna is -40 dBm with a 3 dB beamwidth of 900 in E-plane and 830 in H-plane. Meanwhile, Figure 4.22 shows the
measured radiation pattern of the single patch microstrip antenna incorporated with
LHM. The power received at the antenna is -36 dBm and the 3 dB beam-width is 560
in E-plane and 380 in H-plane. The cross polar isolation of both measured radiation
pattern shows a good results which is higher than 20 dB.
4.5.2
Measurement of Linear Polarized 2 x 2 Array Patch Microstrip Antenna
Incorporated with LHM
The linear polarized 2x2 array microstrip patch antenna is fabricated and
measured. The design procedure has been explained in Chapter 3. Figure 4.23 shows
the fabricated antenna with and without the LHM. The fabricated LHM is placed
approximately 12.5 mm in front of the antenna. The substrate that is used to
89
construct the LHM and the antenna is FR-4 (fire retarded 4) with a thickness of 1.6
mm, dielectric constant of 4.7 and a tangential loss of 0.019 while the dimension of
both structures is similar as in the simulation.
Figure 4.23: Perspective view of the linear polarized 2x2 array patch microstrip
antenna with and without LHM
Table 4.3: The results of the measured linear polarized 2x2 array patch microstrip
antenna
Parameters
S 11
S 21
Value
Minimum value
-30.153 dB at 2.43 GHz
Value at 2.4 GHz
-10.212 dB
bandwidth
2.395 GHz to 2.465 GHz
Maximum value
-32.08 dB at 2.42 GHz
Value at 2.4 GHz
-32.684 dB
Table 4.3 shows the results of the return loss, S 11 and the transmission coefficient,
S 21 of the linear polarized 2x2 array patch microstrip antenna. The return loss, S 11
shows that the resonance frequency is at 2.43 GHz and the bandwidth is from 2.395
GHz to 2.465 GHz. The transmission coefficient, S 21 of the antenna whereas the
maximum value of S 21 is at 2.42 GHz with -32.08 dB.
90
Table 4.4: The results of the measured linear polarized 2x2 array patch microstrip
antenna incorporated with LHM
Parameters
S 11
S 21
Value
Minimum value
-17.478 dB at 2.48 GHz
Value at 2.4 GHz
-7.635 dB
bandwidth
2.455 GHz to 2.515 GHz
Maximum value
-31.01 dB at 2.37 GHz
Value at 2.4 GHz
-30.631 dB
Table 4.4 tabulated the results of linear polarized 2x2 array patch microstrip
antenna integrated with LHM. The outcomes prove the function of the LHM. The
result of the return loss, S 11 shows that the deepest dip is at 2.48 GHz while the
bandwidth is from 2.455 GHz to 2.515 GHz. However, the result of the S 21 shows
that the minimum losses occur at 2.37 GHz with -31.01 dB loss although the return
loss at the similar frequency is -7.395 dB. The transmission coefficient, S 21 at 2.4
GHz is around -31 dB with a return loss of -7.6 dB.
2X2 Array Patch Microstrip Antenna
0
330
-40
30
-45
-50
-55
300
60
-60
-65
-70
-75
270
-80
-40 -45 -50 -55 -60 -65 -70 -75 -80 -75 -70 -65 -60 -55 -50 -45 -40
-75
90
-70
-65
-60
240
120
-55
-50
-45
210
-40
150
180
e-co
e-cross
h-co
h-cross
Figure 4.24: Measured radiation pattern of the linear polarized 2x2 array patch
microstrip antenna
91
2X2 Array Patch Microstrip Antenna with LHM
0
330
30
-40
-50
60
300
-60
-70
270
-80
-40
-50
-60
-70
90
-80
-70
-60
-50
-40
-70
-60
240
120
-50
210
-40
150
180
e-co
e-cross
h-co
h-cross
Figure 4.25: Measured radiation pattern of the linear polarized 2x2 array patch
microstrip antenna with LHM
The focusing effect of the LHM can be observed from the radiation pattern
where the beam is narrower after the integration of the LHM. Figure 4.24 shows the
polar plot radiation pattern of the linear polarized 2x2 array patch microstrip antenna
at 2.4 GHz. The main lobe magnitude is -37 dBm while the 3dB beam-width of Eplane is 480 and the H-plane is 440. Meanwhile, the polar plot radiation pattern of
linear polarized 2x2 array patch microstrip antenna incorporated with LHM at the
same frequency as before is illustrated in Figure 4.25. The main lobe magnitude is 35 dBm while the 3dB beam-width of E-plane is 480 and the H-plane is 380.
Comparison between measured results of radiation pattern and transmission
coefficient at 2.4 GHz shows similar increment in the magnitude after the
incorporation of the LHM. The cross polarization for both radiation pattern shows a
good results which is higher than 20 dB.
92
4.5.3
Measurement of Circular Polarized 2 x 2 Array Patch Microstrip
Antenna Incorporated with LHM
A circular polarized antenna in the present of the LHM is fabricated and
measured. The parameters such as the S 11 , S 21 , gain and the radiation pattern is
analyzed and the effect of the circular polarization signal propagate through the
LHM are investigated. Figure 4.26 shows the fabricated circular polarized 2x2 array
patch microstrip antenna without and with integration of the LHM.
Figure 4.26: Perspective view of the circular polarized 2x2 array patch microstrip
antenna with and without LHM
Table 4.5 shows the results of S 11 and S 21 of the antenna itself. The antenna
is resonating at 2.45 GHz with a bandwidth from 2.40 GHz to 2.59 GHz. The
bandwidth percentage is 7.6 %. The maximum value of transmission coefficient, S 21
is at 2.37 GHz which is around –34 dB.
Table 4.5: The results of the measured circular polarized 2x2 array patch microstrip
antenna
Parameters
S 11
S 21
Value
Minimum value
-30.454 dB at 2.45 GHz
Value at 2.4 GHz
-9.8 dB
bandwidth
2.40 GHz to 2.59 GHz
Maximum value
-34.78 dB at 2.37 GHz
Value at 2.4 GHz
-35.35dB
93
Table 4.6: The results of the measured circular polarized 2x2 array patch microstrip
antenna incorporated with LHM
Parameters
S 11
Value
Minimum value
-14.454 dB at 2.45 GHz
Value at 2.4 GHz
-10.977 dB
bandwidth
2.38 GHz to 2.61 GHz
Maximum value
-30.462 dB at 2.37 GHz
Value at 2.4 GHz
-31.381 dB
S 21
The antenna incorporated with the LHM is resonating at 2.45 GHz as shown
in Table 4.6. The bandwidth of the antenna is from 2.38 GHz to 2.61 GHz with a
percentage of 9.2 %. The maximum value of transmission coefficient, S 21 is at 2.37
GHz with -30.462 dB. The S 21 results shows an increment of gain up to 4 dB at 2.4
GHz and the results of S 11 shows improvement in the bandwidth.
The measurement of the cross polar of the antenna has been done by placing
the antenna horizontal. Table 4.7 shows the S 11 and S 21 results, where the antenna is
resonating at 2.45 GHz with a bandwidth from 2.4 GHz to 2.59 GHz and the
percentage is 7.6 %. The maximum value of transmission coefficient, S 21 is at 2.47
GHz with –34.25 dB. The antenna is then incorporated with LHM and the results of
S 11 and S 21 are shown in Table 4.8. The antenna is resonating at 2.45. The bandwidth
has been improved from 2.38 GHz to 2.61 GHz with the percentage is around 9.2 %.
The maximum value of transmission coefficient, S 21 is -34.6 dB at 2.46 GHz.
Generally, the S 21 results of the antenna incorporated with LHM did not show any
different between the S 21 results of the antenna itself which mean there is no
increment of gain occur.
94
Table 4.7: The results of the measured circular polarized 2x2 array patch microstrip
antenna in cross polar position
Parameters
S 11
S 21
Value
Minimum value
-30.454 dB at 2.45 GHz
Value at 2.4 GHz
-9.8 dB
bandwidth
2.4 GHz to 2.59 GHz
Maximum value
-34.25 dB at 2.47 GHz
Value at 2.4 GHz
-36.363 dB
Table 4.8: The results of the measured circular polarized 2x2 array patch microstrip
antenna incorporated with LHM in cross polar position
Parameters
S 11
S 21
Value
Minimum value
-14.454 dB at 2.45 GHz
Value at 2.4 GHz
-10.977 dB
bandwidth
2.38 GHz to 2.61 GHz
Maximum value
-34.6 dB at 2.46 GHz
Value at 2.4 GHz
-38.374 dB
The radiation pattern of the antenna is measured and the results are shown in
Figure 4.27 and Figure 4.28. Figure 4.27 shows the radiation pattern of the antenna
itself where the power received at the antenna is -40 dBm and the 3 dB beam-width
is 480 in E-plane and 620 in H-plane. The cross polarization of the antenna is 2 dB
which is acceptable for a circular polarized antenna. Meanwhile, Figure 4.28 shows
the radiation pattern of the antenna incorporated with LHM and the power received
at the antenna is -36 dBm which increase up to 4 dB compared to the antenna itself.
The 3 dB beam-width of the antenna is 400 in E-plane and 420 in H-plane. The cross
polarization of the antenna is 7 dB.
95
circular polarized 2X2 array patch microstrip antenna
0
-40
330
30
-45
-50
300
60
-55
-60
-65
-70
270
90
-40
-50
-45
-55
-60
-65 -70
-70 -70
-65
-60
-55
-50
-45
-40
-65
-60
-55
240
120
-50
-45
210
150
-40
180
E-co
E-cross
H-co
H-cross
Figure 4.27: Measured radiation pattern of the circular polarized 2x2 array patch
microstrip antenna
circular polarized 2X2 array patch microstrip antenna
incorparated with LHM
0
-35
330
30
-40
-45
-50
300
60
-55
-60
-65
270
-35
-40
-45
-50
-55
-60
-70
-70
-65 -70-70
90
-65
-60
-55
-50
-45
-40
-35
-65
-60
-55
240
120
-50
-45
-40
210
150
-35
180
E-co
E-cross
H-co
H-cross
Figure 4.28: Measured radiation pattern of the circular polarized 2x2 array patch
microstrip antenna with LHM
96
4.6
Chapter Summary
The simulation and the measurement of the LHM incorporated with the
microstrip antennas have been presented in this chapter. A single patch, linear
polarized 2x2 array patch and circular polarized 2x2 array patch microstrip antennas
has been incorporated with the LHM. The results such as the gain, directivity, return
loss as well as other supporting characteristics of antenna (total efficiency & 3 dB
beam-width) has been tabulated in the given table.
CHAPTER 5
ANALYSIS & DISCUSSION
5.1
Introduction
Subsequent to the chapter 4, this chapter presents the analysis results and the
discussion about the effect of the antennas incorporated with LHM. The comparison
between antennas incorporated with and without LHM for single patch, linear
polarized 2x2 array patch and circular polarized 2x2 array patch microstrip antenna
is tabulated and discussed. On the other hand, the comparison between simulated and
measured results is also presented. The focusing characteristics of the LHM
narrowed the beam-width of the antenna hence increases its gain.
5.2
Analysis and Discussion on Simulation of the Single Patch Microstrip
Antenna Incorporated with LHM
The Return Loss, S 11 holds the vital key in any antenna design as it portrays
how well the signal could be transmitted from the antenna. For an antenna to work
properly, the return loss should be below -10 dB at the desired frequency range. The
gain is the main characteristic of interest in this research and it is very important
especially in microstrip antennas. As other researchers have been trying to increase
the gain by introducing a lot of other methods such as making use of a supersubstrate
of either high permittivity or permeability above the patch antenna [20] and to
sandwich the antenna by dielectric layers of the same permittivity [21], a few has
98
tried to use LHM to increase the gain of the antenna and it has proven true as
predicted by [17] and other researchers through simulations and measurements.
Figure 5.1 shows the comparison of the return loss, S 11 between single patch
microstrip antenna with the single patch microstrip antenna incorporated with LHM.
The resonant frequency is shifted to a higher frequency region from 2.4 GHz to 2.47
GHz and although it is altered, the S 11 at 2.4 GHz is still around -10 dB. The
bandwidth percentage of the antenna is increased after incorporating with LHM from
2.5 % to 4 %.
5
0
S11, dB
-5
-10
-15
-20
-25
1.8
2.0
2.2
2.4
2.6
2.8
3.0
3.2
Frequency, GHz
Single patch microstrip antenna
Single patch microstrip antenna incorporated with LHM
Figure 5.1: Return loss, S 11 of the single patch microstrip antenna incorporated with
LHM
The simulated radiation pattern for both E-plane and H-plane are shown in
Figure 5.2 and Figure 5.3. The radiation patterns are simulated at the frequency of
2.4 GHz. The gain of the antenna increases up to 4.22 dB for both plane after the
introduction of the LHM. The 3 dB beam-width in E-plane become narrower from
780 to 48.50 and in H-plane, from 78.70 to 34.30. The side lobes are also visible. In a
nutshell, the improvement of gain is obvious where the antenna with LHM showed a
more directed beam.
99
E-plane
0
10
330
30
5
0
-5
300
60
-10
-15
-20
270
10
5
0
-25
-10 -15 -20 -25 -20 -15 -10
-20
-5
90
-5
0
5
10
-15
-10
240
120
-5
0
5
10
210
150
180
Single patch microstrip antenna
Single patch microstrip antenna incorporated with LHM
Figure 5.2: Radiation Patterns in E-plane
H-plane
0
330
30
10
5
300
60
0
-5
270
90
-10
10
5
0
-5
-10
-5
0
5
10
-5
0
240
120
5
210
10
150
180
Single patch microstrip antenna
Single patch microstrip antenna incorporated with LHM
Figure 5.3: Radiation Patterns in H-plane
From the results, the introduction of the LHM to the single patch microstrip
antenna has made the performance of the antenna improved. In order to provide a
clear view, comparison of the antenna with and without LHM is tabulated in Table
5.1.
100
Table 5.1: Comparison of the antenna’s performance between single patch
microstrip antenna with and without LHM
Antenna’s parameters at 2.4
Single patch microstrip
Single patch microstrip
GHz
antenna
antenna incorporated
with LHM
Return loss, S11
-16 dB
-10 dB
bandwidth
2.5 %
4%
(2.38 GHz – 2.44 GHz)
(2.4 GHz – 2.5 GHz)
7.281 dBi
11.5 dBi
Directivity
3dB
E-plane
77.80
48.50
beam-
H-plane
78.70
34.30
Total efficiency
43 %
41 %
Front to back lobe ratio
16.93 dB
19.94 dB
width
5.2.1
E-field analysis in the present of LHM in front of the single patch
microstrip antenna
Figure 5.4 (a) showed the E-Field of a typical microstrip antenna operating at
2.4 GHz while Figure 5.4(b) depicted the E-Field of the antenna in the presence of
LHM in E-plane. Meanwhile, Figure 5.5(a) illustrated the E-field of a microstrip
antenna while Figure 5.5(b) shows the E-field of the antenna incorporated with LHM
and both of them were observed in H-plane. It was observed that the E-Field is more
directed once it left the LHM. In a nutshell, the LHM had the ability to focus the
waves and this in turns explained the gain increase in the antenna.
101
(a)
(b)
Figure 5.4: (a) Observation on E-field in E-plane for the single patch microstrip
antenna and (b) Observation on E-field in E-plane for the single patch microstrip
antenna incorporated with LHM
(a)
(b)
Figure 5.5: (a) Observation on E-field in H-plane for the single patch microstrip
antenna and (b) Observation on E-field in H-plane for the single patch microstrip
antenna incorporated with LHM
102
5.3
Analysis and Discussion on Simulation of Linear Polarized 2x2 Array
Patch Microstrip Antenna Incorporated with LHM
The simulated return loss, S 11 of the linear polarized 2x2 array mircostrip
patch antenna with and without LHM is shown in Figure 5.6. The introduction of the
LHM in front of the antenna altered the resonant frequency to a higher region,
shifting it from 2.37 GHz to 2.433 GHz and the bandwidth of the antenna also
become narrower. The return loss at 2.4 GHz is still below -10 dB after incorporating
the LHM.
0
S11, dB
-10
-20
-30
-40
1.8
2.0
2.2
2.4
2.6
2.8
3.0
3.2
Frequency, GHz
2x2 array patch microtrip antenna
2x2 array patch microtrip antenna incorporated with LHM
Figure 5.6: Return loss, S 11 of the linear polarized 2x2 array mircostrip patch
antenna
Figure 5.7(a) shows the comparison of 3D radiation pattern of the antenna
integrated with LHM and Figure 5.7(b) shows the radiation pattern of the antenna
itself. All radiation patterns are simulated at the frequency of 2.4 GHz. Referring to
the radiation pattern in Figure 5.8, the radiated signal seemed more directed after the
insertion of LHM with the 3 dB Beam-width of 42.1o from the previous 55o in Eplane. Meanwhile, Figure 5.9 shows the radiation pattern in H-plane, the 3 dB beamwidth become narrower from 61.8o to 42.1o. This shows a decrement around 18.3o in
beam-width for both planes. The directivity of the antenna increased from 10.0 dBi
to 12.3 dBi which represented an increment of approximately 2.3 dB.
103
(a)
(b)
Figure 5.7: (a) 3D radiation pattern of the linear polarized 2x2 array patch microstrip
antenna incorporated with LHM and (b) 3D radiation pattern of the linear polarized
2x2 array patch microstrip antenna
104
E-plane
0
330
30
10
5
0
300
60
-5
-10
-15
270
10
5
0
-5
-10
-15
-20
-20
-20-20
90
-15
-10
-5
0
5
10
-15
-10
-5
240
120
0
5
10
210
150
180
2x2 array patch microtrip antenna
2x2 array patch microtrip antenna incorporated with LHM
Figure 5.8: Radiation pattern in E-plane
H-plane
0
10
330
30
0
300
60
-10
-20
-30
270
90
10
0
-10
-20
-30
-30 -30
-20
-10
0
10
-20
-10
240
120
0
210
10
150
180
2x2 array patch microtrip antenna
2x2 array patch microtrip antenna incorporated with LHM
Figure 5.9: Radiation pattern in H-plane
On the other hand, the integration of the LHM with the antenna introduces
both side lobe and back lobe. The existent of the side and back lobe are due to the
LHM structure itself. From observation, it seems that some of the energy radiated to
the side of the LHM structure and some are reflected to the back of the antenna.
These unwanted radiations contribute to the side lobe and back lobe.
105
As a whole, the gain and directivity of the antenna have increased in the
presence of LHM. Table 5.2 shows the comparison of the linear polarized 2x2 array
patch microstrip antenna with and without LHM.
Table 5.2: Comparison of the antenna’s performance between linear polarized 2x2
array patch microstrip antenna with and without LHM
Antenna parameters at
Linear polarized 2x2
Linear polarized 2x2 array
2.4 GHz
array patch microstrip
patch microstrip antenna
antenna
incorporated with LHM
Return loss, S11
-11.5 dB
-10 dB
Bandwidth
2.9 %
2.4 %
(2.34 GHz – 2.41 GHz)
(2.4 GHz – 2.46 GHz)
9.96 dBi
12.4 dBi
Directivity
3dB beam-
E-plane
550
38.20
width
H-plane
61.80
42.10
Total efficiency
37.6 %
34.2 %
Front to back lobe ratio
28.8 dB
13.87 dB
5.4
Analysis and Discussion on Simulation of Circular Polarized 2x2 Array
Patch Microstrip Antenna Incorporated with LHM
Figure 5.10 shows the comparison of return loss, S 11 between circular
polarized 2x2 array patch microstrip antenna and circular polarized 2x2 array patch
microstrip antenna in the present of LHM. The introduction of LHM produced 2
resonant frequencies at 2.35 GHz and 2.48 GHz and at the frequency of 2.4 GHz, the
return loss is -8.3 dB.
106
0
-2
-4
S11, dB
-6
-8
-10
-12
-14
-16
1.8
2.0
2.2
2.4
2.6
2.8
3.0
3.2
Frequency, GHz
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.10: Return loss, S 11 of the circular polarized 2x2 array patch microstrip
antenna
E-plane
0
10
330
30
5
0
300
60
-5
-10
-15
270
10
5
0
-5
-10
-20
-15 -20
-15
90
-15
-10
-5
0
5
10
-10
-5
240
120
0
5
210
150
10
180
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.11: Radiation pattern in E-plane
107
H-plane
0
10
330
30
5
0
300
60
-5
-10
-15
270
10
5
0
-5
-10
-20
-15 -20
-15
90
-15
-10
-5
0
5
10
-10
-5
240
120
0
5
210
150
10
180
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.12: Radiation pattern in H-plane
The radiation patterns for both antennas with and without LHM can be
observed in Figure 5.11 and Figure 5.12. Figure 5.11 shows the radiation pattern in
E-plane and Figure 5.12 shows the radiation pattern in H-plane and all radiation
patterns are simulated at 2.4 GHz. The 3 dB beam-width in E-plane for both antenna
have similar value and the 3 dB beam-width for H-plane for antenna incorporated
with LHM has smaller value than the antenna itself. It is due to the focusing effect
created by the LHM.
Table 5.3 shows the comparison between the circular polarized 2x2 array
patch microstrip antenna with and without LHM.
108
Table 5.3: Comparison of the antenna’s performance between circular polarized 2x2
array patch microstrip antenna with and without LHM
Antenna parameters at
Circular polarized 2x2
Circular polarized 2x2
2.4 GHz
array patch microstrip
array patch microstrip
antenna
antenna incorporated
with LHM
Return loss, S 11
-12.07 dB
-8.32 dB
Bandwidth
6.4 %
2.2 %
(2.326 GHz – 2.48 GHz)
(2.327 GHz – 2.38 GHz)
2.8 %
(2.44 GHz – 2.51 GHz)
Directivity
9.826 dBi
10.39 dBi
3dB
E-plane
61.30
47.50
beam-
H-plane
57.10
50.70
Total efficiency
42.3 %
29 %
Front to back lobe ratio
23.28 dB
11.26 dB
width
5.5
Analysis and Discussion on Measurement of the Single Patch Microstrip
Antenna Incorporated with LHM
Figure 5.13 shows the S 11 measurement results and Figure 5.14 shows the S 21
measurement results. From Figure 5.13, the results show a good agreement between
both measurements. The return loss, S 11 for the single patch microstrip antenna and
single patch microstrip antenna incorporated with LHM at 2.4GHz is below -10 dB
which indicate more than 90% radiation has been transmitted. The bandwidth of
single patch microstrip antenna incorporated with LHM is wider that the single patch
microstrip antenna. Figure 5.14 shows the S 21 measurement for both results. An
increment up to 4.366 dB at 2.4 GHz is noticed from the graph.
109
0
S11, dB
-5
-10
-15
-20
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
Frequency, GHz
Single patch microstrip antenna
Single patch microstrip antenna incorporated with LHM
Figure 5.13: Return loss, S 11 of the single patch microstrip antenna incorporated
with and without LHM
-20
-30
S21, dB
-40
-50
-60
-70
-80
-90
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
Frequency, GHz
Single patch microstrip antenna
Single patch microstrip antenna incorporated with LHM
Figure 5.14: Transmission coefficient, S 21 of the single patch microstrip antenna
incorporated with and without LHM
The measurement of the radiation pattern for both single patch microstrip
antennas with and without LHM has been performed at a frequency of 2.4GHz.
Figure 5.15 shows the comparison of the radiation pattern for both antennas in Eplane and Figure 5.16 shows the comparison of radiation pattern in H-plane. From
observation, the gain of the antenna increased up to 4 dB after the insertion of the
LHM. The 3 dB beam-width for E-plane become narrower from 900 to 560. In Hplane, the 3 dB beam-width narrows up to 450 from 830 to 380. The reduction of the 3
110
dB beam-width for both plane shows that the main beam has become narrower and
proves that the LHM focus the wave that pass through it.
E-plane
0
-35
330
30
-40
-45
-50
300
60
-55
-60
-65
-70
270
90
-35 -40 -45 -50 -55 -60 -65 -70
-70
-70 -65 -60 -55 -50 -45 -40 -35
-65
-60
-55
240
120
-50
-45
-40
210
150
-35
180
Single patch microstrip antenna
Single patch microstrip antenna incorporated with LHM
Figure 5.15: Radiation pattern in E-plane
H-plane
0
330
30
-40
-50
300
60
-60
-70
270
90
-80
-40
-50
-60
-70
-80
-70
-60
-50
-40
-70
-60
240
120
-50
210
-40
150
180
Single patch microstrip antenna
Single patch microstrip antenna incorporated with LHM
Figure 5.16: Radiation pattern in H-plane
111
As a whole, the gain of the antenna has increased in the presence of LHM due
to the focusing effect of the LHM. In order to illustrate the comparison between all
the parameters stated above, the performance are compared and presented as shown
in Table 5.4. Table 5.5 shows the comparison between the simulated and measured
results of the same type of antenna.
Table 5.4: Comparison of the antenna’s performance between single patch
microstrip antenna with and without LHM
Antenna parameters at
Single patch microstrip
Single patch microstrip
2.4 GHz
antenna antenna
antenna incorporated
with LHM
Return loss, S 11
-17.488 dB
-15.71 dB
Transmission coefficient,
-36.1 dB
-31.73 dB
2.9 %
4.98 %
(2.36 GHz – 2.43 GHz)
(2.35 GHz – 2.47 GHz)
-40 dBm
-36 dBm
S 21
Bandwidth
Gain
3dB
E-plane
900
560
beam-
H-plane
830
380
Cross
E-plane
35 dB
30 dB
polar
H-plane
29 dB
33 dB
21 dB
17 dB
width
isolation
Front to back lobe ratio
112
Table 5.5: Comparison between simulated and measured single patch microstrip
antenna incorporated with LHM
Antenna parameters at
Simulated single patch
Measured single patch
2.4 GHz
microstrip antenna
microstrip antenna
incorporated with LHM
incorporated with LHM
Return loss, S 11
-10 dB
-15.71 dB
Bandwidth
4%
4.98 %
(2.4 GHz – 2.5 GHz)
(2.35 GHz – 2.47 GHz)
4.22 dB
4 dB
Gain increment
3dB beam-
E-plane
48.50
560
width
H-plane
34.30
380
19.94 dB
17 dB
Front to back lobe ratio
The results for simulation and measurement in term of gain increment are
similar. The other results are different between simulation and measurement due to
the imperfect fabrication process.
H-plane
E-plane
0
0
330
330
30
1.0
0.8
0.8
0.6
300
270
0.8
0.6
0.4
0.6
300
60
0.4
0.4
0.2
0.2
90
0.0
1.0
0.2
0.0
0.2
0.4
0.6
0.8
1.0
270
0.8
0.6
0.4
0.2
90
0.0
0.2
0.4
0.6
0.8
1.0
0.2
0.4
0.4
120
0.6
240
120
0.6
0.8
0.8
210
60
0.0
1.0
0.2
240
30
1.0
1.0
150
210
1.0
150
180
180
Simulated single patch microstrip antenna incorporated with LHM
Measured single patch microstrip antenna incorporated with LHM
Simulated single patch microstrip antenna incorporated with LHM
Measured single patch microstrip antenna incorporated with LHM
(a)
(b)
Figure 5.17: (a) Comparison between simulated and measured radiation patterns in
E-plane and (b) Comparison between simulated and measured radiation patterns in
H-plane
113
Figure 5.17(a) and 5.17(b) shows the radiation pattern in E-plane and H-plane
for both simulated and measured. The shape of the simulated radiation pattern is
approximately similar to the measured radiation pattern.
5.6
Analysis and Discussion on Measurement of the Linear Polarized 2x2
Array Patch Microstrip Antenna Incorporated with LHM
Figure 5.18 shows the comparison of S 11 measured results and Figure 5.19
shows the comparison of S 21 measured results for both antennas with and without
LHM.
0
-5
S11, dB
-10
-15
-20
-25
-30
-35
1.4
1.6
1.8
2.0
2.2
2.4
2.6
2.8
3.0
3.2
Frequency, GHz
2x2 array patch microstrip antenna
2x2 array patch microstrip antenna incorporated with LHM
Figure 5.18: Return loss, S 11 of the linear polarized 2x2 Array Patch Microstrip
Antenna Incorporated with and without LHM
114
-20
-30
-40
S21, dB
-50
-60
-70
-80
-90
-100
1.4
1.6
1.8
2.0
2.2
2.4
2.6
2.8
3.0
3.2
Frequency, GHz
2x2 array patch microstrip antenna
2x2 array patch microstrip antenna incorporated with LHM
Figure 5.19: Transmission coefficient, S 21 of the linear polarized 2x2 Array Patch
Microstrip Antenna Incorporated with and without LHM
The resonant frequency is shifted to a higher frequency region from 2.43 GHz
to 2.48 GHz. The resonant frequency at 2.4 GHz before the placement of the LHM is
-10.2 dB and -7.6 dB after the LHM is placed in front of the antenna. Although the
return loss is a bit degraded, the result of S 21 shows an increment in magnitude up to
2 dB at 2.4 GHz. In other words, the focusing effect of the LHM is still operating at
2.4 GHz even though the return loss at that frequency is degraded. Above 2.52 GHz,
the Transmission coefficient went down to the lower value and acting similar as a
stop band after the incorporation of the LHM.
The radiation pattern of the antenna with and without LHM is measured in
order to prove the focusing effect of the LHM. Figure 5.20 shows the radiation
pattern for both antennas with and without LHM in E-plane at the frequency of 2.4
GHz. The 3 dB beam-width of the antenna incorporated with LHM is 480 similar
compared to the antenna without the LHM. The radiation in H-plane shown in Figure
5.21 shows the same effect after inserting the LHM onto the antenna. The 3 dB
beam-width become narrower from 440 to 380 after incorporating the LHM to the
antenna. The front to back ratio and front to side ratio shows approximate similar
value for both antennas with and without LHM. In order to make it more clearer,
comparison of the antenna’s parameter between linear polarized 2x2 array patch
microstrip antenna with and without LHM is presented in Table 5.6.
115
E-plane
0
330
-35
30
-40
-45
-50
300
60
-55
-60
-65
-70
270
90
-35 -40 -45 -50 -55 -60 -65 -70
-70
-70 -65 -60 -55 -50 -45 -40 -35
-65
-60
-55
240
120
-50
-45
-40
210
150
-35
180
2x2 array patch microstrip antenna
2x2 array patch microstrip antenna incorporated with LHM
Figure 5.20: Radiation pattern in E-plane
H-plane
0
330
-35
30
-40
-45
-50
300
60
-55
-60
-65
-70
270
-75
90
-35 -40 -45 -50 -55 -60 -65 -70 -75 -70 -65 -60 -55 -50 -45 -40 -35
-70
-65
-60
-55
240
120
-50
-45
-40
210
150
-35
180
2x2 array patch microstrip antenna
2x2 array patch microstrip antenna incorporated with LHM
Figure 5.21: Radiation pattern in H-plane
116
Table 5.6: Comparison of the antenna’s performance between linear polarized 2x2
array patch microstrip antenna with and without LHM
Antenna parameters at 2.4
Linear polarized 2x2
Linear polarized 2x2
GHz
array patch microstrip
array patch microstrip
antenna
antenna incorporated
with LHM
Return loss, S 11
-10.2 dB
-7.6 dB
Transmission coefficient,
-33 dB
-31 dB
2.3 %
2.8 %
(2.395 GHz – 2.45 GHz)
(2.45 GHz – 2.52 GHz)
-37 dBm
-35 dBm
S 21
Bandwidth
Gain
3dB beam-
E-plane
480
480
width
H-plane
440
380
Cross polar
E-plane
36 dB
43 dB
isolation
H-plane
32 dB
45 dB
Front to back lobe ratio
14 dB
16 dB
Table 5.7: Comparison between simulated and measured linear polarized 2x2 array
patch microstrip antenna incorporated with LHM
Antenna parameters at
Simulated linear
Measured linear polarized
2.4 GHz
polarized 2x2 array
2x2 array patch microstrip
patch microstrip antenna
antenna incorporated with
incorporated with LHM
LHM
Return loss, S 11
-10 dB
-7.6 dB
Bandwidth
2.4 %
2.8 %
(2.4 GHz – 2.46 GHz)
(2.45 GHz – 2.52 GHz)
2.44 dB
2 dB
Gain increment
3dB beam-
E-plane
42.10
480
width
H-plane
38.20
380
13.87 dB
16 dB
Front to back lobe ratio
117
From Table 5.7, the simulated return loss, S 11 , bandwidth and the 3 dB beamwidth are almost similar to the measured one. The increment of the gain is also alike
for simulated and measured and the other performances were different due to the
imperfect fabrication process.
Figure 5.22(a) and 5.22(b) shows the radiation pattern of the antenna
incorporated with LHM for both simulated and measured. Both results correlate well
with each other where the shapes of the radiation pattern for simulated and measured
are similar.
H-plane
E-plane
0
0
330
330
30
1.0
0.8
0.8
0.6
300
270
0.8
0.6
0.4
0.6
300
60
0.4
0.4
0.2
0.2
0.0
1.0
90
0.0
0.2
0.2
0.4
0.6
0.8
1.0
270
0.8
0.6
0.4
0.2
90
0.0
0.2
0.4
0.6
0.8
1.0
0.2
0.4
0.4
120
0.6
240
120
0.6
0.8
0.8
210
60
0.0
1.0
0.2
240
30
1.0
1.0
150
210
1.0
Simulated 2x2 array patch microstrip antenna incorporated with LHM
Measured 2x2 array patch microstrip antenna incorporated with LHM
(a)
150
180
180
Simulated 2x2 array patch microstrip antenna incorporated with LHM
Measured 2x2 array patch microstrip antenna incorporated with LHM
(b)
Figure 5.22: (a) Comparison between simulated and measured radiation patterns in
E-plane and (b) Comparison between simulated and measured radiation patterns in
H-plane
5.7
Analysis and Discussion on Measurement of the Circular Polarized 2x2
Array Patch Microstrip Antenna Incorporated with LHM
Figure 5.23 shows the measured return loss, S 11 for both antennas with and
without LHM. The circular polarized 2x2 array patch microstrip antenna operating
118
around 2.38 GHz to 2.61 GHz and note that the antenna incorporated with LHM, its
return loss is a bit above -10 dB (-9.96 dB) at 2.5 GHz as predicted in simulation. At
2.4 GHz, the antenna incorporated with LHM has a return loss of -11 dB.
0
-5
S11, dB
-10
-15
-20
-25
-30
-35
1.4
1.6
1.8
2.0
2.2
2.4
2.6
2.8
3.0
3.2
Frequency, GHz
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.23: Return losses, S 11 circular polarized 2x2 array patch microstrip antenna
incorporated with and without LHM
The Transmission coefficient, S 21 of both antennas with and without LHM in
co-polar position are shown in Figure 5.24. As observed, the magnitude value of S 21
increased at 2.2 GHz to 2.46 GHz with a maximum value at 2.37 GHz. The
increment is 3.97 dB from -35.35 dB to -31.38 dB at 2.4 GHz. Above 2.46 GHz, it
seem like the wave is unable to propagate through the LHM.
The Transmission coefficient, S 21 in cross-polar position of both antennas
with and without LHM are shown in Figure 5.25. From observation, the insertion
loss of both antennas has similar value. At 2.4 GHz, the value of S 21 decreased from
-36.36 dB to -38.37 dB. This is due to the structure of the LHM itself. As been
discussed in Chapter 2, the SRR can exhibit a plasmonic-type permeability frequency
which will introduce to negative permeability if the H-field is perpendicular to the
plane of the SRR and the CLS/wire can exhibit plasmonic-type permittivity
frequency which will introduce to negative permittivity if the E-field is parallel to the
CLS/wire. In this case, the E-field an H-field of the antenna itself is inversed in cross
polar position.
119
co-polar
-20
-30
-40
S21, dB
-50
-60
-70
-80
-90
-100
1.4
1.6
1.8
2.0
2.2
2.4
2.6
2.8
3.0
3.2
Frequency, GHz
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.24: Transmission coefficient, S 21 in co-polar position of the circular
polarized 2x2 array patch microstrip antenna incorporated with and without LHM
cross-polar
-30
S21, dB
-40
-50
-60
-70
-80
1.4
1.6
1.8
2.0
2.2
2.4
2.6
2.8
3.0
3.2
Frequency, GHz
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.25: Transmission coefficient, S 21 in cross-polar position of the circular
polarized 2x2 array patch microstrip antenna incorporated with and without LHM
120
E-plane
0
-35
330
30
-40
-45
300
60
-50
-55
-60
270
90
-35
-40
-45
-50
-55
-60
-60
-60
-55
-50
-45
-40
-35
-55
-50
240
120
-45
-40
210
150
-35
180
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.26: Co-polar radiation patterns in E-plane
H-plane
0
-35
330
30
-40
-45
-50
300
60
-55
-60
-65
270
-35
-40
-45
-50
-55
-60
-70
-70
-65 -70-70
90
-65
-60
-55
-50
-45
-40
-35
-65
-60
-55
240
120
-50
-45
-40
210
150
-35
180
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.27: Cross-polar radiation patterns in H-plane
121
E-plane
0
-40
330
30
-45
-50
-55
300
60
-60
-65
-70
270
90
-40
-45
-50
-55
-60
-65 -70
-70 -70
-65
-60
-55
-50
-45
-40
-65
-60
240
120
-55
-50
-45
210
150
-40
180
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.28: Co-polar radiation patterns in E-plane
H-plane
0
30
330
-45
-50
300
60
-55
-60
-65
270
90
-45
-50
-55
-60
-65
-65
-65
-60
-55
-50
-45
-60
-55
240
120
-50
-45
210
150
180
2x2 array patch circular polarized microstrip antenna
2x2 array patch circular polarized microstrip antenna incorporated with LHM
Figure 5.29: Cross-polar radiation patterns in H-plane
Figure 5.26 to Figure 5.29 shows the measured radiation pattern of the
antenna at 2.4 GHz. Figure 5.26 shows the co-polar radiation pattern in E-plane and
Figure 5.27 shows the co-polar radiation pattern in H-plane. The cross-polar
122
radiation pattern in E-plane is shown in Figure 5.28 and Figure 5.29 shows the crosspolar radiation pattern in H-plane. The gain of the antenna is increased up to 4 dB
after the insertion of the LHM. The shape of the radiation pattern observed in Figure
5.26 and Figure 5.27 shows that the beam becomes narrower and the 3 dB beamwidth goes to 420 from 620 in H-plane and 400 from 480 in E-plane. In the meantime,
the radiation pattern of the cross-polar for both planes shows a contrary result. The
gain of the antenna decreased down to 2 dB after the introduction of the LHM.
The cross-polar isolation of the antenna is 1 dB meanwhile the cross-polar
isolation for the antenna incorporated with LHM is 7 dB. In conclusion, the antenna
did not acted as a circular polarized antenna after the incorporation of the LHM as
the cross-polar isolation is larger then 3 dB.
Table 5.8 shows the comparison of all the antennas’ performance between the
circular polarized 2x2 array patch microstrip antenna with and without LHM.
Meanwhile, Table 5.9 shows the comparison of the simulated and measured circular
polarized 2x2 array patch microstrip antenna.
Table 5.8: Comparison of the antenna’s performance between circular polarized 2x2
array patch microstrip antenna with and without LHM
Antenna parameters at 2.4
Circular polarized
Circular polarized 2x2
GHz
2x2 array patch
array patch microstrip
microstrip antenna
antenna incorporated
with LHM
Return loss, S 11
-9.8 dB
-11 dB
Transmission coefficient, S 21
-35.35 dB
-31.38 dB
-36.36 dB
-38.37 dB
7.6 %
9.29 %
(2.4 GHz – 2.59
(2.38 GHz – 2.61 GHz)
(co-polar)
Transmission coefficient, S 21
(cross-polar)
Bandwidth
GHz)
Gain
-40 dBm
-36 dBm
123
3dB
E-plane
480
400
beam-
H-plane
620
420
Cross
E-plane
1 dB
7 dB
polar
H-plane
2 dB
7 dB
16 dB
19 dB
width
isolation
Front to back lobe ratio
Table 5.9: Comparison between simulated and measured circular polarized 2x2 array
patch microstrip antenna incorporated with LHM
Antenna parameters at
Simulated circular
Measured circular
2.4 GHz
polarized 2x2 array
polarized 2x2 array patch
patch microstrip antenna
microstrip antenna
incorporated with LHM
incorporated with LHM
Return loss, S 11
-8.32 dB
-11 dB
Bandwidth
2.2 %
9.29 %
(2.327 GHz – 2.38 GHz)
(2.38 GHz – 2.61 GHz)
2.8 %
(2.44 GHz – 2.51 GHz)
Gain increment
0.56 dB
4 dB
3dB
E-plane
47.50
400
beam-
H-plane
50.70
420
13.87 dB
19 dB
width
Front to back lobe ratio
The simulated result is a bit different from the measured results. The
bandwidth of the measured antenna is larger than the simulated antenna where two
bandwidths were combined to a single bandwidth after fabrication. The increment of
the gain is far beyond the expectation as in the simulation. As for the 3 dB beamwidth, the measured antenna is much narrower than the simulated once and the front
to back ratio was improved after the fabrication.
124
H-plane
E-plane
0
0
330
330
30
1.0
0.8
0.8
0.6
300
270
0.8
0.6
0.4
0.6
300
60
0.4
0.4
0.2
0.2
0.0
1.0
0.2
90
0.0
0.2
0.4
0.6
0.8
1.0
270
0.8
0.6
0.4
90
0.0
0.2
0.2
0.4
0.6
0.8
1.0
0.2
0.4
0.4
120
0.6
240
120
0.6
0.8
0.8
210
60
0.0
1.0
0.2
240
30
1.0
1.0
150
210
1.0
150
180
180
Simulated 2x2 array patch circular polarized microstrip antenna incorporated with LHM
Measured 2x2 array patch circular polarized microstrip antenna incorporated with LHM
(a)
Simulated 2x2 array patch circular polarized microstrip antenna incorporated with LHM
Measured 2x2 array patch circular polarized microstrip antenna incorporated with LHM
(b)
Figure 5.30: (a) Comparison between simulated and measured radiation patterns in
E-plane and (b) Comparison between simulated and measured radiation patterns in
H-plane
Figure 5.30 shows the radiation pattern for simulated and measured antenna
in both planes. As can be seen, the shape of the radiation pattern is similar for the
simulated and measured antennas.
5.8
Chapter Summary
The two linear polarized antennas which are single patch and 2x2 array patch
microstrip antenna shows an increment of gain and the 3dB beam-width also become
narrow. The circular polarized 2x2 array patch microstrip antenna shows an
increment of gain in E-co and H-co while the gain decreased in E-cross and H-cross.
The circular polarized 2x2 array patch microstrip antenna becomes a linear polarized
due to the increment of the value of cross polar isolation of the antenna. In a nutshell,
the LHM improved the gain of the antenna as well as focusing the wave propagating
through it. Besides, the 3dB beam-width had decreased for both E-Field and H-Field
respectively. Therefore, this again showed that the beam become directional due to
the focusing effect of the LHM. The LHM works properly for linear polarized
125
antenna while a LHM incorporated with a circular polarized antenna will change the
polarization to linear polarized and still improved the gain of the antenna.
CHAPTER 6
CONCLUSION
6.1
Overall Conclusion
An application oriented study of the state of the art of Left-Handed Metamaterial
(LHM), microstrip antenna and the integration of the LHM with the antennas to improve
the antenna performance has been carried out. A brief background of the project is
discussed providing the background, problem statement, objectives, scope, methodology
and anticipated outcome of the project.
Literature review as an importance key of the research has been carried out. The
peculiar properties of the LHM such as negative refraction index and backward wave
have been discussed thoroughly. The manner on how the structure exhibits the negative
ε and µ has also been discussed. A few key contribution papers related to the LHM and
antenna has been discussed and further analyzed.
The LHM is consisting of a single split ring resonator, MSRR which has been
modified, place in the middle of a pair capacitance loaded strip, CLS in planar form. The
MSRR has four slots in the middle of the structure which create wider range of negative
permittivity and permeability. . The MSRR exhibits negative permeability while the
CLS exhibits negative permittivity. The understanding of the behaviors of the LHM
characteristic is essential in order to design the structure in the required frequency.
127
Parametric studies on the parameters such as the lengths, the widths and the gaps of the
structure has been carried out and the results of S-parameters and the frequency range of
the negative ε and µ is analyzed. The configuration of the simulation of the LHM in
order to get the S-parameters has been proposed and the method and calculation to
determine the frequency range of the negative ε and µ from the S-parameters data has
been discussed.
The design of three types of antenna has been discussed. These three antennas
are single patch microstrip antenna, linear polarized 2x2 array patch microstrip antenna
and circular polarized 2x2 array patch microstrip antenna. The technique to design the
power divider and quarter-wave transformer are also been discussed.
The integration of the LHM with these three types of antenna has been carried
out through simulation and measurement. The introduction of the LHM to the antenna
certainly effects and alters the return loss of the antenna. The results correlate well with
those simulated ones and the measured. Realistically, the LHM structure is fabricated
using etching technique of a metallically clad FR4 dielectric board.
The comparison in the results between the microstrip antenna and the microstrip
antenna incorporated with LHM has been carried out. The integration improved the
antenna’s gain and produced a more directional radiation pattern beam. Some of the
integration improved the bandwidth of the antenna. The concept of the negative
refraction index of the LHM has been proved where it act as a lens resulting the gain of
the antenna increased and the half power beam-width become narrow.
128
6.2
Key Contribution
The latest development of the LHM especially towards the antenna application
has been previewed. The unique properties of the LHM have been understood and the
way the LHM operate and exhibit the negative ε and µ has been studied.
The technique to simulate the LHM unit cell structure in order to collect the Sparameters data has been proposed. The calculation method of the ε and µ through
extraction from the S-parameters has been studied and mastered. Different configuration
of the LHM has been simulated and the best configuration is selected for further studies.
Parametric studies has been carried out and by varying the parameters such as lengths,
widths and gaps of the LHM, they produced different frequency range of negative ε and
µ.
The introduction of the air gaps between each LHM unit cells and the design of
the LHM in a planar form reduces the cost of fabrication and decrease the fabrication
time.
With this, all the contribution factors leads to design of the antenna incorporated
with LHM. The simulation of the single patch microstrip antenna, linear polarized 2x2
array patch microstrip antenna and circular polarized 2x2 array patch microstrip antenna
incorporated with LHM has been presented and the measurement to validate the
simulation has been carried out.
The integration of the LHM to the antennas through simulation improves the
antenna’s gain, return loss and the shape of the antenna’s radiation pattern become
directional. The fabrication and measurement of these antennas proves the simulation is
accurate where the results correlate well with each others.
129
6.3
Future Research
The LHM structure will be further investigated. A new design of LHM that could
exhibit both negative for ε and µ would be interesting to be explored. The new design
should be smaller and less bulky and with this feature, the fabrication cost can be
reduced.
A new approach to apply the LHM in different situation should be studied. By
implementing the stop band and pass band of the LHM, it can be use as a filter for the
antenna. It can be used in UWB antenna where a certain band needed to be stop while
others are required.
A beam shift can be produced using the LHM and a study should be done in this
area of work. Implementing diode to the LHM with specific configuration and switching
it in a certain part LHM would alter the beam of the radiation pattern of the antenna.
The antenna incorporated with LHM is suitable for point to point WLAN and
WiMAX and the LHM should be designed at these bands. This will increase the market
value of the LHM.
130
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[1]
George
V.
eleftheriades
and
Keith
G.
Balmain,
Negative-Refraction
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[2]
Constantine A. Balanis, Antenna theory, 3nd Ed., Hoboken, N. J.: Wiley, 1998.
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Nader Engheta and Richard W. Ziolkowski Metamaterials Physics and
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[6]
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APPENDIX A
List of Publications
Patent intellectual properties (IP)
1.
LEFT HANDED METAMATERIAL
Mohamad Kamal A. Rahim, PI 2009 2983
2.
LEFT HANDED METAMATERIAL INCORPARATED WITH
MICROTRIP ANTENNA
Mohamad Kamal A. Rahim, PI 2009 2919
Published Journal
1.
MICROSTRIP ANTENNA’S GAIN ENHANCEMENT USING LEFT
HANDED METAMATERIAL STRUCTURE - Huda A. Majid, Mohamad
Kamal A. Rahim and Thelaha Masri, Progress In Electromagnetics Research
M, Vol. 8, 235 – 247, 2009.
Published Paper
1.
INVESTIGATION
OF
LEFT
HANDED
METAMATERIAL
IN
MICROSTRIP ANTENNA APPLICATION - Huda A. Majid and
Mohamad
Kamal
A.
Rahim,
Asia
Pasific
Conference
on
Applied
Electromagnetics (APACE 2007), Renaissance Melaka Hotel, Melaka,
Malaysia.
132
2.
LEFT
HANDED
METAMATERIAL
INCORPORATED
WITH
MICROSTRIP ANTENNA AT 6 GHZ – Huda A. Majid, Mohamad Kamal
A. Rahim, Thelaha Masri and Osman Ayop, International Symposium on
Antennas and Propagation (ISAP 2008), Taipei International Convention
Center, Taipei, Taiwan.
3.
LEFT HANDED MATEMATERIAL DESIGN FOR MICROSTRIP
ANTENNA APPLICATION – Huda A. Majid, Mohamad Kamal A. Rahim
and Thelaha Masri, IEEE International RF and Microwave Conference (RFM
2008), Kuala Lumpur, Malaysia.
4.
MEASUREMENT OF THE MICROSTRIP ANTENNA INCORPORATED
WITH LEFT HANDED METAMATERIAL – Huda A. Majid, Mohamad
Kamal A. Rahim, Thelaha Masri and Mohd Nazri A. Karim, Student Conference
on Research and Development (SCOReD 2008), Universiti Teknologi Malaysia,
Johor, Malaysia.
5.
LEFT
HANDED
METAMATERIAL
INCORPORATED
WITH
CIRCULAR POLARIZED MICROSTRIP ANTENNA - Huda A. Majid,
Mohamad Kamal A. Rahim and Thelaha Masri, International Symposium on
Antennas and Propagation (ISAP 2009), 20-23 October 2009, Imperial Queen’s
Park Hotel, Bangkok, Thailand.
133
APPENDIX B
Example of the Calculation on the modified NRW method using MathCAD
f :=
A :=
0
0
2
1
2.005
2
2.01
3
2.015
4
2.02
5
2.025
6
2.03
7
8
B :=
0
C :=
0
0
0
-4.471
0
138.649
0
-2.074
1
-4.48
1
138.345
1
-2.078
2
-4.488
2
138.06
2
-2.08
3
-4.496
3
137.794
3
-2.08
4
-4.503
4
137.546
4
-2.079
5
-4.508
5
137.315
5
-2.075
6
-4.511
6
137.098
6
-2.07
2.035
7
-4.512
7
136.892
7
-2.065
2.04
8
-4.51
8
136.693
8
-2.058
9
2.045
9
-4.506
9
136.498
9
-2.052
10
2.05
10
-4.498
10
136.303
10
-2.045
11
2.055
11
-4.488
11
136.105
11
-2.04
12
2.06
12
-4.475
12
135.9
12
-2.035
13
2.065
13
-4.46
13
135.685
13
-2.031
14
2.07
14
-4.443
14
135.457
14
-2.029
15
...
15
...
15
...
15
...
A is magnitude of S11
B is phase of S11
C is magnitude of S21
D is phase of S21
f is frequency
n := 1 , 2 .. 1000
134
B
n
X :=
n
180
⋅π
( n)
realX := A ⋅ cos X
n
n
( n)
imageX := A ⋅ sin X
n
n
0
realX = 0
1
2
0
3.347
imageX = 0
1
-2.977
...
2
...
0
0
imageXX := imageX⋅ ( 0 + i)
0
0
imageXX = 1
-2.977i
2
-3i
3
...
0
D
Y :=
n
n
180
⋅π
( n)
realY := C ⋅ cos Y
n
n
n
n
0
realY = 0
1
2
( n)
imageY := C ⋅ sin Y
0
1.42
imageY = 0
1
1.518
...
2
...
0
imageYY := imageY⋅ ( 0 + i)
0
0
imageYY = 1
0
1.518i
2
1.514i
3
...
0
135
s11 := realX + imageXX
s21 := realY + imageYY
0
0
0
0
0
0
s11 = 1
3.347-2.977i
s21 = 1
1.42+1.518i
2
3.338-3i
2
1.427+1.514i
3
...
3
...
h := f ⋅ GHz
d := 1.635mm
εο := 8.854187817e
− 12
µο := 1.256637061e
Note :
d = thickness of slab
c = speed of light
w = radian frequency
8m
c = 2.998 × 10
−6
w := 2⋅ π ⋅ h
s
εr = relative permittivity
μr = relative permeability
v1 := s21 + s11
v2 := s21 − s11
µr :=
n
2⋅ (1 − v2n) ⋅ c
(
w ⋅ d ⋅ i⋅ 1 + v2
n
εr :=
)
n
n
2⋅ c
2
n
w ⋅ d ⋅ i 1 + v1
n
n
0
Re( µr ) = 0
1
1 − v1
⋅
0
-12.423
Re( εr ) = 0
1
2.401
...
2
...
0
0
136
APPENDIX C
Refraction Index
4
3
2
1
Refraction
index, εr & µr
0
-1
-2
-3
-4
2.0
2.2
2.4
2.6
2.8
3.0
Frequency, GHz
Permittivity
Permeability
Refractive Index
Figure C0: Value of ε r , µ r and refractive index for unit cell with 8 mm air gap
Refraction index, n
n = -0.4 at 2.45 GHz
n = -0.67 at 2.4 GHz
n = -1.0 at 2.36 GHz
n = -1.7 at 2.3 GHz
Effect of the refraction index, n toward the gain of the microstrip antenna
Figure C1, C2, C3 and C4 show the radiation pattern at 4 different frequencies
with different value of refraction index occurs. Table C1 shows the comparison of those
radiation patterns.
137
Figure C1: Radiation pattern at 2.3 GHz
Figure C2: Radiation pattern at 2.36 GHz
Figure C3: Radiation pattern at 2.4 GHz
138
Figure C4: Radiation pattern at 2.45 GHz
As observed, the highest gain occurs at 2.4 GHz where the refraction index is -1.
At 2.4 GHz, the gain is considered high where the refraction index is -0.67. At the
refraction index of -1.7, the gain is also high where it is operating at 2.30 GHz.
Table C1: Gain comparison at different frequencies with different value of refraction
index.
Frequency
Gain
Refraction index, n
2.30 GHz
10.85 dBi
-1.7
2.36 GHz
11.57 dBi
-1.0
2.40 GHz
11.50 dBi
-0.67
2.45 GHz
6.60 dBi
-0.4
Finding:
•
Tolerance for refraction index for LHM that could increase the gain of the
antenna is;
–
•
+/- 0.4
So, the refraction index, n of the LHM should be;
–
n = -1 + 0.4
However, in measurement the highest gain occur at 2.4 GHz. The shifting of the
operating frequency is due to inconsistency of the dielectric constant value of the
139
substrate board (FR4) used. The inconsistency usually occurs at higher frequency where
it shifts the frequency to higher region.
140
APPENDIX D
Scaling the Dimension of the Left Handed Metamaterial Structure
One of the ways to change or shift the resonant frequency and also the value of ε r
and µ r is scaling the whole structure. The initial dimension of the structure has been
elaborated in detail in Chapter 3. Table D1 has been tabulated in order to acquire a range
of negative ε r and negative µ r from 1 GHz to 5 GHz.
Table D1: Correlation between the scaling factor and the frequency range of negative
permittivity, ε r and negative permeability, µ r
Scaling factor
Frequency range of negative permittivity, ε r &
permeability, µ r
0.5
4.60 GHz – 5.00 GHz
0.6
3.77 GHz – 4.28 GHz
0.7
3.20 GHz – 3.69 GHz
0.8
2.76 GHz – 3.30 GHz
1
2.15 GHz – 2.70 GHz
1.3
1.58 GHz – 2.12 GHz
1.5
1.34 GHz – 1.47 GHz
141
APPENDIX E
Wet Etching Process
The first step in the fabrication process is wet etching technique. The material
used to fabricate the LHM structure and microstrip antennas is FR4 board which has
substrate thickness of 1.6 mm, copper thickness of 0.035 mm, relative permittivity of 4.6
and tangential loss of 0.019. Firstly, the cover of the photo resist microstrip board is
being removed. After that, the transparent mask is placed on top of the microstrip
antennas and LHM structure layout area. Lastly, mask and microstrip board are exposed
to ultra violet (UV) light where the layer were not expose to UV light become
polymerized or soluble (hard) as shown in Figure E1. The region which is not soluble is
removed by an acid call developer as shown in Figure E2. This process is done in the
dark room. Next, the region which is not exposed is removed by using a strong chemical
acid as shown in Figure E3.
Figure E1: UV- light generator
142
Figure E2: Acid / developer (to remove first layer)
Figure E3: Chemical acid (to remove second layer)
The etching process is very important to obtain good structure for LHM and
microstrip antennas from the fabrication. Hence, the procedures must be followed
accordingly to ensure that the output from fabrication is as same as defined in the
simulation. Good transparent layout, appropriate time to expose the structure to the UV
light and the quality of the developer and chemical play important roles to obtain good
fabrication result.