a Low Power Mixer 3 V Receiver IF Subsystem AD61009

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FEATURES
Complete Receiver-on-a-Chip: Monoceiver® Mixer
–15 dBm 1 dB Compression Point
–8 dBm Input Third Order Intercept
500 MHz RF and LO Bandwidths
Linear IF Amplifier
Linear-in-dB Gain Control
Manual Gain Control
Quadrature Demodulator
On-Board Phase-Locked Quadrature Oscillator
Demodulates IFs from 1 MHz to 12 MHz
Can Also Demodulate AM, CW, SSB
Low Power
25 mW at 3 V
CMOS Compatible Power-Down
APPLICATIONS
GSM and TETRA Receivers
Satellite Terminals
Battery-Powered Communications Receivers
GENERAL DESCRIPTION
The AD61009 is a 3 V low power receiver IF subsystem for operation at input frequencies as high as 500 MHz and IFs from
400 kHz to 12 MHz. It consists of a mixer, IF amplifiers, I and
Q demodulators, a phase-locked quadrature oscillator, and a
biasing system with external power-down.
The AD61009’s low noise, high intercept mixer is a doublybalanced Gilbert cell type. It has a nominal –15 dBm input
referred 1 dB compression point and a –8 dBm input referred
third-order intercept. The mixer section of the AD61009 also
includes a local oscillator (LO) preamplifier, which lowers the
required LO drive to –16 dBm.
Low Power Mixer
3 V Receiver IF Subsystem
AD61009
PIN CONFIGURATION
20-Lead SSOP
(RS Suffix)
FDIN 1
20 VPS1
COM1 2
19 FLTR
PRUP 3
18 IOUT
LOIP 4
17 QOUT
AD61009 16 VPS2
TOP VIEW 15 DMIP
(Not to Scale)
14 IFOP
GREF 7
RFLO 5
RFHI 6
MXOP 8
13 COM2
VMID 9
12 GAIN
IFHI 10
11 IFLO
A quadrature VCO phase-locked to the IF drives the I and Q
demodulators. The I and Q demodulators can also demodulate AM; when the AD61009’s quadrature VCO is phase locked
to the received signal, the in-phase demodulator becomes a
synchronous product detector for AM. The VCO can also be
phase-locked to an external beat-frequency oscillator (BFO),
and the demodulator serves as a product detector for CW or
SSB reception. Finally, the AD61009 can be used to demodulate BPSK using an external Costas Loop for carrier recovery.
In MGC operation, the AD61009 accepts an external gaincontrol voltage input from an external AGC detector or a DAC.
Monoceiver is a registered trademark of Analog Devices, Inc.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 2001
AD61009–SPECIFICATIONS (@ T = 25ⴗC, Supply = 3.0 V, IF = 10.7 MHz, unless otherwise noted)
A
Model
Conditions
Min
AD61009ARS
Typ
Max
Unit
DYNAMIC PERFORMANCE
MIXER
Maximum RF and LO Frequency Range
Maximum Mixer Input Voltage
Input 1 dB Compression Point
Input Third-Order Intercept
Noise Figure
Maximum Output Voltage at MXOP
Mixer Output Bandwidth at MXOP
LO Drive Level
LO Input Impedance
Isolation, RF to IF
Isolation, LO to IF
Isolation, LO to RF
Isolation, IF to RF
IF AMPLIFIERS
Noise Figure
Input 1 dB Compression Point
Output Third-Order Intercept
Maximum IF Output Voltage at IFOP
Output Resistance at IFOP
Bandwidth
GAIN CONTROL
Gain Control Range
Gain Scaling
Gain Scaling Accuracy
Bias Current at GAIN
Bias Current at GREF
Input Resistance at GAIN, GREF
I AND Q DEMODULATORS
Required DC Bias at DMIP
Input Resistance at DMIP
Input Bias Current at DMIP
Maximum Input Voltage
Amplitude Balance
Quadrature Error
Phase Noise in Degrees
Demodulation Gain
Maximum Output Voltage
Output Offset Voltage
Output Bandwidth
PLL
Required DC Bias at FDIN
Input Resistance at FDIN
Input Bias Current at FDIN
Frequency Range
Required Input Drive Level
Acquisition Time to ± 3°
POWER-DOWN INTERFACE
Logical Threshold
Input Current for Logical High
Turn-On Response Time
Standby Current
For Conversion Gain > 20 dB
For Linear Operation; Between RFHI and RFLO
RF Input Terminated in 50 Ω
RF Input Terminated in 50 Ω
Matched Input, Max Gain, f = 83 MHz, IF = 10.7 MHz
Matched Input, Max Gain, f = 144 MHz, IF = 10.7 MHz
ZIF = 165 Ω, at Input Compression
–3 dB, ZIF = 165 Ω
Mixer LO Input Terminated in 50 Ω
LOIP to VMID
RF = 240 MHz, IF = 10.7 MHz, LO = 229.3 MHz
RF = 240 MHz, IF = 10.7 MHz, LO = 229.3 MHz
RF = 240 MHz, IF = 10.7 MHz, LO = 229.3 MHz
RF = 240 MHz, IF = 10.7 MHz, LO = 229.3 MHz
500
± 54
–15
–5
14
12
± 1.3
45
–16
1
30
20
40
70
MHz
mV
dBm
dBm
dB
dB
V
MHz
dBm
kΩ
dB
dB
dB
dB
Max Gain, f = 10.7 MHz
IF = 10.7 MHz
IF = 10.7 MHz
ZIF = 600 Ω
From IFOP to VMID
–3 dB at IFOP, Max Gain
17
–15
18
± 560
15
45
dB
dBm
dBm
mV
Ω
MHz
90
20
75/VR
±1
5
1
1
dB
mV/dB
dB/V
dB
µA
µA
MΩ
VPOS/2
50
2
±150
±75
± 0.2
–1.2
–100
18
±1.23
10
1.5
V dc
kΩ
µA
mV
mV
dB
Degrees
dBc/Hz
dB
V
mV
MHz
(See Figures 10 and 11)
Mixer + IF Section, GREF to 1.5 V
GREF to 1.5 V
GREF to General Reference Voltage VR
GREF to 1.5 V, 80 dB Span
From DMIP to VMID
IF > 3 MHz
IF ≤ 3 MHz
IF = 10.7 MHz, Outputs at 600 mV p-p, F = 100 kHz
IF = 10.7 MHz, Outputs at 600 mV p-p, F = 100 kHz
IF = 10.7 MHz, F = 10 kHz
Sine Wave Input, Baseband Output
RL ≥ 20 kΩ
Measured from IOUT, QOUT to VMID
Sine Wave Input, Baseband Output
17.4
–100
From FDIN to VMID
Sine Wave Input at Pin 1
IF = 10.7 MHz
For Power Up on Logical High
To PLL Locked
POWER SUPPLY
Supply Range
Supply Current
OPERATING TEMPERATURE
TMIN to TMAX
–3.5
+100
V dc
kΩ
nA
MHz
mV
µs
2
75
16.5
550
V dc
µA
µs
µA
8.5
–25
–40
18.8
VPOS/2
50
200
1.0 to 12
400
16.5
2.85
Operation to 2.85 V Minimum Supply Voltage
Operation to 4.5 V Minimum Supply Voltage
+1.5
5.5
12.5
V
mA
+85
+85
°C
°C
Specifications subject to change without notice.
–2–
REV. 0
AD61009
ABSOLUTE MAXIMUM RATINGS 1
NOTES
1
Stresses above those listed under Absolute Maximum Rating may cause permanent
damage to the device. This is a stress rating only; functional operation of the device
at these or any other conditions above those indicated in the operational section
of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
2
Thermal Characteristics: 20-lead SSOP Package: θJA = 126°C/W.
Supply Voltage VPS1, VPS2 to COM1, COM2 . . . . . . . 5.5 V
Internal Power Dissipation2 . . . . . . . . . . . . . . . . . . . . 600 mW
2.7 V to 5.5 V Operating Temperature Range
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –25°C to +85°C
4.5 V to 5.5 V Operating Temperature Range
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . . 300°C
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD61009 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
ORDERING GUIDE
Model
Temperature Range
Package Description
Package Option
AD61009ARS
–25°C to +85°C for 2.7 V to 5.5 V
Operation; –40°C to +85°C for 4.5 V
to 5.5 V Operation
20-Lead Plastic SSOP
RS-20
REV. 0
–3–
AD61009
PIN FUNCTION DESCRIPTIONS
Pin
Mnemonic
Reads
Function
1
FDIN
Frequency Detector Input
2
3
COM1
PRUP
Common #1
Power-Up Input
4
LOIP
Local Oscillator Input
5
6
7
8
RFLO
RFHI
GREF
MXOP
RF “Low” Input
RF “High” Input
Gain Reference Input
Mixer Output
9
10
11
12
VMID
IFHI
IFLO
GAIN
Midsupply Bias Voltage
IF “High” Input
IF “Low” Voltage
Gain Control Input
13
14
COM2
IFOP
Common #2
IF Output
15
DMIP
Demodulator Input
16
17
VPS2
QOUT
VPOS Supply #2
Quadrature Output
18
IOUT
In-Phase Output
19
20
FLTR
VPS1
PLL Loop Filter
VPOS Supply #1
PLL input for I/Q demodulator quadrature oscillator, ± 400 mV
drive required from external oscillator. Must be biased at VP/2.
Supply common for RF front end and main bias.
3 V/5 V CMOS compatible power-up control; logical high =
powered-up; max input level = VPS1 = VPS2.
LO input, ac coupled ± 54 mV LO input required (–16 dBm for
50 Ω input termination).
Usually connected to ac ground.
AC coupled, ± 56 mV, max RF input for linear operation.
High impedance input, typically 1.5 V, sets gain scaling.
High impedance, single-sided current output, ± 1.3 V max voltage
output (± 6 mA max current output).
Output of the midsupply bias generator (VMID = VPOS/2).
AC coupled IF input, ± 56 mV max input for linear operation.
Reference node for IF input; auto-offset null.
High impedance input, 0 V–2 V using 3 V supply, max gain at
V = 0.
Supply common for IF stages and demodulator.
Low impedance, single-sided voltage output, 5 dBm (± 560 mV)
max.
Signal input to I and Q demodulators ± 150 mV max input at IF
> 3 MHz for linear operation; ± 75 mV max input at IF < 3 MHz
for linear operation. Must be biased at VP/2.
Supply to high-level IF, PLL, and demodulators.
Low impedance Q baseband output; ± 1.23 V full scale in 20 kΩ
min load; ac coupled.
Low impedance I baseband output; ± 1.23 V full scale in 20 kΩ
min load; ac coupled.
Series RC PLL Loop filter, connected to ground.
Supply to mixer, low level IF, PLL, and gain control.
PIN CONNECTION
20-Lead SSOP (RS-20)
FDIN 1
20 VPS1
COM1 2
19 FLTR
PRUP 3
18 IOUT
LOIP 4
17 QOUT
AD61009 16 VPS2
TOP VIEW 15 DMIP
(Not to Scale)
14 IFOP
GREF 7
RFLO 5
RFHI 6
MXOP 8
13 COM2
VMID 9
12 GAIN
IFHI 10
11 IFLO
–4–
REV. 0
Typical Performance Characteristics–AD61009
50⍀
HP8764B
HP8656B
IEEE
0
RF_OUT
SYNTHESIZER
1
HP8656B
0
S1
RF_OUT
IEEE
S0
1
50⍀
SYNTHESIZER
V
RFHI
0
1
0
X
R
S1
IFHI
P6205
TEK1105
OUT IN1
OUT1
FET PROBE
IFOP
X10
VPOS
0
VNEG
IEEE
IN2
SPOS
DMIP
SNEG
DCPS
FDIN
HP34401A
VPOS
I
R5
PRUP
1k⍀
GAIN
DP8200
VPOS
IEEE
HP8765B
C
S0 V
S1
PLL
QOUT
LO
DMM
V
1
OUT2
PROBE
SUPPLY
IOUT
HI
CPIB
1
50⍀
LOIP
HP6633A
S0
MXOP
L
HP8656B
RF_OUT
IEEE
SYNTHESIZER
HP8764B
50⍀
CHARACTERIZATION
BOARD
VNEG
SPOS
0
SNEG
1
V REF
BIAS
HP8765B
C
S0 V
S1
TPC 1. Mixer/Amplifier Test Set
HP8720C
PORT_1
IEEE_488
PORT_2
NETWORK AN
0
HP346B
28V
CHARACTERIZATION
BOARD
HP8765B
HP8765B
NOISE SOURCE
C
RFHI
X
R
S0 V S1
MXOP
C
1
S1 V S0
L
HP8656B
RF_OUT
IEEE
SYNTHESIZER
LOIP
IFHI
IFOP
IOUT
DMIP
FDIN
PLL
QOUT
HP6633A
VPOS
IEEE
VPOS
VNEG
PRUP
SPOS
GAIN
BIAS
SNEG
DCPS
DP8200
VPOS
VNEG
IEEE
SPOS
V REF
SNEG
TPC 2. Mixer Noise Figure Test Set
REV. 0
0
50⍀
HP8970A
NOISE
1
–5–
RF_IN
28V_OUT
NOISE FIGURE METER
HP8594E
RF_IN
IEEE
SPEC AN
AD61009
CHARACTERIZATION
BOARD
RFHI
X
R
MXOP
L
LOIP
HP346B
28V
P6205
IFHI
NOISE
IFOP
X10
FET
OUT IN1
PROBE
TEK1103
OUT1
28V_OUT
NOISE FIGURE METER
NOISE SOURCE
IN2
OUT2
PROBE SUPPLY
DMIP
FDIN
HP8970A
RF_IN
IOUT
PLL
QOUT
HP6633A
VPOS
VPOS
IEEE
VNEG
PRUP
SPOS
GAIN
BIAS
SNEG
DCPS
DP8200
VPOS
VNEG
IEEE
SPOS
V REF
SNEG
TPC 3. IF Amp Noise Figure Test Set
CHARACTERIZATION
BOARD
50⍀
HP8764B
RFHI
0
HP8656B
IEEE
X
R
MXOP
L
1
0
RF_OUT
SYNTHESIZER
S0
LOIP
S1
50⍀
1
IFHI
IFOP
DMIP
IOUT
V
HP3326A
DCFM
OUTPUT_1
IEEE
OUTPUT_2
DUAL SYNTHESIZER
P6205
FDIN
QOUT
VPOS
IEEE
VPOS
VNEG
PRUP
SPOS
GAIN
BIAS
OUT IN1
1103
OUT1
FET PROBE
PLL
HP6633A
X10
P6205
OUT IN2
OUT2
X10
PROBE
FET PROBE
SUPPLY
0
HP8765B
1
HP8765B 0
C
S0 V S1
C
HP8694E
RF_IN
IEEE
1
SPEC AN
S1 V S0
HP54120
CH1
SNEG
CH2
DCPS
CH3
DP8200
CH4
VPOS
TRIG
VNEG
IEEE
IEEE_488
DIGITAL
OSCILLOSCOPE
SPOS
SNEG
V REF
TPC 4. PLL/Demodulator Test Set
–6–
REV. 0
AD61009
CHARACTERIZATION
BOARD
RFHI
MXOP
R
X
L
LOIP
HP6633A
VPOS
IFHI
IFOP
VNEG
IEEE
SPOS
SNEG
DCPS
DP8200
FDIN
VNEG
IEEE
IOUT
DMIP
VPOS
PLL
QOUT
SPOS
SNEG
V REF
VPOS
R1
499k⍀
HP34401A
HI
GAIN
LO
I
GPIB
DMM
BIAS
PRUP
TPC 5. GAIN Pin Bias Test Set
CHARACTERIZATION
BOARD
RFHI
MXOP
R
X
L
LOIP
HP6633A
VPOS
IFHI
IFOP
DMIP
IOUT
VNEG
IEEE
SPOS
SNEG
DCPS
DP8200
VPOS
FDIN
VNEG
IEEE
PLL
QOUT
SPOS
SNEG
VPOS
V REF
R1
499k⍀
HP34401A
BIAS
PRUP
HI
GPIB
DMM
GAIN
LO
I
TPC 6. Demodulator Bias Test Set
CHARACTERIZATION
BOARD
HP3325B
IEEE
RFHI
RF_OUT
SYNTHESIZER
MXOP
R
X
L
LOIP
HP6633A
VPOS
HP8594E
IFHI
VNEG
IEEE
IFOP
SPOS
SPEC AN
SNEG
DCPS
HP6633A
DMIP
VPOS
FDIN
VNEG
IEEE
IOUT
PLL
QOUT
SPOS
SNEG
DCPS
VPOS
R1
10k⍀
HP34401A
GAIN
LO
DMM
BIAS
PRUP
HI
GPIB
I
TPC 7. Power-Up Threshold Test Set
REV. 0
RF_IN
–7–
IEEE
AD61009
CHARACTERIZATION
BOARD
RFHI
MXOP
R
X
L
LOIP
IFHI
FL6082A
RF_OUT
IEEE
MOD_OUT
X10
IOUT
DMIP
FDIN
1103
P6205
IFOP
50⍀
OUT IN1
CH1
CH2
CH3
CH4
TRIG
IEEE_488
DIGITAL
OSCILLOSCOPE
PLL
QOUT
HP6633A
VPOS
VPOS
IEEE
HP54120
OUT1
FET PROBE
P6205
OUT IN2
OUT2
X10
PROBE SUPPLY
FET PROBE
VNEG
PRUP
SPOS
GAIN
NOTE: MUST BE 3 RESISTOR POWER DIVIDER
BIAS
SNEG
DCPS
DP8200
VPOS
VNEG
IEEE
SPOS
V REF
SNEG
HP8112
IEEE
PULSE_OUT
PULSE GENERATOR
TPC 8. Power-Up Test Set
CHARACTERIZATION
BOARD
RFHI
MXOP
R
X
L
LOIP
HP8656B
IEEE
RF_OUT
IFHI
P6205
IFOP
X10
R1
1k⍀
SYNTHESIZER
HP8594E
1103
OUT IN1
OUT1
FET PROBE
RF_IN
IEEE
SPEC AN
IN2
OUT2
PROBE SUPPLY
IOUT
DMIP
FDIN
PLL
QOUT
HP6633A
VPOS
IEEE
VPOS
VNEG
PRUP
SPOS
GAIN
BIAS
SNEG
DCPS
TPC 9. IF Output Impedance Test Set
–8–
REV. 0
AD61009
CHARACTERIZATION
BOARD
RFHI
MXOP
R
X
L
LOIP
IFHI
IFOP
20⍀
dB
HP54120
P6205
FL6082A
DMIP
RF_OUT
IEEE
IOUT
FDIN
X10
QOUT
X10
HP6633A
VPOS
VPOS
IEEE
VNEG
PRUP
SPOS
GAIN
OUT IN1
OUT1
CH1
CH2
FET PROBE
P6205
PLL
MOD_OUT
1103
FET PROBE
BIAS
CH3
OUT IN2
OUT2
CH4
TRIG
PROBE SUPPLY
IEEE_488
DIGITAL
OSCILLOSCOPE
SNEG
DCPS
DP8200
VPOS
VNEG
IEEE
SPOS
V REF
SNEG
TPC 10. PLL Settling Time Test Set
CHARACTERIZATION
BOARD
RFHI
HP3325B
IEEE
MXOP
R
X
L
LOIP
RF_OUT
SYNTHESIZER
IFHI
IFOP
DMIP
IOUT
HP3326
DCFM
OUTPUT_1
IEEE
OUTPUT_2
DUAL SYNTHESIZER
P6205
FDIN
PLL
QOUT
VPOS
VPOS
VNEG
PRUP
SPOS
GAIN
1103
OUT IN1
0
OUT IN2
FET PROBE
BIAS
OUT1
FET PROBE
P6205
X10
HP6633A
IEEE
X10
OUT2
PROBE SUPPLY
SNEG
DCPS
DP8200
VPOS
VNEG
IEEE
SPOS
SNEG
V REF
TPC 11. Quadrature Accuracy Test Set
REV. 0
–9–
HP8765B
HP8694E
1
C
S0 V S1
RF_IN
IEEE
SPEC AN
AD61009
VPOS
C15
0.1␮F
GND
C11
10nF
FDIN
4.99k⍀
R10
0.1␮F
C13
0.1␮F
C1
R8
51.1⍀
0⍀
R12
PRUP
C10
1nF
LOIP
R7
51.1⍀
C9
1nF
RFHI
MXOP
*
IFHI
C16
1nF
R6
51.1⍀
R13
301⍀
1 FDIN
VPS1 20
2 COM1
FLTR 19
3 PRUP
IOUT 18
4 LOIP
QOUT 17
AD61009
5 RFLO
6 RFHI
DMIP 15
7 GREF
IFOP 14
8 MXOP
COM2 13
9 VMID
GAIN 12
10 IFHI
R14
54.9⍀
R5
332⍀
R9
51.1⍀
VPS2 16
R1
1k⍀
IOUT
0.1␮F
C2
*
QOUT
*
R2
316⍀
IFOP
*
IFLO 11
C6
0.1␮F
C7
1nF
C3
10nF
C8
0.1␮F
C5
1nF
GAIN
*
DMIP
*
0.1␮F
NOTE: CONNECTIONS MARKED
* ARE DC COUPLED.
TPC 12. Characterization Board
–10–
REV. 0
AD61009
30
20
19
VGAIN = 0.3V
25
18
16
20
CONVERSION GAIN – dB
SSB NF – dB
17
VPOS = 5V, IF = 20MHz
15
VPOS = 3V, IF = 20MHz
14
13
VGAIN = 0.6V
15
10
VGAIN = 1.2V
5
VGAIN = 1.8V
0
VGAIN = 2.4V
12
11
10
50
–5
VPOS = 5V, IF = 10MHz
70
90
VPOS = 3V, IF = 10MHz
110 130 150 170 190
RF FREQUENCY – MHz
210
230
–10
0.1
250
4500
4.0
4000
3.5
80
70
CUBIC FIT OF IF_GAIN (TEMP)
60
2.5
2500
2.0
2000
1.5
1500
R SHUNT COMPONENT
1.0
1000
IF AMP GAIN
50
40
GAIN – dB
C SHUNT COMPONENT
3000
3.0
CAPACITANCE – pF
3500
30
20
10
CUBIC FIT OF CONV_GAIN (TEMP)
MIXER CG
0
0.5
500
0
0
50
100
150
200
250
300
350
400
450
–10
0
500
–20
–50 –40 –30 –20 –10
0
10 20 30 40 50 60 70 80 90 100 110 120 130
TEMPERATURE – ⴗC
FREQUENCY – MHz
TPC 14. Mixer Input Impedance vs. Frequency,
VPOS = 3 V, V GAIN = 0.8 V
TPC 17. Mixer Conversion Gain and IF Amplifier Gain vs.
Temperature, VPOS = 3 V, VGAIN = 0.3 V, VREF = 1.5 V, IF =
10.7 MHz, RF = 250 MHz
80
30
25
VGAIN = 0.00V
70
VGAIN = 0.54V
20
CUBIC FIT OF IF_GAIN (VPOS)
IF AMP GAIN
60
15
10
VGAIN = 1.62V
5
GAIN – dB
CONVERSION GAIN – dB
100
10
TPC 16. Mixer Conversion Gain vs. IF, T = 25 °C,
VPOS = 3 V, VREF = 1.5 V
TPC 13. Mixer Noise Figure vs. Frequency
RESISTANCE – ⍀
1
INTERMEDIATE FREQUENCY – MHz
VGAIN = 1.08V
0
–5
50
40
CUBIC FIT OF CONV_GAIN (VPOS)
30
–10
VGAIN = 2.16V
–15
20
10
–20
0
REV. 0
3 3.2 3.4 3.6 3.8 4 4.2 4.4 4.6 4.8
5 5.2 5.4 5.6 5.8 6
SUPPLY – Volts
RADIO FREQUENCY – MHz
TPC 15. Mixer Conversion Gain vs. Frequency,
T = 25 °C, VPOS = 2.7 V, VREF = 1.35 V, IF = 10.7 MHz
MIXER CG
2.4 2.6 2.8
50 100 150 200 250 300 350 400 450 500 550 600
TPC 18. Mixer Conversion Gain and IF Amplifier Gain vs.
Supply Voltage, T = 25 °C, VGAIN = 0.3 V, VREF = 1.5 V, IF =
10.7 MHz, RF = 250 MHz
–11–
AD61009
–90.00
80
VGAIN = 0.3V
70
–100.00
VGAIN = 0.6V
PHASE NOISE – dBc
IF AMPLIFIER GAIN – dB
60
50
VGAIN = 1.2V
40
30
VGAIN = 1.8V
20
–110.00
–120.00
–130.00
10
VGAIN = 2.4V
–140.00
0
–150.00
–10
0.1
1
10
100
1.00E+02
INTERMEDIATE FREQUENCY – MHz
1.00E+03
1.00E+04
1.00E+05
1.00E+06
1.00E+07
CARRIER FREQUENCY OFFSET, f(fm) – Hz
TPC 19. IF Amplifier Gain vs. Frequency,
T = 25 °C, VPOS = 3 V, VREF = 1.5 V
TPC 22. PLL Phase Noise L (F) vs. Frequency,
VPOS = 3 V, C3 = 0.1 µ F, IF = 10.7 MHz
2.5
10
8
FLTR PIN VOLTAGE – Volts
6
IF AMP
ERROR – dB
4
2
0
–2
MIXER
–4
2
–6
–8
1.5
0.1
–10
0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2
2.2 2.4 2.6 2.8 3
1
10
100
PLL FREQUENCY – MHz
GAIN VOLTAGE – Volts
TPC 20. Gain Error vs. Gain Control Voltage,
Representative Part
TPC 23. PLL Loop Voltage at FLTR (KVCO) vs. Frequency
8
7
6
COUNT
5
4
3
2
996.200␮s
1.00870ms
1.02120ms
TIMEBASE = 2.5␮s/DIV
DELAY
MEMORY 1 = 100.0mV/DIV
OFFSET = 127.3mV
TIMEBASE = 2.50␮s/DIV
DELAY
MEMORY 2 = 20.00mV/DIV
OFFSET = 155.2mV
TIMEBASE = 2.50␮s/DIV
DELAY
= 1.00870ms
STOP
= 1.01700ms
DELTA T
= 16.5199␮s
START
= 1.00048ms
= 1.00870ms
1
= 1.00870ms
0
85
86
87
88
89
90
91
92
93
QUADRATURE ANGLE – Degrees
94
95
TRIGGER ON EXTERNAL AT POS. EDGE AT 134.0mV
TPC 21. PLL Acquisition Time
TPC 24. Demodulator Quadrature Angle, Histogram,
T = 25 °C, VPOS = 3 V, IF = 10.7 MHz
–12–
REV. 0
AD61009
20
30
I_GAIN_CORR
19
25
18
17
IGAIN – dB
COUNT
20
15
10
CUBIC FIT OF I_GAIN_CORR (TEMP)
16
15
14
13
12
5
11
10
0
–2
–1
0
1
IQ GAIN BALANCE – dB
2.5
2
3.5
3
4
4.5
SUPPLY – Volts
5
5.5
6
TPC 28. Demodulator Gain vs. Supply Voltage
TPC 25. Demodulator Gain Balance, Histogram,
T = 25 °C, VPOS = 3 V, IF = 10.7 MHz
40
20
19
35
18
30
17
25
16
COUNT
IGAIN – dB
I_GAIN_CORR
15
QUADRATIC FIT OF I_GAIN_CORR (IFF)
14
20
15
13
10
12
5
11
10
0
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
17
17.2
17.4
17.6
BASEBAND FREQUENCY – MHz
17.8
18
18.2
18.4
18.6
18.8
DEMODULATOR GAIN – dB
TPC 26. Demodulator Gain vs. Frequency
TPC 29. Demodulator Gain Histogram,
T = 25 °C, VPOS = 3 V, IF = 10.7 MHz
20
I_GAIN_CORR
19
18
17
IGAIN – dB
CUBIC FIT OF I_GAIN_CORR (TEMP)
16
15
14
13
40.2127ms
12
11
10
–50 –40 –30 –20 –10 0
10 20 30 40 50 60 70 80 90 100 110 120 130
TEMPERATURE – ⴗC
40.2377ms
40.2627ms
TIMEBASE = 500␮s/DIV
DELAY
MEMORY 1 = 100.0mV/DIV
OFFSET = 154.0mV
TIMEBASE = 5.00␮s/DIV
DELAY
MEMORY 2 = 60.00mV/DIV
OFFSET = 209.0mV
TIMEBASE = 5.00␮s/DIV
DELAY
= 40.2377ms
STOP
= 40.2485ms
DELTA T
= 15.7990␮s
START
= 40.2327ms
= 40.2377ms
= 40.2377ms
TRIGGER ON EXTERNAL AT POS. EDGE AT 40.0mV
TPC 27. Demodulator Gain vs. Temperature
REV. 0
TPC 30. Power-Up Response Time to PLL Stable
–13–
AD61009
Mixer
SUPPLY CURRENT – mA
15
The UHF mixer is an improved Gilbert cell design, and can
operate from low frequencies (it is internally dc-coupled) up to
an RF input of 500 MHz. The dynamic range at the input of the
mixer is determined, at the upper end, by the maximum input
signal level of ± 56 mV between RFHI and RFLO up to which
the mixer remains linear, and, at the lower end, by the noise
level. It is customary to define the linearity of a mixer in terms
of the 1 dB gain-compression point and third-order intercept,
which for the AD61009 are –15 dBm and –8 dBm, respectively,
in a 50 Ω system.
10
The mixer’s RF input port is differential, that is, pin RFLO is
functionally identical to RFHI, and these nodes are internally
biased; we will generally assume that RFLO is decoupled to ac
ground. The RF port can be modeled as a parallel RC circuit as
shown in Figure 2.
5
0
0.5
1
1.5
GAIN VOLTAGE – Volts
2
2.5
TPC 31. Power Supply Current vs. Gain Control Voltage,
GREF = 1.5 V
AD61009
PRODUCT OVERVIEW
The AD61009 provides most of the active circuitry required to
realize a complete low power, single-conversion superheterodyne
receiver, or most of a double-conversion receiver, at input frequencies up to 500 MHz, and with an IF of from 400 kHz to
12 MHz. The internal I/Q demodulators, and their associated
phase locked-loop, which can provide carrier recovery from
the IF, support a wide variety of modulation modes, including
n-PSK, n-QAM, and AM. A single positive supply voltage of
3 V is required (2.85 V minimum, 5.5 V maximum) at a typical
supply current of 8.5 mA at midgain. In the following discussion, VP will be used to denote the power supply voltage, which
will be assumed to be 3 V.
C1
C2
RFHI
CIN
L1
RIN
RFLO
C3
C1, C2, L1: OPTIONAL MATCHING CIRCUIT
C3: COUPLES RFLO TO AC GROUND
Figure 2. Mixer Port Modeled as a Parallel RC Network; an
Optional Matching Network Is also Shown
Figure 1 shows the main sections of the AD61009. It consists of
a variable-gain UHF mixer and linear four-stage IF strip, which
together provide a voltage controlled gain range of more than
90 dB; followed by dual demodulators, each comprising a multiplier followed by a two-pole, 2 MHz low-pass filter; and driven
by a phase-locked loop providing the inphase and quadrature
clocks. A biasing system with CMOS compatible power-down
completes the AD61009.
The local oscillator (LO) input is internally biased at VP/2 via a
nominal 1000 Ω resistor internally connected from pin LOIP to
VMID. The LO interface includes a preamplifier which minimizes
the drive requirements, thus simplifying the oscillator design
and reducing LO leakage from the RF port. Internally, this
single-sided input is actually differential; the noninverting input
is referenced to pin VMID. The LO requires a single-sided drive
of ± 50 mV, or –16 dBm in a 50 Ω system.
LOIP
RFHI
VMID
IOUT
IFHI
MXOP
BPF
IFOP
RFLO
BPF OR
LPF
FDIN
DMIP
VQFO
FLTR
VMID
IFLO
QOUT
MIDPOINT
BIAS
GENERATOR
GAIN
VPS1
VPS2
BIAS
GENERATOR
PTAT
VOLTAGE
AD61009
GREF
PRUP
COM1
COM2
Figure 1. Functional Block Diagram
–14–
REV. 0
AD61009
The mixer’s output passes through both a low-pass filter and a
buffer, which provides an internal differential to single-ended
signal conversion with a bandwidth of approximately 45 MHz.
Its output at pin MXOP is in the form of a single-ended current.
This approach eliminates the 6 dB voltage loss of the usual series
termination by replacing it with shunt terminations at the both
the input and the output of the filter. The nominal conversion
gain is specified for operation into a total IF bandpass filter
(BPF) load of 165 Ω, that is, a 330 Ω filter, doubly-terminated
as shown in Figure 33. Note that these loads are connected to
bias point VMID, which is always at the midpoint of the supply
(that is, VP/2).
The maximum permissible signal level at MXOP is determined
by both voltage and current limitations. Using a 3 V supply and
VMID at 1.5 V, the maximum swing is about ± 1.3 V. To attain
a voltage swing of ± 1 V in the standard IF filter load of 165 Ω
load requires a peak drive current of about ± 6 mA, which is well
within the linear capability of the mixer. However, these upper
limits for voltage and current should not be confused with issues
related to the mixer gain, already discussed. In an operational
system, the AGC voltage will determine the mixer gain, and
hence the signal level at the IF input pin IFHI; it will always be
less than ± 56 mV (–15 dBm into 50 Ω), which is the limit of the
IF amplifier’s linear range.
The conversion gain is measured between the mixer input and
the input of this filter, and varies between 1.5 dB and 26.5 dB
for a 165 Ω load impedance. Using filters of higher impedance,
the conversion gain can always be maintained at its specified
value or made even higher; for filters of lower impedance, of say
ZO, the conversion gain will be lowered by 10 log10(165/ZO).
Thus, the use of a 50 Ω filter will result in a conversion gain that
is 5.2 dB lower. Figure 3 shows filter matching networks and
Table I lists resistor values.
IF Amplifier
R2
MXOP 8
1nF
BPF
R1
VMID
10 IFHI
R3
9
11 IFLO
100nF
100nF
Most of the gain in the AD61009 arises in the IF amplifier strip,
which comprises four stages. The first three are fully differential
and each has a gain span of 25 dB for the nominal AGC voltage
range. Thus, in conjunction with the mixer’s variable gain, the
total gain exceeds 90 dB. The final IF stage has a fixed gain of
20 dB, and it also provides differential to single-ended conversion.
The IF input is differential, at IFHI (noninverting relative to the
output IFOP) and IFLO (inverting). Figure 4 shows a simplified
schematic of the IF interface. The offset voltage of this stage
would cause a large dc output error at high gain, so it is nulled by
a low-pass feedback path from the IF output, also shown in
TPC 25. Unlike the mixer output, the signal at IFOP is a lowimpedance single-sided voltage, centered at VP/2 by the dc
feedback loop. It may be loaded by a resistance as low as 50 Ω,
which will normally be connected to VMID.
AD61009
IFHI
Figure 3. Suggested IF Filter Matching Network. The
Values of R1 and R2 Are Selected to Keep the Impedance
at Pin MXOP at 165 Ω
10k⍀
VMID
IFOP
IFLO
Table I. AD61009 Filter Termination Resistor Values for
Common IFs
IF
450 kHz
455 kHz
6.5 MHz
10.7 MHz
Filter
Impedance
Filter Termination Resistor
Values1 for 24 dB of Mixer Gain
1500 Ω
1500 Ω
1000 Ω
330 Ω
R1
174 Ω
174 Ω
215 Ω
330 Ω
R2
1330 Ω
1330 Ω
787 Ω
0Ω
NOTE
1
Resistor values were calculated such that R1 + R2 = Z FILTER and
R1储 (R2 + ZFILTER) = 165 Ω.
REV. 0
R3
1500 Ω
1500 Ω
1000 Ω
330 Ω
10k⍀
OFFSET FEEDBACK
LOOP
Figure 4. Simplified Schematic of the IF Interface
The IF’s small-signal bandwidth is approximately 45 MHz from
IFHI and IFLO through IFOP. The peak output at IFOP is
± 560 mV at VP = 3 V and ± 400 mV at the minimum VP of
2.7 V. This allows some headroom at the demodulator inputs
(pin DMIP), which accept a maximum input of ± 150 mV for
IFs > 3 MHz and ± 75 mV for IFs ≤ 3 MHz (at IFs ≤ 3 MHz,
the drive to the demodulators must be reduced to avoid saturating the output amplifiers with higher order mixing products that
are no longer removed by the onboard low-pass filters).
–15–
AD61009
Since there is no band-limiting in the IF strip, the outputreferred noise can be quite high; in a typical application and
at a gain of 75 dB it is about 100 mV rms, making post-IF filtering
desirable. IFOP may be also used as an IF output for driving
an A/D converter, external demodulator, or external AGC
detector. Figure 5 shows methods of matching the optional
second IF filter.
VPOS
AD61009
2RT
RT
IFOP
BPF
2RT
DMIP
a. Biasing DMIP from Power Supply (Assumes BPF AC
Coupled Internally)
AD61009
Alternatively, pin GREF can be tied to an external voltage
reference, VR, provided, for example, by an AD1582 (2.5 V)
or AD1580 (1.21 V) voltage reference, to provide supplyindependent gain scaling of VR/75 (volts per dB). Since it uses
the same reference voltage, the numerical input to this DAC
provides an accurate RSSI value in digital form, no longer
requiring the reference voltage to have high absolute accuracy.
I/Q Demodulators
RT
IFOP
AD61009 features temperature-compensation of the gain scaling. The gain control scaling is proportional to the reference
voltage applied to the pin GREF. When this pin is tied to the
midpoint of the supply (VMID), the scale is nominally 20 mV/
dB (50 dB/V) for VP = 3 V. Under these conditions, the lower
80 dB of gain range (mixer plus IF) corresponds to a control
voltage of 0.4 V ≤ VG ≤ 2.0 V. The final centering of this 1.6 V
range depends on the insertion losses of the IF filters used. More
generally, the gain scaling using these connections is VP/150
(volts per dB), so becomes 33.3 mV/dB (30 dB/V) using a 5 V
supply, with a proportional change in the AGC range, to 0.33 V ≤
VG ≤ 3 V, Table II lists gain control voltages and scale factors
for power supply voltages from 3 V to 5.5 V.
BPF
DMIP
RT
VMID
CBYPASS
b. Biasing DMIP from VMID (Assumes BPF AC Coupled
Internally)
Figure 5. Input and Output Matching of the Optional
Second IF Filter
Gain Scaling and RSSI
The AD61009’s overall gain, expressed in decibels, is linearin-dB with respect to the AGC voltage VG at pin GAIN. The
gain of all sections is maximum when VG is zero, and reduces
progressively up to VG = 2.2 V (for VP = 3 V; in general, up to a
limit VP – 0.8 V). The gain of all stages changes in parallel. The
Both demodulators (I and Q) receive their inputs at pin DMIP.
Internally, this single-sided input is actually differential; the
noninverting input is referenced to pin VMID. Each demodulator comprises a full-wave synchronous detector followed by a
2 MHz, two-pole low-pass filter, producing single-sided outputs
at pins IOUT and QOT. Using the I and Q demodulators for
IFs above 12 MHz is precluded by the 1 MHz to 12 MHz
response of the PLL used in the demodulator section. Pin DMIP
requires an external bias source at VP/2; Figure 6 shows suggested methods.
Outputs IOUT and QOUT are centered at VP/2 and can swing
up to ± 1.23 V even at the low supply voltage of 2.85 V. The
conversion gain of the I and Q demodulators is 18 dB (X8),
requiring a maximum input amplitude at DMIP of ± 150 mV
for IFs > 3 MHz.
Table II. AD61009 Gain and Manual Gain Control Voltage vs. Power Supply Voltage
Power Supply
Voltage
(V)
GREF
(= VMID)
(V)
Scale Factor
(dB/V)
Scale Factor
(mV/dB)
Gain Control
Voltage Input Range
(V)
3.0
3.5
4.0
4.5
5.0
5.5
1.5
1.75
2.0
2.25
2.5
2.75
50.00
42.86
37.50
33.33
30.00
27.27
20.00
23.33
26.67
30.00
33.33
36.67
0.400–2.000
0.467–2.333
0.533–2.667
0.600–3.000
0.667–3.333
0.733–3.667
NOTE
Maximum gain occurs for gain control voltage = 0 V.
–16–
REV. 0
AD61009
The reference signal may be provided from an external source,
in the form of a high-level clock, typically a low level signal
(± 400 mV) since there is an input amplifier between FDIN and
the loop’s phase detector. For example, the IF output itself can
be used by connecting DMIP to FDIN, which will then provide
automatic carrier recover for synchronous AM detection and
take advantage of any post-IF filtering. Pin FDIN must be
biased at VP/2; Figure 9 shows suggested methods.
VPOS
AD61009
2RT
RT
IFOP
BPF
2RT
DMIP
a. Biasing DMIP from Power Supply (Assumes BPF
AC-Coupled Internally)
AD61009
IFOP
RT
BPF
In practice, the probability of a phase mismatch at power-up is
high, so the worst-case linear settling period to full lock needs to
be considered in making filter choices. This is typically 16.5 µs at
an IF of 10.7 MHz for a ±100 mV signal at DMIP and FDIN.
DMIP
RT
VMID
The VFQO operates from 1 MHz to 12 MHz and is controlled by
the voltage between VPOS and FLTR. In normal operation, a
series RC network, forming the PLL loop filter, is connected
from FLTR to ground. The use of an integral sample-hold
system ensures that the frequency-control voltage on pin FLTR
remains held during power-down, so reacquisition of the carrier
typically occurs in 16.5 µs.
CBYPASS
Bias System
b. Biasing DMIP from VMID (Assumes BPF
AC-Coupled Internally)
Figure 6. Suggested Methods for Biasing Pin DMIP
at VP /2
For IFs < 3 MHz, the on-chip low-pass filters (2 MHz cutoff)
do not attenuate the IF or feedthrough products; thus, the
maximum input voltage at DMIP must be limited to ± 75 mV
to allow sufficient headroom at the I and Q outputs for not only
the desired baseband signal but also the unattenuated higherorder demodulation products. These products can be removed
by an external low-pass filter.
Phase-Locked Loop
The demodulators are driven by quadrature signals that are
provided by a variable frequency quadrature oscillator (VFQO),
phase locked to a reference signal applied to pin FDIN. When
this signal is at the IF, inphase and quadrature baseband outputs
are generated at IOUT and QOUT, respectively. The quadrature accuracy of this VFQO is typically –1.2° at 10.7 MHz. The
PLL uses a sequential-phase detector that comprises low power
emitter-coupled logic and a charge pump (Figure 7).
The AD61009 operates from a single supply, VP, usually of 3 V,
at a typical supply current of 8.5 mA at midgain and T = 27°C,
corresponding to a power consumption of 25 mW. Any voltage
from 2.85 V to 5.5 V may be used.
The bias system includes a fast-acting active-high CMOScompatible power-up switch, allowing the part to idle at 550 µA
when disabled. Biasing is proportional-to-absolute-temperature
(PTAT) to ensure stable gain with temperature.
An independent regulator generates a voltage at the midpoint
of the supply (VP/2) which appears at the VMID pin, at a low
impedance. This voltage does not shut down, ensuring that the
major signal interfaces (e.g., mixer-to-IF and IF-to-demodulators)
remain biased at all times, thus minimizing transient disturbances
at power-up and allowing the use of substantial decoupling
capacitors on this node. The quiescent consumption of this
regulator is included in the idling current.
VPOS
FDIN
EXTERNAL
FREQUENCY
REFERENCE
VF
R
SEQUENTIAL
PHASE
DETECTOR
U
D
C
ID~
40␮A
50k⍀
a. Biasing FDIN from Supply when Using
External Frequency Reference
IU~
40␮A
F
AD61009
50k⍀
R
I-CLOCK
VARIABLEFREQUENCY
QUADRATURE
OSCILLATOR
AD61009
90ⴗ
FDIN
EXTERNAL
FREQUENCY
REFERENCE
Q-CLOCK
(ECL OUTPUTS)
50k⍀
VMID
CBYPASS
REFERENCE CARRIER
(FDIN AFTER LIMITING)
Figure 7. Simplified Schematic of the PLL and
Quadrature VCO
b. Biasing FDIN from VMID when Using
External Frequency Reference
Figure 8. Suggested Methods for Biasing Pin FDIN
at VP / 2
REV. 0
–17–
AD61009
USING THE AD61009
90dB
In this section, we will focus on a few areas of special importance and include a few general application tips. As is true of
any wideband high gain component, great care is needed in PC
board layout. The location of the particular grounding points
must be considered with due regard to possibility of unwanted
signal coupling, particularly from IFOP to RFHI or IFHI or both.
80dB
70dB
(67.5dB)
60dB
IF GAIN
50dB
The high sensitivity of the AD61009 leads to the possibility that
unwanted local EM signals may have an effect on the performance.
During system development, carefully-shielded test assemblies
should be used. The best solution is to use a fully-enclosed
box enclosing all components, with the minimum number of
needed signal connectors (RF, LO, I and Q outputs) in miniature coax form.
40dB
30dB
(21.5dB)
20dB
MIXER GAIN
10dB
(7.5dB)
(1.5dB)
0dB
0.4V
0
The I and Q output leads can include small series resistors
(about 100 Ω) inside the shielded box without significant loss
of performance, provided the external loading during testing
is light (that is, a resistive load of more than 20 kΩ and capacitances of a few picofarads). These help to keep unwanted RF
emanations out of the interior.
1.8V
1V
NORMAL OPERATING RANGE
2.2V
2V
Vg
Figure 10. Gain Distribution for GREF = 1.5 V
90dB
80dB
70dB
The power supply should be connected via a through-hole
capacitor with a ferrite bead on both inside and outside leads.
Close to the IC pins, two capacitors of different value should be
used to decouple the main supply (VP) and the midpoint supply
pin, VMID. Guidance on these matters is also generally included
in applications schematics.
(67.5dB)
60dB
50dB
IF GAIN
40dB
30dB
Gain Distribution
(21.5dB)
20dB
As in all receivers, the most critical decisions in effectively using
the AD61009 relate to the partitioning of gain between the
various subsections (Mixer, IF Amplifier, Demodulators) and
the placement of filters, so as to achieve the highest overall signalto-noise ratio and lowest intermodulation distortion.
MIXER GAIN
10dB
(7.5dB)
(1.5dB)
0dB
0
0.328V
1.64V
1V
NORMAL OPERATING RANGE
Figure 9 shows the main RF/IF signal path at maximum and
minimum signal levels.
2V
Vg
Figure 11. Gain Distribution for GREF = 1.23 V
As noted earlier, the gain in dB is reduced linearly with the
voltage VG on the GAIN pin. Figure 10 shows how the mixer
and IF strip gains vary with VG when GREF is connected to VMID
(1.5 V) and a supply voltage of 3 V is used. Figure 11 shows
how these vary when GREF is connected to a 1.23 V reference.
I
ⴞ54mV
MAX INPUT
ⴞ1.3V
MAX OUTPUT
RFHI
MXOP
ⴞ54mV
MAX INPUT
ⴞ560mV
MAX OUTPUT
IFOP
IFHI
IF BPF
LOIP
CONSTANT
–16dBm
(ⴞ50mV)
330⍀
330⍀
(TYPICAL
IMPEDANCE)
ⴞ154mV
MAX INPUT
ⴞ1.23V
MAX OUTPUT
IOUT
DMIP
QOUT
IF BPF
(VMID)
Q
(LOCATION OF OPTIONAL
SECOND IF FILTER)
Figure 9. Signal Levels for Minimum and Maximum Gain
–18–
REV. 0
AD61009
Using the AD61009 with a Fast PRUP Control Signal
If the AD61009 is used in a system in which the PRUP signal
(Pin 3) is applied with a rise time less than 35 µs, anomalous
behavior occasionally occurs. The problem is intermittent, so it
will not occur every time the part is powered up under these
conditions. It does not occur for any other normal operating conditions when the PRUP signal has a rise time slower than 35 µs.
Symptoms of operation with too fast a PRUP signal include low
gain, oscillations at the I or Q outputs of the device or no valid
data occurring at the output of the AD61009. The problem
causes no permanent damage to the AD61009, so it will often
operate normally when reset.
Fortunately, there is a very simple solution to the fast PRUP
problem. If the PRUP signal (Pin 3) is slowed down so that
the rise time of the signal edge is greater than 35 µs, the anomalous behavior will not occur. This can be realized by a simple
RC circuit connected to the PRUP pin, where R = 4.7 kΩ and
C = 1.5 nF. This circuit is shown in Figure 12.
FROM PRUP
CONTROL SIGNAL
AD61009
4.7k⍀
PRUP
1.5nF
All designs incorporating the AD61009 should include
this circuitry.
Note that connecting the PRUP pin to the supply voltage will
not eliminate the problem since the supply voltage may have a
rise time faster than 35 µs. With this configuration, the 4.7 kΩ
series R and 1.5 nF shunt C should be placed between the
supply and the PRUP pin as shown in Figure 12.
AD61009 EVALUATION BOARD
The AD61009 evaluation board (Figures 13 and 14) consists of
an AD61009, ground plane, I/O connectors, and a 10.7 MHz
bandpass filter. The RF and LO ports are terminated in 50 Ω
to provide a broadband match to external signal generators to
allow a choice of RF and LO input frequencies. The IF filter is
at 10.7 MHz and has 330 Ω input and output terminations; the
board is laid out to allow the user to substitute other filters for
other IFs.
The board provides SMA connectors for the RF and LO port
inputs, the demodulated I and Q outputs, the manual gain control (MGC) input, the PLL input, and the power-up input. In
addition, the IF output is also available at an SMA connector;
this may be connected to the PLL input for carrier recovery to
realize synchronous AM and FM detection via the I and Q
demodulators, respectively. Table III lists the AD61009 Evaluation Board’s I/O Connectors and their functions.
Figure 12. Proper Configuration of AD61009 PRUP Signal
REV. 0
–19–
AD61009
VPOS
C15
0.1␮F
GND
JUMPER
C11
10nF
FDIN
R8
51.1⍀
R11
OPEN
C12
0.1␮F
C1
0.1␮F
FDIN
COM1
R12
4.7k⍀
PRUP
C13 0
C10
1nF
VPS1
PRUP
R7
51.1⍀
C14 0
R6
51.1⍀
1k⍀
IOUT
C17
1.5nF
C16 1nF
C9
1nF
R5
RFLO
JUMPER
R4
OPEN
332⍀
R3
332⍀
C3 10nF
R1
FLTR
I
QOUT
LOIP
LO
RF
R10
4.99k⍀
C20.1␮F
AD61009 VPS2
RFHI
DMIP
GREF
IFOP
MXOP
COM2
C4
47pF
R2
316⍀
IF
GAIN
VMID
IFHI
Q
C5
1nF
IFLO
GAIN
C6
0.1␮F
C7
1nF
C8
0.1␮F
AD607 EVALUATION BOARD
(AS RECEIVED)
VPOS
R15
50k⍀
R13
50k⍀
C18
SHORT
FDIN
FDIN
R14
51.1⍀
C17
10nF
R12
OPEN
VMID
MOD FOR LARGE MAGNITUDE
AC-COUPLED INPUT
VPOS
R17
OPEN
R18
OPEN
C19
ANYTHING
FDIN
FDIN
R19
RSOURCE
C20
SHORT
R16
OPEN
VMID
MOD FOR DC-COUPLED INPUT
Figure 13. Evaluation Board
–20–
REV. 0
AD61009
FDIN
C15
J10
R8
PRUP
R11
C1
R12
R1
C11
C17
C13
LOIP
C12
R10
C10
R7
C16
C14
C9
R6
J9
C3
C3'
U1
C2
R2
C4
R4
R5
C6
R3
C8
IFOP
C7
RFHI
FILT
ANALOG
DEVICES
AD61009
EVALUATION BOARD REV B
a. Topside
IOUT
R9
QOUT
C5
GAIN
b. Bottom Side
Figure 14. Evaluation Board Layout
REV. 0
–21–
AD61009
Table III. AD61009 Evaluation Board Input and Output Connections
Reference
Designation
Connector
Type
Description
Coupling
Approximate
Signal Level
J1
SMA
Frequency
Detector Input
DC
± 400 mV
J2
SMA
Power Up
DC
J3
SMA
LO Input
AC
J4
SMA
RF Input
AC
J5
SMA
MGC Input
DC
J6
SMA
IF Output
AC
CMOS Logic
Level Input
–16 dBm
(± 50 mV)
–15 dBm max
(± 54 mV)
0.4 V to 2.0 V
(3 V Supply)
(GREF = VMID)
NA
J7
SMA
Q Output
AC
NA
J8
SMA
I Output
AC
NA
J9
Jumper
Ties GREF
to VMID
NA
NA
J10
Jumper
NA
NA
T1
Terminal Pin
DC
DC
T2
Terminal Pin
Ties Power-Up
to Positive
Supply
Power Supply
Positive Input
(VPS1, VPS2)
Power Supply
Return (GND)
DC
0V
–22–
Comments
This pin needs to be biased at VMID
and ac coupled when driven by an
external signal generator.
Tied to Positive Supply by Jumper J10.
Input is terminated in 50 Ω.
Input is terminated in 50 Ω.
Jumper is set for Manual Gain Control
Input; See Table I for Control Voltage
Values.
This signal level depends on the
AD61009’s gain setting.
This signal level depends on the
AD61009’s gain setting.
This signal level depends on the
AD61009’s gain setting.
Sets gain-control Scale Factor (SF);
SF = 75/VMID in dB/V, where
VMID = VPOS/2.
Remove to test Power-Up/-Down.
2.85 V to 5.5 V
Draws 8.5 mA at midgain connection.
REV. 0
AD61009
In operation (Figure 15), the AD61009 evaluation board draws
about 8.5 mA at midgain (59 dB). Use high impedance probes
to monitor signals from the demodulated I and Q outputs and
the IF output. The MGC voltage should be set such that the
signal level at DMIP does not exceed ± 150 mV; signal levels
above this will overload the I and Q demodulators. The insertion loss between IFOP and DMIP is typically 3 dB if a simple
low-pass filter (R8 and C2) is used and higher if a reverseterminated bandpass filter is used.
HP 6632A
PROGRAMMABLE
POWER SUPPLY
2.7V–6V
FLUKE 6082A
SYNTHESIZED
SIGNAL GENERATOR
240MHz
MCL
ZFSC–2–1
COMBINER
HP 8656A
SYNTHESIZED
SIGNAL GENERATOR
240.02MHz
HP 9920
IEEE CONTROLLER
HP9121
DISK DRIVE
HP 3326
SYNTHESIZED
SIGNAL GENERATOR
10.710MHz
VPOS
RF
FDIN
I OUTPUT
AD607
EVALUATION
BOARD
LO
Q OUTPUT
TEKTRONIX
11402A
OSCILLOSCOPE
WITH 11A32
PLUGIN
MGC
HP 8656A
SYNTHESIZED
SIGNAL GENERATOR
229.3MHz
DATA PRECISION
DVC8200
PROGRAMMABLE
VOLTAGE SOURCE
IEEE–488 BUS
Figure 15. Evaluation Board Test Setup
REV. 0
–23–
AD61009
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
20-Lead Plastic SSOP (RS-20)
11
1
10
0.07 (1.78)
0.066 (1.67)
0.0256
(0.65)
BSC
8°
SEATING 0.009 (0.229) 0°
PLANE
0.005 (0.127)
0.037 (0.94)
0.022 (0.559)
PRINTED IN U.S.A.
0.078 (1.98) PIN 1
0.068 (1.73)
0.008 (0.203)
0.002 (0.050)
C02347–0–1/01 (rev. 0)
20
0.212 (5.38)
0.205 (5.21)
0.311 (7.9)
0.301 (7.64)
0.295 (7.50)
0.271 (6.90)
–24–
REV. 0
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